JP6428334B2 - Switched reluctance motor controller - Google Patents

Description

  The present invention relates to a control device applied to a switched reluctance motor having a plurality of windings that can be magnetically coupled to each other.

  This type of control device performs non-interference control that cancels the influence of mutual magnetic interference between a plurality of windings when a current flows through a plurality of windings constituting a switched reluctance motor (hereinafter referred to as SRM). It has been known. Specifically, non-interference control is performed based on a motor model including the self-inductance of each of the plurality of windings and the mutual inductance between the plurality of windings. In this model, each of the self-inductance and the mutual inductance is a parameter that depends on the current flowing in the self-phase (see Non-Patent Document 1 below). Thereby, high response of current control in the SRM is achieved.

Han-Kyung Bae, et. Al, “A Novel Approach to Control of Switched Reluctance Motors Considering Mutual Inductance” IEEE Industrial Electronics Society. IECON2000, Vol.1, pp.369-374

  Here, even when the above-described non-interference control is performed, the present inventors faced a situation in which the current controllability in the SRM is greatly deteriorated, for example, the current flowing through the winding is greatly deviated from the command value. For this reason, there is still room for improvement in the technology for improving the controllability of the current (control amount) of the SRM.

  The main object of the present invention is to provide a control device for a switched reluctance motor capable of improving the controllability of the control amount of the switched reluctance motor.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

  The present invention includes a switched reluctance motor having a plurality of windings (22A, 22B, 22C) that can be magnetically coupled to each other, and a voltage applying means (20) that applies a voltage to each of the plurality of windings. Applied to a motor control system, a current flowing through the winding or a magnetic flux interlinking with the winding is set as a control amount, and the control amount of each of the plurality of windings is controlled to the command value. Command voltage calculation means (35A to 35C, 52A to 52C) for calculating a command voltage applied to each of the windings, and each of the plurality of windings based on the command voltage calculated by the command voltage calculation means Operating means for operating the voltage applying means to apply a voltage to the winding, and among the plurality of windings, one winding to be calculated for the command voltage is a self-phase winding, and the remainder The winding is an other-phase winding, and the command voltage calculating means includes a self-inductance of the self-phase winding and a mutual inductance between the self-phase winding and the other-phase winding. The command voltage is calculated based on a model of the motor that depends on all the currents flowing through the windings.

  The present inventor, for each of a plurality of windings, each of the self-inductance of the self-phase winding and the mutual inductance between the self-phase winding and the other-phase winding flows in each of the plurality of windings. The knowledge that it depends on everything was acquired. And according to the model of the motor in which each of the self-inductance of the self-phase winding and the mutual inductance between the self-phase winding and the other-phase winding depends on all the currents flowing in each of the plurality of windings, In addition to the current flowing in the self-phase winding, we found that the effect of magnetic saturation due to the current flowing in the other-phase winding can be quantified with high accuracy.

  Therefore, in the above invention, a command voltage to be applied to each of the plurality of windings is calculated based on a motor model in which each of the self-inductance and the mutual inductance depends on all the currents flowing in each of the plurality of windings. Based on the calculated command voltage, a voltage is applied to each of the plurality of windings by operating the voltage applying unit. Thereby, in each of a some winding, it can avoid that control amount deviates large from command value. Therefore, the controllability of the motor control amount can be improved.

1 is an overall configuration diagram of a motor control system according to a first embodiment. Sectional drawing which shows the outline of SRM. The figure which shows the electricity supply aspect of all the volume winding SRM. The block diagram which shows the outline | summary of electric current feedback control. The figure for demonstrating the setting method of a feedback gain. The time chart which shows the current feedback control when the electric current concerning related technology is small. The time chart which shows the current feedback control in case the electric current concerning related technology is large. The time chart which shows the current feedback control in case the electric current concerning 1st Embodiment is small. The time chart which shows the current feedback control in case the electric current concerning 1st Embodiment is large. The block diagram which shows the outline | summary of the magnetic flux feedback control concerning 2nd Embodiment. The figure which shows the relationship between the electrical angular velocity concerning 3rd Embodiment, and each component of interference voltage. The flowchart which shows an interference voltage calculation process. The figure which shows the electricity supply aspect of the single node winding SRM concerning other embodiment.

(First embodiment)
Hereinafter, a first embodiment in which a control device according to the present invention is applied to a full-pitch switched reluctance motor as an in-vehicle main machine will be described with reference to the drawings.

  As shown in FIG. 1, a battery 10 as a DC power source is a secondary battery having a voltage between terminals of several hundred volts, for example. As the battery 10, for example, a lithium ion secondary battery or a nickel hydride secondary battery can be used.

  A power conversion circuit 20 (corresponding to “voltage application means”) is connected to the battery 10 via a smoothing capacitor 12. A motor generator as an in-vehicle main machine is connected to the power conversion circuit 20. In the present embodiment, a three-phase full-pitch switched reluctance motor including an A-phase winding 22A, a B-phase winding 22B, and a C-phase winding 22C is used as the motor generator. Hereinafter, the switched reluctance motor is simply referred to as SRM.

  The A phase of the power conversion circuit 20 includes a series connection of an A phase upper arm switch SAp and an A phase lower arm diode DAn, and a series connection of an A phase upper arm diode DAp and an A phase lower arm switch SAn. . The B phase of the power conversion circuit 20 includes a series connection body of a B phase upper arm switch SBp and a B phase lower arm diode DBn, and a series connection body of a B phase upper arm diode DBp and a B phase lower arm switch SBn. . The C phase of the power conversion circuit 20 includes a series connection body of a C phase upper arm switch SCp and a C phase lower arm diode DCn, and a series connection body of a C phase upper arm diode DCp and a C phase lower arm switch SCn. . In the present embodiment, IGBTs are used as the A to C phase upper arm switches SAp, SBp, SCp and the A to C phase lower arm switches SAn, SBn, SCn.

  Here, the connection mode of the A phase will be described in detail. A connection point between the A-phase upper arm switch SAp and the A-phase lower arm diode DAn and a connection point between the A-phase upper arm diode DAp and the A-phase lower arm switch SAn are connected via the A-phase winding 22A. The emitter of the A-phase upper arm switch SAp and the cathode of the A-phase lower arm diode DAn are connected to each other, and the collector of the A-phase upper arm switch SAp is connected to the positive terminal of the battery 10. The anode of the A-phase lower arm diode DAn is connected to the negative terminal of the battery 10. On the other hand, the anode of the A-phase upper arm diode DAp and the collector of the A-phase lower arm switch SAn are connected to each other, and the cathode of the A-phase upper arm diode DAp is connected to the positive terminal of the battery 10. The emitter of the A-phase lower arm switch SAn is connected to the negative terminal of the battery 10.

  In addition, the connection aspect of B phase and C phase is the same as that of A phase. For this reason, in this embodiment, the detailed description of the connection aspect about B phase and C phase is abbreviate | omitted.

  The control device 30 includes a central processing unit (CPU) and a storage unit (memory) (not shown), and executes various programs stored in the memory by the central processing unit, so that a torque, which is a control amount of the SRM, can be commanded. The torque command value T *, which is a value, is controlled. The detection value of the rotation angle sensor 40 that detects the electrical angle of the rotor 26 of the SRM is input to the control device 30. The electrical angle is a value that changes in a range of 0 ° to less than 360 °. The control device 30 also detects an A-phase current sensor 42A that detects a current flowing through the A-phase winding 22A (hereinafter referred to as A-phase current) and a current (hereinafter referred to as B-phase current) that flows through the B-phase winding 22B. The detected value of the phase B current sensor 42B and the phase C current sensor 42C for detecting the current flowing in the phase C winding 22C (hereinafter referred to as phase C current) are input. In the present embodiment, the A, B, C phase current sensors 42A, 42B, 42C correspond to “current detection means”.

  Based on the detection values of these various sensors, the control device 30 controls each switch SAp, SAn, SBp, SBn, SCp, SCn constituting the power conversion circuit 20 to control the torque of the SRM to the torque command value T *. On the other hand, each switch SAp, SAn, SBp, SBn, SCp, SCn is operated by outputting operation signals gAp, gAn, gBp, gBn, gCp, gCn for instructing switching between the on state and the off state.

  Specifically, the control device 30 includes a speed calculation unit 31 and an operation signal generation unit 32. The speed calculation unit 31 calculates the electrical angular speed ω by time differentiation of the electrical angle θ detected by the rotation angle sensor 40. The operation signal generator 32 operates the switches SAp, SAn, SBp, SBn, SCp, SCn based on the torque command value T *, the electrical angle θ, the electrical angular velocity ω, and the phase currents Ia, Ib, Ic. Operation signals gAp, gAn, gBp, gBn, gCp, and gCn are generated. In the present embodiment, the operation signal generator 32 includes “operation means”.

  Incidentally, the torque command value T * is input to the control device 30 from, for example, a control device higher than the control device 30 (for example, a control device that supervises vehicle travel control). In addition, each phase of the SRM is independent, and furthermore, each process related to each phase in the control device 30 is basically the same process.

  FIG. 2 shows an outline of the whole volume SRM. The full volume winding SRM includes a cylindrical stator 24 and a columnar rotor 26. In the present embodiment, six salient poles 25, that is, salient poles 25 that are twice the number of phases are provided on the inner surface of the stator 24 at equal intervals in the circumferential direction. On the other hand, on the outside of the rotor 26, four salient poles 27 are provided at equal intervals in the circumferential direction. In the present embodiment, each of the A-phase winding 22A, the B-phase winding 22B, and the C-phase winding 22C is provided between the salient poles 25 of the stator 24 and is opposed to the rotational direction by a mechanical angle of 180 °. Is concentrated in the slot. The winding is wound so that the energization direction is opposite in one slot and the other slot. In addition, the slots adjacent to each other in the circumferential direction are wound so that the energization direction is opposite.

  FIG. 3 shows an energization mode for each phase winding 22A, 22B, 22C of the full-pitch winding SRM. Specifically, FIG. 3A shows the transition of the electrical angle θ, and FIGS. 3B to 3D show the transition of the A, B, C phase currents Ia, Ib, Ic.

  As shown in the figure, in the full-pitch winding SRM, each of the A-phase winding 22A, the B-phase winding 22B, and the C-phase winding 22C is out of phase with each other by 120 °, and at least two windings are energized. The voltage is applied in the order of the A-phase winding 22A, the B-phase winding 22B, and the C-phase winding 22C so that a part of the period overlaps.

  By the way, the magnetic flux generated by energizing any one of the phase windings 22A, 22B, and 22C is also linked to the remaining windings. For example, if the B-phase winding 22B is energized by applying a positive voltage, the magnetic flux generated by the energization is also linked to the A-phase winding 22A and the C-phase winding 22C whose order of energization is changed. Inductive electromotive force (interference voltage) is generated in the A and C phase windings 22A and 22B. For this reason, there is a concern that the current controllability of the SRM may be reduced by the interference voltage during a period in which at least two of the phase windings 22A, 22B, and 22C are energized. In particular, in the full-pitch winding SRM, since there is a period in which all of the A, B, and C phase windings 22A, 22B, and 22C are energized, there is a concern that the current controllability is significantly reduced.

  Therefore, in the present embodiment, current feedback control is performed so as to suppress the influence of the interference voltage on the current controllability. Hereinafter, after describing the design method of the control system, the current feedback control performed by the operation signal generation unit 32 will be described.

  First, a control system design method will be described.

  The voltage equation of the three-phase SRM is expressed by the following equation (eq1).

上式（ｅｑ１）において、Ｖａ，Ｖｂ，ＶｃはＡ，Ｂ，Ｃ相巻線２２Ａ，２２Ｂ，２２Ｃの印加電圧を示し、Ａ，Ｂ，Ｃ相巻線２２Ａ，２２Ｂ，２２Ｃの巻線抵抗を示し、Ｉａ，Ｉｂ，ＩｃはＡ，Ｂ，Ｃ相電流を示し、λａ，λｂ，λｃはＡ，Ｂ，Ｃ相巻線２２Ａ，２２Ｂ，２２Ｃの鎖交磁束を示す。ここで本実施形態では、Ａ，Ｂ，Ｃ相鎖交磁束λａ，λｂ，λｃを下式（ｅｑ２）で表す。

In the above equation (eq1), Va, Vb, and Vc indicate the voltages applied to the A, B, and C phase windings 22A, 22B, and 22C, and the winding resistances of the A, B, and C phase windings 22A, 22B, and 22C Ia, Ib, and Ic indicate A, B, and C phase currents, and λa, λb, and λc indicate interlinkage magnetic fluxes of the A, B, and C phase windings 22A, 22B, and 22C. Here, in the present embodiment, the A, B, C phase interlinkage magnetic fluxes λa, λb, λc are expressed by the following equation (eq2).

上式（ｅｑ２）において、Ｌａ，Ｌｂ，ＬｃはＡ，Ｂ，Ｃ相巻線２２Ａ，２２Ｂ，２２Ｃの自己インダクタンスを示し、Ｍｉｊ（ｉ，ｊ＝Ａ，Ｂ，Ｃであり、ｉ≠ｊ）は、ｊ相巻線２２ｊに対するｉ相巻線２２ｉの相互インダクタンスを示す。ここで本実施形態では、各インダクタンスＬａ，Ｌｂ，Ｌｃ，Ｍａｂ，Ｍａｃ，Ｍｂａ，Ｍｂｃ，Ｍｃａ，Ｍｃｂの全てが、Ａ，Ｂ，Ｃ相電流Ｉａ，Ｉｂ，Ｉｃと電気角θとの関数として表されるとする。この場合、各鎖交磁束λａ，λｂ，λｃの時間微分値は下式（ｅｑ３）で表される。

In the above equation (eq2), La, Lb, and Lc indicate the self-inductances of the A, B, and C phase windings 22A, 22B, and 22C, and Mij (i, j = A, B, C, i ≠ j). Indicates the mutual inductance of the i-phase winding 22i with respect to the j-phase winding 22j. Here, in the present embodiment, all the inductances La, Lb, Lc, Mab, Mac, Mba, Mbc, Mca, Mcb are all functions as a function of the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ. Let it be represented. In this case, the time differential values of the interlinkage magnetic fluxes λa, λb, and λc are expressed by the following equation (eq3).

上式（ｅｑ３）に上式（ｅｑ２）を代入すると、下式（ｅｑ４）が導かれる。

Substituting the above equation (eq2) into the above equation (eq3) leads to the following equation (eq4).

上式（ｅｑ４）のＡ相について、下式（ｅｑ５）のように記号を置き換える。

For the A phase of the above formula (eq4), the symbols are replaced as in the following formula (eq5).

上式（ｅｑ４）のＢ相について、下式（ｅｑ６）のように記号を置き換える。

For the B phase of the above formula (eq4), the symbols are replaced as in the following formula (eq6).

上式（ｅｑ４）のＣ相について、下式（ｅｑ７）のように記号を置き換える。

For the C phase of the above formula (eq4), the symbols are replaced as in the following formula (eq7).

上式（ｅｑ２）を上式（ｅｑ１）に代入し、各相電流Ｉａ，Ｉｂ，Ｉｃについて解くと、下式（ｅｑ８）〜（ｅｑ１０）が導かれる。

Substituting the above equation (eq2) into the above equation (eq1) and solving for each phase current Ia, Ib, Ic, the following equations (eq8) to (eq10) are derived.

上式（ｅｑ８）を下式（ｅｑ１１）のように変形する。

The above equation (eq8) is transformed into the following equation (eq11).

上式（ｅｑ１１）において、ＶＩＮＡはＡ相巻線２２Ａに印加される干渉電圧を示す。

In the above equation (eq11), VINA represents an interference voltage applied to the A-phase winding 22A.

  The above equation (eq9) is transformed into the following equation (eq12).

上式（ｅｑ１２）において、ＶＩＮＢはＢ相巻線２２Ｂに印加される干渉電圧を示す。

In the above equation (eq12), VINB represents an interference voltage applied to the B-phase winding 22B.

  The above equation (eq10) is transformed into the following equation (eq13).

上式（ｅｑ１３）において、ＶＩＮＣはＣ相巻線２２Ｃに印加される干渉電圧を示す。上式（ｅｑ１１）〜（ｅｑ１３）から、図４に示すＳＲＭのモデルが定まる。このモータモデルは、巻線の印加電圧を入力とし、制御量である電流を出力とするモデルである。モデルでは、Ａ，Ｂ，Ｃ相巻線２２Ａ，２２Ｂ，２２Ｃに印加されるＡ，Ｂ，Ｃ相干渉電圧ＶＩＮＡ，ＶＩＮＢ，ＶＩＮＣが示されている。本実施形態では、各干渉電圧ＶＩＮＡ，ＶＩＮＢ，ＶＩＮＣの影響を除去する非干渉制御を行う。

In the above equation (eq13), VINC represents an interference voltage applied to the C-phase winding 22C. From the above equations (eq11) to (eq13), the SRM model shown in FIG. 4 is determined. This motor model is a model in which an applied voltage of a winding is input and a current that is a controlled variable is output. In the model, A, B, C phase interference voltages VINA, VINB, VINC applied to the A, B, C phase windings 22A, 22B, 22C are shown. In the present embodiment, non-interference control is performed to remove the influence of the interference voltages VINA, VINB, and VINC.

  Next, current feedback control performed by the operation signal generation unit 32 will be described with reference to FIG.

  The command current generator 33 is a current command value that flows in the A, B, C phase windings 22A, 22B, 22C based on the torque command value T * and the electrical angle θ detected by the rotation angle sensor 40. A, B and C phase command currents Ia *, Ib * and Ic * are calculated.

  The A, B, C phase current deviation calculation units 34A, 34B, 34C are detected by the A, B, C phase command currents Ia *, Ib *, Ic * and the A, B, C phase current sensors 42A, 42B, 42C. Current deviations ΔIa, ΔIb, ΔIc from the calculated A, B, C phase currents Ia, Ib, Ic are calculated. Specifically, by subtracting A, B, C phase currents Ia, Ib, Ic from A, B, C phase command currents Ia *, Ib *, Ic *, A, B, C phase current deviations ΔIa, ΔIb, ΔIc is calculated.

  The A, B, and C phase current controllers 35A, 35B, and 35C receive the A, B, and C phase current deviations ΔIa, ΔIb, and ΔIc, and input the A, B, and C phase currents Ia, Ib, and Ic to A, B, As operation amounts for feedback control to the C-phase command currents Ia *, Ib *, and Ic *, A, B, and C-phase basic voltages Va1, Vb1, and Vc1 are calculated. In the present embodiment, the A, B, C phase basic voltages Va1, Vb1, Vc1 are calculated by proportional integral control based on the A, B, C phase current deviations ΔIa, ΔIb, ΔIc.

  In particular, in the present embodiment, the proportional gain Kpa and the integral gain Kia for calculating the A-phase basic voltage Va1 are calculated by the following equation (eq14).

上式（ｅｑ１４）において、ωｃｃは、制御帯域幅を定める交差角周波数［ｒａｄ／ｓ］を示す。ここで本実施形態では、比例ゲインＫｐａのＡ０を、以下に説明する手法によって算出する。本実施形態では、各相鎖交磁束λａ，λｂ，λｃが、Ａ，Ｂ，Ｃ相電流Ｉａ，Ｉｂ，Ｉｃと電気角θとを独立変数として数式化され、制御装置３０の備えるメモリに予め記憶されている。数式化された磁束は、例えば、実験又は磁界解析により得られるものである。このため、検出されたＡ，Ｂ，Ｃ相電流Ｉａ，Ｉｂ，Ｉｃ，電気角θで各相鎖交磁束λａ，λｂ，λｃを偏微分することにより、上式（ｅｑ５）〜（ｅｑ７）に示した各偏微分値を算出することができる。そして、算出した各偏微分値に基づいて、上式（ｅｑ８）に示したＤａ，Ａａを算出し、算出したＤａ，Ａａに基づいて、上式（ｅｑ１１）に示したＡ０を算出する。鎖交磁束λａ，λｂ，λｃを数式化して記憶させる構成によれば、例えば、メモリに記憶されるゲイン算出用の情報量を低減しつつ、偏微分演算により各偏微分値を簡易に算出することができる。

In the above equation (eq14), ωcc represents the cross angular frequency [rad / s] that defines the control bandwidth. Here, in this embodiment, A0 of the proportional gain Kpa is calculated by the method described below. In the present embodiment, the phase-linkage magnetic fluxes λa, λb, λc are expressed numerically by using the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ as independent variables, and stored in advance in a memory provided in the control device 30. It is remembered. The mathematically converted magnetic flux is obtained by experiment or magnetic field analysis, for example. Therefore, the above equations (eq5) to (eq7) are obtained by partially differentiating the phase linkage magnetic fluxes λa, λb, λc with the detected A, B, C phase currents Ia, Ib, Ic and the electrical angle θ. Each partial differential value shown can be calculated. Then, Da and Aa shown in the above equation (eq8) are calculated based on the calculated partial differential values, and A0 shown in the above equation (eq11) is calculated based on the calculated Da and Aa. According to the configuration in which the linkage magnetic fluxes λa, λb, and λc are mathematically stored, for example, each partial differential value is easily calculated by partial differential calculation while reducing the amount of information for gain calculation stored in the memory. be able to.

  A proportional gain Kpb and an integral gain Kib for calculating the B-phase basic voltage Vb1 are calculated by the following equation (eq15).

ここで比例ゲインＫｐｂのＢ０は、上記Ａ０の算出手法と同様な手法によって算出すればよい。詳しくは、各相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θに基づいて、上式（ｅｑ５）〜（ｅｑ７）に示した各偏微分値を算出し、算出した各偏微分値に基づいて、上式（ｅｑ９）に示したＤｂ，Ｂｂを算出する。そして、算出したＤｂ，Ｂｂに基づいて、上式（ｅｑ１２）に示したＢ０を算出する。

Here, B0 of the proportional gain Kpb may be calculated by a method similar to the method of calculating A0. Specifically, the partial differential values shown in the above equations (eq5) to (eq7) are calculated based on the phase currents Ia, Ib, Ic and the electrical angle θ, and based on the calculated partial differential values, Db and Bb shown in the equation (eq9) are calculated. Based on the calculated Db and Bb, B0 shown in the above equation (eq12) is calculated.

  A proportional gain Kpc and an integral gain Kic for calculating the C-phase basic voltage Vc1 are calculated by the following equation (eq16).

ここで比例ゲインＫｐｃのＣ０は、上記Ａ０の算出手法と同様な手法によって算出すればよい。詳しくは、各相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θに基づいて、上式（ｅｑ５）〜（ｅｑ７）に示した各偏微分値を算出し、算出した各偏微分値に基づいて、上式（ｅｑ１０）に示したＤｃ，Ｃｃを算出する。そして、算出したＤｃ，Ｃｃに基づいて、上式（ｅｑ１３）に示したＣ０を算出する。

Here, C0 of the proportional gain Kpc may be calculated by a method similar to the calculation method of A0. Specifically, the partial differential values shown in the above equations (eq5) to (eq7) are calculated based on the phase currents Ia, Ib, Ic and the electrical angle θ, and based on the calculated partial differential values, Dc and Cc shown in the equation (eq10) are calculated. Then, based on the calculated Dc and Cc, C0 shown in the above equation (eq13) is calculated.

  Here, in this embodiment, each proportional gain Kpa, Kpb, Kpc and each integral gain Kia, Kib, Kic are set by pole-zero cancellation. Hereinafter, the phase A will be described as an example with reference to FIG. FIG. 5 shows a block diagram when the A-phase interference voltage VINA is completely removed by non-interference control.

  When the gains Kpa and Kia for canceling out the denominator pole of the transfer function of the SRM model shown in FIG. 5A and the zero of the numerator of the transfer function of the A-phase current controller 35A are calculated, the above equation (eq14) is obtained. The indicated gain is derived. In this embodiment, the poles of the SRM model are stable. When the gains Kpa and Kia of the above equation (eq14) are substituted into the transfer function of the A-phase current controller 35A and the transfer function of the A-phase current controller 35A and the transfer function of the SRM model are summarized, FIG. The block diagram shown in FIG. Since the transfer function F shown in FIG. 5B is an integral type, the steady-state deviation can be set to zero. In the block diagram shown in FIG. 5B, the closed-loop transfer function having the A-phase command current ia * as input and the A-phase current ia as output is a first-order lag system.

  Incidentally, in the present embodiment, each of the current controllers 35A to 35C corresponds to “command voltage calculation means” and “gain calculation means”.

  Returning to the description of FIG. 4 above, the A, B, C phase interference voltage calculation units 36A, 36B, 36C (corresponding to “interference voltage calculation means”) are used for the respective phase currents Ia, Ib, Ic, electrical angle θ, Using the angular velocity ω as an input, A, B, C phase interference voltages VINA, VINB, VINC are calculated based on the equation of each phase interference voltage shown in the above equations (eq11) to (eq13). More specifically, the phase A will be described as an example. First, the phase-linkage magnetic fluxes λa, λb, and λc expressed by the A, B, and C phase currents Ia, Ib, and Ic and the electrical angle θ are partially differentiated. Thus, the partial differential values shown in the above equations (eq5) to (eq7) can be calculated. Then, Aa, Ab, Ac, and Aw shown in the above equation (eq8) are calculated based on the calculated partial differential values. Then, based on the calculated Aa, Ab, Ac, Aw, B, C-phase currents Ib, Ic, electrical angular velocity ω, and B, C-phase voltages Vb, Vc, the A-phase interference voltage shown in the above equation (eq11) VINA is calculated. Here, as the B and C phase voltages Vb and Vc, for example, the B and C phase command voltages Va *, Vb *, and Vc * calculated in the previous processing cycle may be used. According to the configuration in which the linkage fluxes λa, λb, and λc are mathematically stored, for example, each partial differential value is easily calculated by partial differential calculation while reducing the amount of information for calculating the interference voltage stored in the memory. can do.

  The A, B, C phase non-interacting units 37A, 37B, 37C (corresponding to "voltage compensation means") are converted from the A, B, C phase basic voltages Va1, Vb1, Vc1 to the A, B, C phase interference voltage VINA. , VINB, VINC are subtracted and output. Each output value becomes A, B, C phase command voltage Va *, Vb *, Vc *. The operation signal generation unit 32 applies each of the switches SAp, SAn, SBp, SBn, SCp, SCn to apply the command voltages Va *, Vb *, Vc * to the A, B, C phase windings 22A, 22B, 22C. Operation signals gAp, gAn, gBp, gBn, gCp, and gCn are generated. Here, for example, the command voltage is set by combining a positive voltage mode in which both the upper and lower arm switches are turned on, a negative voltage application mode in which both are turned off, and a zero voltage mode in which either the upper or lower arm switch is turned on. What is necessary is just to apply.

  In the present embodiment, the flux linkage information λa, λb, each of the flux linkage information λa, λb, which is expressed by the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ as independent variables is stored in the memory. Although it is set as λc, it is not limited to this. For example, each partial differential value LAt, MABt, MACt, dλat,..., LCt, dλct shown in the above equations (eq5) to (eq7) is related to each phase current Ia, Ib, Ic and electrical angle θ. May be magnetic flux information. Then, an interference voltage and a feedback gain (for example, the proportional gain) may be calculated based on each stored partial differential value.

  Subsequently, the effects of the present embodiment will be described with reference to FIGS. Here, the related technique is a technique for calculating the interference voltage and the proportional gains as shown in the following equation (eq17) with respect to the current dependency of each self-inductance La, Lb, Lc and each mutual inductance Mij.

図６及び図７に、関連技術にかかる各相電流Ｉａ，Ｉｂ，Ｉｃ及び各相指令電流Ｉａ＊，Ｉｂ＊，Ｉｃ＊の推移を示す。ここで図５における各指令電流（例えば１００Ａ）は、図６における各指令電流（例えば１５０Ａ）よりも小さく設定されている。

6 and 7 show transitions of the phase currents Ia, Ib, Ic and the phase command currents Ia *, Ib *, Ic * according to the related art. Here, each command current (for example, 100 A) in FIG. 5 is set smaller than each command current (for example, 150 A) in FIG.

  As shown in FIG. 6, in a period in which current flows through at least two windings, current controllability and control stability are reduced due to the influence of mutual magnetic interference. Here, as shown in FIG. 7, when the command current is further increased, the influence of mutual magnetic interference is increased and current control is broken.

  Next, FIGS. 8 and 9 show transitions of the phase currents Ia, Ib, Ic and the phase command currents Ia *, Ib *, Ic * according to the present embodiment. Here, each command current (for example, 100 A) in FIG. 8 is set smaller than each command current (for example, 150 A) in FIG. 9.

  In the present embodiment, unlike the above equation (eq17), all the inductances La, Lb, Lc, Mab, Mac, Mba, Mbc, Mca, Mcb are all converted to A, B, C phase currents Ia, Ib, Ic. Expressed as a function of the electrical angle θ, each interference voltage VINA, VINB, VINC and each proportional gain Kpa, Kpb, Kpc were calculated. For this reason, as shown in FIG. 9, even when the command current increases, each phase current can follow the command current without breaking the current control.

  Incidentally, in the region where the electrical angular velocity ω is high, the interference voltage of each phase is high, so that the influence of the interference voltage on the current control becomes large. Even in this case, according to the present embodiment, the influence of the interference voltage on the current control can be suppressed.

  As described above, in the present embodiment, all of the A, B, C phase windings 22A, 22B, 22C are all based on the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ. The A, B, and C phase interference voltages VINA, VINB, and VINC depending on the phase current were calculated. The A, B, C phase basic voltages Va1, Vb1, Vc1 calculated by the A, B, C phase current controllers 35A, 35B, 35C are compensated by the A, B, C phase interference voltages VINA, VINB, VINC. A voltage was applied to the A, B, C phase windings 22A, 22B, 22C.

  The proportional gains Kpa, Kpb, Kpc used in the A, B, C phase current controllers 35A, 35B, 35C were calculated based on the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ.

  Thereby, it can be avoided that the A, B, C phase currents Ia, Ib, Ic greatly deviate from the A, B, C phase command currents Ia *, Ib *, Ic *. Therefore, the current controllability and control stability of the SRM can be improved.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In this embodiment, instead of the current feedback control, magnetic flux feedback control is performed in which the control amount of the SRM is the linkage flux.

  First, a control system design method will be described.

  The above formula (eq2) is represented as the following formula (eq18).

上式（ｅｑ１８）をＡ，Ｂ，Ｃ相電流Ｉａ，Ｉｂ，Ｉｃについて解くと、下式（ｅｑ１９）が導かれる。

When the above equation (eq18) is solved for the A, B, and C phase currents Ia, Ib, and Ic, the following equation (eq19) is derived.

上式（ｅｑ１９）を上式（ｅｑ１）に代入し、各鎖交磁束について解くと、下式（ｅｑ２０）〜（ｅｑ２２）が導かれる。なお、以降、ｄｅｔ［Ａ］＝Ｄｅｔとする。

By substituting the above equation (eq19) into the above equation (eq1) and solving for each flux linkage, the following equations (eq20) to (eq22) are derived. Hereinafter, it is assumed that det [A] = Det.

上式（ｅｑ２０）〜（ｅｑ２２）において、ＶｉＡ，ＶｉＢ，ＶｉＣは、本実施形態にかかるＡ，Ｂ，Ｃ相干渉電圧を示す。上式（ｅｑ２０）〜（ｅｑ２２）から、図１０に示すＳＲＭのモデルが定まる。このモータモデルは、巻線の印加電圧を入力とし、制御量である磁束を出力とするモデルである。

In the above equations (eq20) to (eq22), ViA, ViB, and ViC indicate the A, B, and C phase interference voltages according to the present embodiment. From the above equations (eq20) to (eq22), the SRM model shown in FIG. 10 is determined. This motor model is a model in which an applied voltage of a winding is input and a magnetic flux as a control amount is output.

  Next, magnetic flux feedback control performed by the operation signal generation unit 32 will be described with reference to FIG.

  The command current generator 50 is based on the torque command value T * and the electrical angle θ, and is the A, B, C phase that is the command value of the interlinkage magnetic flux of the A, B, C phase windings 22A, 22B, 22C. Command magnetic fluxes λa *, λb *, and λc * are calculated.

  The A, B, and C phase magnetic flux deviation calculation units 51A, 51B, and 51C are used for the A, B, and C phase command magnetic fluxes λa *, λb *, and λc * and the current A, B, and C phase windings 22A, 22B, and 22C. Magnetic flux deviations Δλa, Δλb, and Δλc with respect to the A, B, and C phase interlinkage magnetic fluxes λa, λb, and λc, which are the interlinkage magnetic fluxes, are calculated. Here, the A, B, C phase interlinkage magnetic fluxes λa, λb, λc are calculated from the interlinkage magnetic flux expressed based on the A, B, C phase currents Ia, Ib, Ic and the electrical angle θ. do it.

  The A, B, and C phase magnetic flux controllers 52A, 52B, and 52C receive the A, B, and C phase magnetic flux deviations Δλa, Δλb, and Δλc, and input the A, B, and C phase flux linkages λa, λb, and λc to A, As operation amounts for feedback control to the B and C phase command magnetic fluxes λa *, λb *, and λc *, A, B, and C phase basic voltages Va1, Vb1, and Vc1 are calculated. In the present embodiment, the A, B, C phase basic voltages Va1, Vb1, Vc1 are calculated by proportional integral control based on the A, B, C phase current deviations ΔIa, ΔIb, ΔIc.

  In particular, in this embodiment, the proportional gain Kpad and the integral gain Kiad for calculating the A-phase basic voltage Va1 are calculated by the following equation (eq23).

ここで積分ゲインＫｉａｄのＡｄ０は、以下に説明する手法によって算出すればよい。詳しくは、本実施形態では、各自己インダクタンスＬａ，Ｌｂ，Ｌｃと、各相互インダクタンスＭａｂ，Ｍａｃ，Ｍｂａ，Ｍｂｃ，Ｍｃａ，Ｍｃｂとが、Ａ，Ｂ，Ｃ相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θと関係付けられてメモリに予め記憶されている。このため、各相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θに基づいて、上式（ｅｑ２０）に示したＡｄ０を算出すればよい。

Here, Ad0 of the integral gain Kiad may be calculated by the method described below. Specifically, in the present embodiment, each self-inductance La, Lb, Lc and each mutual inductance Mab, Mac, Mba, Mbc, Mca, Mcb are represented by A, B, C-phase currents Ia, Ib, Ic and electrical angles. It is related to θ and stored in advance in the memory. For this reason, Ad0 shown in the above equation (eq20) may be calculated based on the phase currents Ia, Ib, Ic and the electrical angle θ.

  The proportional gain Kpbd and integral gain Kibd for calculating the B-phase basic voltage Vb1 are calculated by the following equation (eq24).

ここで積分ゲインＫｉｂｄのＢｄ０は、上式（ｅｑ２１）に示すように、各相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θに基づいて算出すればよい。

Here, Bd0 of the integral gain Kibd may be calculated based on the phase currents Ia, Ib, Ic and the electrical angle θ as shown in the above equation (eq21).

  A proportional gain Kpcd and an integral gain Kicd for calculating the C-phase basic voltage Vc1 are calculated by the following equation (eq25).

ここで積分ゲインＫｉｃｄのＢｄ０は、上式（ｅｑ２２）に示すように、各相電流Ｉａ，Ｉｂ，Ｉｃ及び電気角θに基づいて算出すればよい。なお、各比例ゲインＫｐａｄ，Ｋｐｂｄ，Ｋｐｃｄ及び各積分ゲインＫｉａｄ，Ｋｉｂｄ，Ｋｉｃｄは、上記第１実施形態と同様に、極零相殺によって定めたものである。

Here, Bd0 of the integral gain Kidd may be calculated based on the phase currents Ia, Ib, Ic and the electrical angle θ as shown in the above equation (eq22). Each proportional gain Kpad, Kpbd, Kpcd and each integral gain Kiad, Kibd, Kidd are determined by pole-zero cancellation, as in the first embodiment.

  The A, B, and C phase interference voltage calculation units 53A, 53B, and 53C receive the phase currents Ia, Ib, and Ic and the electrical angle θ, and input A, B, and C shown in the above equations (eq20) to (eq22). Phase interference voltages ViA, ViB and ViC are calculated.

  The A, B, C phase non-interacting parts 54A, 54B, 54C subtract the A, B, C phase interference voltages ViA, ViB, ViC from the A, B, C phase basic voltages Va1, Vb1, Vc1 and output them. To do. Each output value becomes A, B, C phase command voltage Va *, Vb *, Vc *. The operation signal generation unit 32 applies each of the switches SAp, SAn, SBp, SBn, SCp, SCn to apply the command voltages Va *, Vb *, Vc * to the A, B, C phase windings 22A, 22B, 22C. Operation signals gAp, gAn, gBp, gBn, gCp, and gCn are generated.

  According to the present embodiment described above, it is possible to obtain the same effect as that of the first embodiment.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, the method for calculating the interference voltage is changed in a region where the electrical angular velocity ω is high. Hereinafter, the phase A will be described as an example.

  The interference voltage shown in the above equation (eq11) can be transformed as in the following equation (eq26).

上式（ｅｑ２６）の右辺において、第１項は巻線抵抗Ｒに比例する抵抗項ＲＦを示し、第２項は電気角速度ωに比例する速度項ωＦを示し、第３項は巻線印加電圧に比例する電圧項を示す。ここで図１１に、電気角速度ωが高い場合（５０００ｒｐｍを例示）と低い場合（１００ｒｐｍを例示）との電気角１周期における抵抗項ＲＦと速度項ωＦとを示す。なお、図１１において、電気角速度が高い場合の抵抗項ＲＦ，速度項ωＦのグラフの縦軸１メモリのスケールは、電気角速度が低い場合の抵抗項ＲＦ，速度項ωＦのグラフの縦軸１メモリのスケールよりも大きい。

In the right side of the above equation (eq26), the first term represents the resistance term RF proportional to the winding resistance R, the second term represents the speed term ωF proportional to the electrical angular velocity ω, and the third term represents the winding applied voltage. The voltage term proportional to is shown. Here, FIG. 11 shows a resistance term RF and a velocity term ωF in one cycle of the electrical angle when the electrical angular velocity ω is high (example of 5000 rpm) and low (example 100 rpm). In FIG. 11, the scale of the vertical axis 1 memory of the graph of the resistance term RF and the velocity term ωF when the electrical angular velocity is high is the vertical axis 1 memory of the graph of the resistance term RF and the velocity term ωF when the electrical angular velocity is low. Greater than the scale.

  As shown in the figure, when the electrical angular velocity is low, the resistance term RF and the velocity term ωF have similar values. On the other hand, when the electrical angular velocity is high, the resistance term RF is sufficiently smaller than the velocity term ωF. This means that the influence of the resistance term RF on the current controllability is small when the electrical angular velocity is high. For this reason, when the electrical angular velocity is high, the resistance term RF can be ignored.

  FIG. 12 shows a procedure of interference voltage calculation processing according to the present embodiment. This process is repeatedly executed, for example, at a predetermined processing cycle by each of the interference voltage calculation units 36A to 36C. In FIG. 12, the A phase is described as an example, but this process is the same for the B and C phases.

  In this series of processes, first, in step S10, it is determined whether or not the electrical angular velocity ω exceeds the threshold velocity ωth. This process is a process for determining whether or not the resistance term RF can be ignored.

  When a negative determination is made in step S10, the process proceeds to step S11, and the interference voltage VINA is calculated as an added value of the resistance term RF, the speed term ωF, and the voltage term VF.

  On the other hand, if an affirmative determination is made in step S10, it is determined that the resistance term RF can be ignored, and the process proceeds to step S12. In step S12, the interference voltage VINA is calculated as an addition value of the speed term ωF and the voltage term VF without calculating the resistance term RF.

  Thus, in this embodiment, when the electrical angular velocity ω is higher than the threshold velocity ωth, the interference voltages VINA, VINB, and VINC are calculated without calculating the resistance term RF. Thereby, the calculation load of the control apparatus 30 can be reduced.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the first embodiment, the command voltages Va *, Vb *, and Vc * of the previous processing cycle are used for calculating the interference voltage and the feedback gain, but the present invention is not limited to this. For example, the control system may be provided with voltage detection means for detecting the applied voltage of each phase winding 22A, 22B, 22C, and the detection value of the voltage detection means may be used instead of the command voltage.

  In each of the above embodiments, instead of the configuration for calculating the basic voltage by proportional-integral control, for example, a configuration for calculating the basic voltage by proportional-integral-derivative control may be adopted.

  In each of the above-described embodiments, the control device 30 controls the three-phase full-volume SRM. However, the control device 30 may control all-volume SRM other than the three-phase. In this case, if the full volume SRM is the N phase, the command current of each phase may be shifted by “360 ° / X”.

  Further, the SRM is not limited to a full-pitch concentrated winding, but may be a single-pump concentrated winding. In this case, each phase winding may be energized as shown in FIG. Even in this configuration, there is a period in which the two windings are energized, and therefore the application of the present invention is effective.

  As the SRM control amount, for example, the rotational speed of the rotor may be used instead of the torque.

  -The application target of the present invention is not limited to a motor as an in-vehicle main machine, but may be a motor as an in-vehicle auxiliary machine, for example. Further, the application target of the present invention is not limited to the in-vehicle motor.

  22A, 22B, 22C ... A, B, C phase winding, 20 ... power conversion circuit, 30 ... control device.

Claims (11)

A switched reluctance motor having a plurality of windings (22A, 22B, 22C) magnetically coupled to each other;
Voltage application means (20) for applying a voltage to each of the plurality of windings, and is applied to a motor control system,
A current flowing through the winding or a magnetic flux interlinking with the winding is used as a control amount, and the control amount of each of the plurality of windings is applied to each of the plurality of windings in order to control the command value. Command voltage calculating means (35A to 35C, 52A to 52C) for calculating the command voltage;
Operating means for operating the voltage applying means to apply a voltage to each of the plurality of windings based on the command voltage calculated by the command voltage calculating means,
Among the plurality of windings, one winding to be calculated for the command voltage is a self-phase winding, and the remaining winding is another phase winding,
The command voltage calculation means is configured such that each of the self-inductance of the self-phase winding and the mutual inductance between the self-phase winding and the other-phase winding flows in each of the plurality of windings. A control device for a switched reluctance motor, wherein the command voltage is calculated based on a model of the dependent motor. The motor control system includes current detection means (42A, 42B, 42C) for detecting a current flowing in each of the plurality of windings,
For each of the plurality of windings, an interference voltage including the self-inductance and the mutual inductance based on a current flowing through each of the plurality of windings detected by the current detection unit, the self-phase Interference voltage calculating means (36A to 36C; 53A to 53C) for calculating an interference voltage generated in the self-phase winding by magnetic coupling between the winding and the other-phase winding;
Voltage compensation means (37A to 37C; 54A to 54C) for compensating the command voltage calculated by the command voltage calculation means for each of the plurality of windings with the interference voltage calculated by the interference voltage calculation means. And further comprising
The control device for a switched reluctance motor according to claim 1, wherein the operating means operates the voltage applying means so as to apply the command voltage compensated by the voltage compensating means to each of the plurality of windings. Information on the interlinkage magnetic flux of the winding associated with the current flowing through each of the plurality of windings, or the interlinkage magnetic flux of the winding associated with the current flowing through each of the plurality of windings. Further comprising storage means for preliminarily storing magnetic flux information including information related to a value obtained by partial differentiation of the current flowing through the winding;
The switched reluctance motor control device according to claim 2, wherein the interference voltage calculation unit calculates the interference voltage based on the magnetic flux information stored in the storage unit. The magnetic flux information includes information on the interlinkage magnetic flux expressed as an independent variable using a current flowing through each of the plurality of windings and an electric angle of the motor.
The interference voltage calculation means performs a partial differentiation on the mathematical interlinkage magnetic flux with a current flowing through the winding, and a partial differentiation on the electrical flux based on the electrical angle. 4. The switched reluctance motor control apparatus according to claim 3, wherein the interference voltage is calculated based on each value. The controlled variable is a current flowing through the winding,
The interference voltage includes a term whose absolute value is proportional to the resistance of the winding, and a term whose absolute value is proportional to the electrical angular velocity of the motor,
5. The interference voltage calculation unit calculates the interference voltage by excluding a term proportional to the resistance of the winding on condition that the electrical angular velocity exceeds a predetermined angular velocity. Switched reluctance motor control device according to the item. The motor control system includes current detection means (42A, 42B, 42C) for detecting a current flowing in each of the plurality of windings,
The command voltage calculation means constitutes a controller that calculates the command voltage as an operation amount for feedback control of the control amount to the command value for each of the plurality of windings,
A gain for calculating a feedback gain including the self-inductance and the mutual inductance based on a current flowing through each of the plurality of windings detected by the current detection unit, the feedback gain being used in the feedback control. The control device for a switched reluctance motor according to any one of claims 1 to 5, further comprising calculation means (35A to 35C; 52A to 52C). In the model, an applied voltage of the winding is input, and the control amount is output.
The switched reluctance motor control according to claim 6, wherein the feedback gain is set as a value that cancels out the pole of the model expressed by a transfer function and a zero point of the controller expressed by the transfer function. apparatus. Information on the interlinkage magnetic flux of the winding associated with the current flowing through each of the plurality of windings, or the interlinkage magnetic flux of the winding associated with the current flowing through each of the plurality of windings. Further comprising storage means for preliminarily storing magnetic flux information including information related to a value obtained by partial differentiation of the current flowing through the winding;
The switched reluctance motor control device according to claim 6 or 7, wherein the gain calculating means calculates the feedback gain based on the magnetic flux information stored in the storage means. The magnetic flux information includes information on the interlinkage magnetic flux expressed as an independent variable using a current flowing through each of the plurality of windings and an electric angle of the motor.
The gain calculating means performs partial differentiation of the mathematically calculated flux linkage with a current flowing through the winding, and partial differentiation of the mathematically calculated flux linkage with the electrical angle. 9. The switched reluctance motor control apparatus according to claim 8, wherein the feedback gain is calculated based on a value.   The operation means is configured to periodically energize each of the plurality of windings, and to apply a part of the energization period of at least two of the plurality of windings to overlap each other. The control apparatus of the switched reluctance motor according to any one of claims 1 to 9. The motor is three-phase,
11. The switched reluctance motor control device according to claim 10, wherein the operating means operates the voltage applying means such that a part of energization periods of the plurality of windings overlap each other.

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