JP5469897B2 - AC motor control device and AC motor drive system - Google Patents

Description

  The present invention relates to an AC motor control device and an AC motor drive system, and more particularly to an AC motor control device and an AC motor drive system suitable for a permanent magnet synchronous motor.

  AC motors, in particular permanent magnet synchronous motors (hereinafter referred to as PM motors), are expanding their application in home appliances, industries, automobiles, etc., taking advantage of their small size and high efficiency. In particular, in recent years, as a result of further miniaturization, motors have appeared in which the saturation characteristics of the magnetic circuit constituting the motor are remarkable. In such a motor, since the inductance that has been treated as a constant in the prior art varies greatly depending on the current, the current control accuracy deteriorates.

  For such a problem, Patent Document 1 discloses a technique for changing an electric constant setting value of an AC motor in accordance with an electric current. This technique is a technique for improving the torque accuracy by providing a current command generation unit with a relationship between magnetic flux and current of a synchronous motor as a nonlinear function (hereinafter, referred to as conventional technique 1).

  Further, in Patent Document 2, the control gain used for the current feedback control is set to a value proportional to the fluctuation rate Δφ / Δi of the own-axis linkage magnetic flux number fluctuation Δφ with respect to the own-axis current fluctuation Δi. The accuracy is improved (hereinafter referred to as Conventional Technology 2).

JP 2001-161099 A JP 2008-141835 A

  According to the prior art 1, an improvement in torque accuracy can be expected by calculating a current command value in consideration of magnetic flux saturation characteristics. However, no means for improving the transient response of the current control system is specified.

  On the other hand, in the prior art 2, the transient gain of the current control system is improved by setting the control gain used for the current feedback control to a function that takes into account the saturation characteristics of the magnetic flux or a variable value based on a table. However, a feedback control system that takes into account even the interference effect of the d-axis and q-axis magnetic fluxes has not been constructed.

  The present invention has been made in consideration of the above points, and a current control transient response as set can be obtained even for a PM motor in which magnetic flux saturation is remarkable and there is a large amount of mutual interference magnetic flux between shafts. An object is to provide an AC motor control device and an AC motor drive system.

  In order to solve the above-described problems, the present invention configures a feedback control system that takes into account the mutual interference effect of magnetic flux between the d-axis and the q-axis. In the present invention, a non-linear function of magnetic flux and current is prepared, and using these, the magnetic flux deviation in each voltage command and the calculation corresponding to the magnetic flux in each axis are executed. A current control transient response as set is realized even for a PM motor having a large amount of mutual interference magnetic flux between them. In particular, the voltage command value to the power converter defined on one of the dq control axes orthogonal to each other in the rotational coordinate system of the motor is the current command value and current detection value defined on the same axis. And a deviation between the current command value defined on the other axis and the detected current value.

  Specifically, in order to achieve the above object, the present invention relates to an AC motor control device that controls the output voltage of a power converter that drives an AC motor, and the control device is orthogonal to the rotational coordinate system of the motor. Among the two control axes, the voltage command value to the power converter defined on one axis is the difference between the current command value defined on the same axis and the current detection value, and the other axis. It is generated using information on both of the deviation between the current command value and the current detection value defined in (1).

  Further, according to the present invention, in the AC motor control device, the voltage command value to the power converter defined on the one axis is converted into a deviation between the current command value defined on the same axis and the current detection value. On the other hand, a value obtained by multiplying a current detection value defined on the same axis and a function having at least one of the current detection value defined on the other axis as a variable and the current detection value defined on the other axis. A function having at least one of the current detection value defined on the same axis and the current detection value defined on the other axis as a variable with respect to the deviation between the current command value and the current detection value It produces | generates based on the sum with the value which multiplied by.

  Further, according to the present invention, in the AC motor control device, the voltage command value to the power converter defined on the one axis is converted into a deviation between the current command value defined on the same axis and the current detection value. On the other hand, the current detection value defined on the same axis and the value obtained by multiplying the table data referring to at least one of the current detection value defined on the other axis, and the definition on the other axis Table data referring to at least one of the current detection value defined on the same axis and the current detection value defined on the other axis with respect to the deviation between the current command value and the current detection value It produces | generates based on the sum with the value which multiplied by.

  Furthermore, the present invention provides an AC motor control device, wherein the function is a coil interlinkage flux function of the motor defined on one of two control axes orthogonal to each other in the rotational coordinate system of the motor. , A function that is partially differentiated by a current defined on the same axis or the other axis.

  Furthermore, the present invention provides an AC motor control apparatus, wherein the table data is a coil linkage flux function of the motor defined on one of two control axes orthogonal to each other in the rotational coordinate system of the motor. Alternatively, it is a function or table data obtained by partial differentiation of the table data of the coil linkage magnetic flux function with a current defined on the same axis or the other axis.

Furthermore, in the control apparatus for an AC motor according to the present invention, when the two control axes orthogonal to each other in the rotational coordinate system of the motor are d-axis and q-axis, respectively, the current command value defined on the d-axis is expressed as i _d ^*, defines the current detection value which is defined on the d-axis i _d, the i _d ^* and i _d and deviation .DELTA.i _d of the current command value defined on the q-axis i _q ^*, on the q-axis The detected current value is i _q , the deviation between i _q ^* and i _q is Δi _q , the coil linkage magnetic flux function defined on the d axis is Φ _d ( _id , i _q ), and Φ _d (i _d, i _q) of the current i _d ∂Φ a partially differentiating the function with _d (i _d, i _q) / ∂i _d, polarized wherein [Phi _d a (i _d, i _q) in the current i _q When the differentiated function is ∂Φ _d (i _d , i _q ) / qi _q , the voltage command value v _d ^* to the power converter defined on the d axis is calculated from (Equation 1). defined on the d-axis Is characterized in that generated using serial coil flux linkage function _{Φ d (i d, i q} ) of the current i _d, i _q on the change amount _{ΔΦ d (i d, i q} ).

Furthermore, in the control apparatus for an AC motor according to the present invention, when the two control axes orthogonal to each other in the rotational coordinate system of the motor are d-axis and q-axis, respectively, the current command value defined on the d-axis is expressed as i _d ^*, defines the current detection value which is defined on the d-axis i _d, the i _d ^* and i _d and deviation .DELTA.i _d of the current command value defined on the q-axis i _q ^*, on the q-axis The detected current value is i _q , the deviation between i _q ^* and i _q is Δi _q , the coil linkage magnetic flux function defined on the q axis is Φ _q ( _id , i _q ), and Φ _q (i _d, i _q) polarized the current i ∂Fai a partially differentiating the function with _{d q (i d, i q} ) / ∂i d, wherein [Phi _q (i _d, i _q) with the current i _q When the differentiated function is ∂Φ _q (i _d , i _q ) / ∂i _q , the voltage command value v _q ^* to the power converter defined on the q axis is calculated from (Equation 2). defined on the q-axis Is characterized in that generated using serial coil flux linkage function _{Φ q (i d, i q} ) of the current i _d, i _q on the change amount _{ΔΦ q (i d, i q} ).

Further, in the present invention is a controller for an AC motor, said defined on the d-axis coil flux linkage function _{Φ d (i d, i q} ) as a current defined on the d-axis i _d, q-axis When the current defined above is i _q , the permanent magnet magnetic flux is φ _m , and the constants are k ₁ , k ₂ , k ₃ , k ₄ , K ₁₀ , and I _m0 , if i _d > 0 (Equation 3) If i _d <0, (Equation 4) is used.

Further, in the present invention is a controller for an AC motor, said defined over q-axis coil flux linkage function [Phi _q (i _d, i _q) as a current defined on the d-axis i _d, q-axis It is characterized by using (Formula 5) when the current defined above is i _q and the constants are k ₅ , k ₆ , and K ₂₀ .

Furthermore, in the control apparatus for an AC motor according to the present invention, when k ₁ , k ₂ , k ₃ , k ₄ , K ₁₀ , and I _m0 are constants, the coil linkage magnetic flux function Φ _d defined on the d-axis is used. (i _d, i _q) the function obtained by partially differentiating the current i _d a ∂Fai _d (i _d, i _q) as / ∂i _d, i _d> 0 if the (expression 6), i _d <0 If you are using (equation 7), wherein the coil flux linkage function _{Φ d (i d, i q} ) the function obtained by partially differentiating the current _{i q ∂Φ d (i d,} i q) as / ∂i _q , (Equation 8) is used.

Furthermore, the present invention is the control apparatus for an AC motor, when the k _5, k _6, K ₂₀ constant, the defined over q-axis coil flux linkage function _{Φ q (i d, i q} ) the the partially differentiated by the current i _d function _{∂Φ q (i d, i q} ) / ∂i using (equation 9) as _d, the coil flux linkage function [Phi _q (i _d, i _q) the current i the partially differentiated with _q function _{∂Φ q (i d, i q} ) as / ∂i _q, is characterized in the use of (equation 10).

Further, in the present invention is a controller for an AC motor, said defined on the d-axis coil flux linkage function _{Φ d (i d, i q} ) as a current defined on the d-axis i _d, q-axis When the current defined above is i _q and the constants are K ₁ , K ₂ , K ₃ , φ ₀ , I ₀ , (Equation 11) is used.

Further, in the present invention is a controller for an AC motor, said defined over q-axis coil flux linkage function [Phi _q (i _d, i _q) as a current defined on the d-axis i _d, q-axis It is characterized by using (formula 12) when the current defined above is i _q and the constants are K ₄ , K ₅ , K ₆ , and I ₁ .

Further, in the present invention is a controller for an AC _{motor, K 1, K 2, K} 3, when the a I ₀ constant, the coil flux linkage function defined on the d-axis Φ _d (i _d, i _q ) the function obtained by partially differentiating the current i _d ∂Φ _d (i _d, as i _q) / ∂i _d using (expression 13), the coil flux linkage function [Phi _d (i _d, a i _q) As the function ∂Φ _d (i _d , i _q ) / qi _q partially differentiated by the current i _q , if i _q > 0, use (Expression 14), and if i _q <0, (Expression 15) It is characterized by using.

Further, in the present invention is a controller for an AC _{motor, K 4, K 5, K} 6, when the a I ₁ constant, the coil flux linkage functions defined on q-axis Φ _q (i _d, i _q the function ∂Φ _q (i _d which) was partially differentiated by the current _{i d, i q) / ∂i} d as i _d + I _1> 0 If the equation (16), if i _d + I ₁ <0 ( using equation 17), the coil flux linkage function Φ _q (i _d, the function i _q) obtained by partially differentiating the current i _q ∂Φ _q (i _d, as i _q) / ∂i _q, (formula 18) is used.

  Furthermore, the present invention provides a control device for an AC motor, wherein when the motor does not have magnetic flux interference between the control shafts and the motor is in a stopped state, a current detection value defined on one shaft is obtained. When this is changed, the current detection value defined on the other axis changes.

  Furthermore, the present invention provides an AC motor control device in which the motor is replaced with a series circuit of three independent resistors and inductances, and the current detection defined on one axis when the motor is stopped. When the value is changed, the current detection value defined on the other axis changes.

  In order to solve the above problem, the present invention applies a pulse-width modulated voltage to an AC motor, drives the AC motor, means for detecting the current of the AC motor, and And a controller for driving the AC motor by adjusting an output voltage output from the power converter. The controller includes two control axes orthogonal to each other in the rotational coordinate system of the motor. The voltage command value for the power converter defined on one axis is defined as the deviation between the current command value defined on the same axis and the current detection value, and on the other axis. The information is generated using information on both the deviation between the current command value and the current detection value.

  In order to solve the above problems, the present invention provides an AC motor, an inverter for driving the AC motor by applying a pulse-width modulated voltage to the AC motor, and a means for detecting the current of the AC motor. And an AC motor drive system that adjusts an output voltage output from the inverter and drives the AC motor, wherein the AC motor drive system includes two control axes that are orthogonal to each other in the rotational coordinate system of the motor. The voltage command value to the power converter defined on one axis is defined as the deviation between the current command value defined on the same axis and the current detection value, and the other axis. It is generated using information on both the deviation between the current command value and the detected current value.

  Furthermore, the present invention is characterized in that the vehicle drive device includes the above-described AC motor control device or AC motor drive system.

  As described above, in the present invention, magnetic flux and current non-linear functions are prepared, and using them, the magnetic flux deviations in the voltage command generation unit and the operations corresponding to the magnetic fluxes of each axis are executed, whereby the magnetic flux Even for PM motors where saturation is significant and there are many inter-interference magnetic fluxes between the axes, the current control transient response as set is realized. In particular, the voltage command value to the power converter defined on one of the dq control axes orthogonal to each other in the rotational coordinate system of the motor is the current command value and current detection value defined on the same axis. And a difference between the current command value defined on the other axis and the detected current value are characteristic features of the solution means.

  According to the present invention, a current control transient response as set can be obtained even for a motor in which magnetic flux saturation is remarkable and there is a large amount of inter-axis interference magnetic flux. As a result, effects such as improved control system stability and reduced overshoot can be obtained as compared with the prior art.

The voltage command production | generation part 1 in Embodiment 1, 2 of this invention. Composition of Chapter 4 (Prior Art 3) written by Hidehiko Sugimoto, “Theory and Design of AC Servo Systems” (general electronic publisher). The schematic diagram of the relationship between the magnetic flux and current of an ideal PM motor. The schematic diagram of the relationship between the magnetic flux and current of a non-linear PM motor. The relationship between current and magnetic flux according to (Equation 23), (Equation 24) and (Equation 25). Relationship between current and magnetic flux according to (Equation 26) and (Equation 27). The block diagram which shows the whole structure (torque control system) of Embodiment 1,2,3,4,5,6 of this invention. The d-axis magnetic flux deviation calculation part based on (Formula 23), (Formula 24), and (Formula 28) in Embodiment 1,3. The q-axis magnetic flux deviation calculation part based on (Formula 25) and (Formula 29) in Embodiment 1,3. The d-axis magnetic flux calculation part based on (Formula 23) and (Formula 24) in Embodiment 1. FIG. The q-axis magnetic flux calculating part based on (Formula 25) in Embodiment 1. FIG. The d-axis magnetic flux deviation calculation part based on (Formula 26) and (Formula 28) in Embodiment 2,4. The q-axis magnetic flux deviation calculation part based on (Formula 27) and (Formula 29) in Embodiment 2,4. The d-axis magnetic flux calculating part based on (Formula 26) in Embodiment 2. FIG. The q-axis magnetic flux calculating part based on (Formula 27) in Embodiment 2. FIG. The voltage command production | generation part 200 in Embodiment 3, 4 of this invention. The d-axis simulation magnetic flux calculating part based on (Formula 41) and (Formula 42) in Embodiment 3. FIG. The q-axis simulation magnetic flux calculating part based on (Formula 43) in Embodiment 3. FIG. The d-axis simulation magnetic flux calculating part based on (Formula 44) in Embodiment 4. FIG. The q-axis simulation magnetic flux calculating part based on (Formula 45) in Embodiment 4. FIG. 6 is a simulation waveform showing a current control transient response in the second and fourth embodiments and the conventional techniques 2 and 3. FIG. The principle explanatory drawing of the prior art 4 (cascade type vector control system). The equivalent conversion block diagram (A) of the cascade type vector controller in the prior art 4. FIG. The voltage command production | generation part in Embodiment 5 of this invention. The equivalent conversion block diagram (B) of the cascade type vector controller in the prior art 4. FIG. The voltage command production | generation part in Embodiment 6 of this invention. The block diagram which shows the whole structure (position sensorless speed control system) of Embodiment 7 of this invention. The block diagram which shows the whole structure of Embodiment 8 of this invention. 9 is an example of an equivalent motor circuit that can be used in the eighth embodiment. The table data format used in Embodiment 9 of the present invention. The block diagram of the vehicle drive device by Embodiment 10 of this invention. The block diagram of the electric rear-wheel drive vehicle by Embodiment 11 of this invention.

  Next, with reference to FIGS. 1-32, embodiment of the control apparatus of the AC motor by this invention is described. In the following embodiments, a permanent magnet type synchronous motor (hereinafter abbreviated as PM motor) will be described as an AC motor. However, other motors (for example, a winding type synchronous motor, a reluctance motor, an induction motor, etc.) will be described. Is also feasible in the same way.

  First, in order to explain the embodiment of the present invention, it explains using a prior art. A voltage command generator in the prior art will be described with reference to FIG. The voltage command generation unit shown in FIG. 2 has almost the same configuration as the content described in Chapter 4 (hereinafter referred to as “Prior Art 3”) edited by Hidehiko Sugimoto, “Theory and Design of AC Servo Systems” (general electronic publisher).

In FIG. 2, i _d ^* is a d-axis current command value, i _q ^* is a q-axis current command value, i _d is a d-axis current detection value, and i _q is a q-axis current detection value. In the prior art 3, the i _d ^* and i _d of d-axis current deviation .DELTA.i _d is computed by a subtracter 102 calculates a i _q ^* and i _q of the q-axis current deviation .DELTA.i _q by subtractor 103. Then, the gain 104, by multiplying the d-axis inductance L _d of the PM motor current deviation .DELTA.i _d, calculates the d-axis magnetic flux deviation .DELTA..PHI _d. Similarly, in the gain 109, the q-axis magnetic flux deviation ΔΦ _q is calculated by multiplying the current deviation Δi _q by the q-axis inductance L _q of the PM motor. The ΔΦ _d and ΔΦ _q obtained in this way can be considered as compensation amounts of the respective axis magnetic flux components corresponding to the respective axis current deviations Δi _d and Δi _q . In the subsequent gains 108 and 112, the magnetic flux deviations ΔΦ _d and ΔΦ _q of the respective axes are multiplied by the current control response angular frequency ωacr (rad / s), respectively, and the voltage command values v _d ′ ^* and v _q before the voltage non-interference control are respectively obtained. By inputting to the adders 121 and 122 that calculate ' ^* , a current feedback control system that takes into account the winding current magnetic flux of the PM motor is constituted by each axis. On the other hand, integrators 105 and 110 with gain ωacr (rad / s) respectively input shaft current deviations Δi _d and Δi _q , and i is obtained by multiplying the integrated value by current control response angular frequency ωacr (rad / s). _d ′ and i _q ′ are output. Further, the gains 107 and 112 output values obtained by multiplying i _d ′ and i _q ′ by the PM motor armature winding resistance R, respectively, and voltage command values v _d ′ ^* and v _q ′ ^* before voltage non-interference control ^. Is input to the adders 121 and 122 for calculating the current feedback control system in consideration of the voltage drop due to the armature winding resistance of the PM motor.

Further, in the prior art 3 in FIG. 2, voltage non-interference control for canceling the voltage induced on the other axis by the magnetic flux defined on one of the two orthogonal control axes is as follows. It is configured. In d axis voltage value v _d ^* generation in gain 114, by multiplying the L _q to i _q, calculates a q-axis magnetic flux [Phi _q, in the multiplier 116, the electrical angular velocity of the PM motor rotor [Phi _q By multiplying by ω ₁ , the speed-induced voltage Φ _q ω ₁ generated by the q-axis magnetic flux Φ _q in the d-axis direction is calculated. The subtractor 117 outputs a d-axis voltage command value v _d ^* obtained by subtracting Φ _q ω ₁ from the d-axis voltage command value v _d ′ ^* before voltage non-interference control. This makes it possible to cancel out the speed induced voltage Φ _q ω ₁ derived from the q-axis magnetic flux Φ _q generated in the d-axis direction inside the PM motor. The same applies when the q-axis voltage command value v _q ^* generated, the gain 115, by multiplying the L _d to i _d, calculates the d-axis current magnetic flux .PHI.i _d, d-axis direction .PHI.i _d by the adder 120 The d-axis magnetic flux Φ _d is calculated by adding the permanent magnet magnetic flux Φ _m and the multiplier 118 multiplies Φ _d by the electrical angular velocity ω ₁ of the PM motor rotor so that the d-axis magnetic flux Φ _d is opposite to the q axis. The speed induced voltage Φ _d ω ₁ generated in the direction is calculated. The adder 119 outputs a q-axis voltage command value v _q ^* obtained by adding Φ _d ω ₁ to the q-axis voltage command value v _q ′ ^* before the voltage non-interference control. This makes it possible to cancel out the speed induced voltage Φ _d ω ₁ derived from the d axis magnetic flux Φ _d generated in the direction opposite to the q axis inside the PM motor.

However, in the prior art 3 of FIG. 2, the d-axis magnetic flux and the q-axis magnetic flux of the PM motor do not have current saturation, and the ideal characteristic is that there is no mutual interference effect of the magnetic flux between the d-axis and the q-axis. Yes. FIG. 3 shows a schematic diagram of the relationship between the magnetic flux and current of such an ideal PM motor. FIG. 3A shows a linear relationship between the d-axis current i _d and the d-axis magnetic flux Φ _d , and the relational expression is expressed by (Equation 19) when the permanent magnet magnetic flux in the d-axis direction is Φ _m. it can.

At this time, the d-axis magnetic flux deviation .DELTA..PHI _d is a value obtained by multiplying the d-axis current deviation .DELTA.i _d in slope (= L _d) of the equation (19) can be expressed by Equation (20).

Similarly, FIG. 3B shows a linear relationship between the q-axis current i _q and the q-axis magnetic flux Φ _q , and the relational expression can be expressed by (Expression 21).

At this time, the q-axis magnetic flux deviation ΔΦ _q is a value obtained by multiplying the slope (= L _q ) of (Expression 21) by the q-axis current deviation Δi _q and can be expressed by (Expression 22).

  Here, in (Equation 19) to (Equation 22), there are two important matters to be confirmed. First, the gains 104 and 109 for calculating the magnetic flux deviation from the current deviation correspond to (Equation 20) and (Equation 22), respectively, and mean the slopes of the graphs of (Equation 19) and (Equation 21), respectively. Is a point. Second, the gains 114 and 115 for calculating the current magnetic flux correspond to the first term on the right side of (Equation 19) and the value of the graph of (Equation 21), respectively.

In the present invention, the configuration means of the voltage command generation unit for the ideal PM motor described above is the configuration of the voltage command generation unit for the PM motor in which the magnetic flux saturation is remarkable and the mutual interference magnetic flux between the axes is large. Apply. That is, non-linear characteristics of the magnetic flux and current are introduced into the calculation formulas for the respective axis magnetic flux deviations and the respective axis magnetic fluxes corresponding to (Equation 19) to (Equation 22). FIG. 4 shows a schematic diagram of the relationship between the magnetic flux and current of a PM motor with strong non-linearity to be controlled by the present invention. FIG. 4A schematically shows the relationship between the d-axis current i _d and the q-axis current i _q on the d-axis magnetic flux Φ _d . Further, when the relational expression is Φ _m for the permanent magnet magnetic flux in the d-axis direction and k ₁ , k ₂ , k ₃ , k ₄ , K ₁₀ , and I _m0 , if i _d > 0 (Equation 23 ), If i _d <0, it is approximately given by (Equation 24).

Similarly, FIG. 4B schematically shows the relationship between the d-axis current i _d and the q-axis current i _q on the q-axis magnetic flux Φ _q, and the relationship is expressed by constants k ₅ , k ₆ , K _{When it is 20} , it is given approximately by (Equation 25).

The magnetic flux calculation formulas of (Equation 23) to (Equation 25) express the magnetic flux saturation and the mutual interference action of the magnetic flux between the d-axis and the q-axis with high accuracy. Further, although different from (Equation 23) to (Equation 25), constants are set as K ₁ , K ₂ , K ₃ , K ₄ , K ₅ , K ₆ as approximate equations for realizing magnetic flux calculation with the same high accuracy. , Φ ₀ , I ₀ , I ₁ , (Equation 26) and (Equation 27) may be used.

The above approximate expression (Expression 23), (Expression 24), (Expression 25) or (Expression 26), (Expression 27) is used instead of (Expression 19) and (Expression 21) to perform voltage non-interference control. Thus, high-accuracy voltage non-interference control can be realized even for a PM motor in which magnetic flux saturation is significant and there is a large amount of mutual interference magnetic flux between the axes. Further, ΔΦ _d and ΔΦ _q used in the current feedback control system are respectively calculated by (Equation 28) and (Equation 29) in consideration of the mutual interference magnetic flux between the axes. These (Equation 28) and (Equation 29) are obtained by extending (Equation 20) and (Equation 22) in the prior art 3 in consideration of the influence of the other-axis current. It is.

  With the above configuration, if (Equation 23) to (Equation 27) accurately represent the non-linear characteristics of the PM motor, high-precision control can be performed regardless of the magnetic flux saturation on the PM motor side and the mutual interference magnetic flux between the shafts. Can be expected. Therefore, the approximation levels of (Equation 23), (Equation 24), and (Equation 25) will be compared with the target value calculated by magnetic field analysis using a certain motor as an example.

FIG. 5A shows the target value calculated by the magnetic field analysis with respect to the magnetic flux Φ _d when the horizontal axis is i _d and i _q is changed to 0A, 100A, 200A, and 300A. It is the graph which compared with the approximate value calculated by the functional formula of Formula 24). FIG. 5B shows the target value calculated by the magnetic field analysis with respect to the magnetic flux Φ _q when the horizontal axis is i _q and i _d is changed to −200 A, −100 A, 0 A, 100 A, and 200 A. It is the graph which compared the approximate value calculated by the functional formula of (Formula 25).

Similarly, FIG. 6A shows the target value calculated by the magnetic field analysis with respect to the magnetic flux Φ _d when the horizontal axis is i _d and i _q is changed to 0A, 100A, 200A, and 300A (formula 26). ), An approximate value calculated by the functional expression of (Expression 27). FIG. 6B shows the target value calculated by the magnetic field analysis with respect to the magnetic flux Φ _q when i _q is taken on the horizontal axis and i _d is changed to −200 A, −100 A, 0 A, 100 A, and 200 A. It is the graph which compared the approximate value calculated with the functional formula of (Formula 27).

From the comparison results of FIGS. 5 (a), 5 (b) and FIGS. 6 (a), 6 (b), by using the functional equation approximation of (Equation 23) to (Equation 27), It can be confirmed that the influence of i _d and i _q on the d-axis magnetic flux φ _d and the q-axis magnetic flux φ _q can be satisfactorily approximated even for a motor having a strong and nonlinear characteristic. Therefore, high-precision calculations of (Expression 28) and (Expression 29) including partial differential functions based on i _d and i _q of (Expression 23) to (Expression 27) can also be realized.

  As described above, in the present invention, magnetic flux and current non-linear functions are prepared, and using them, the magnetic flux deviations in the voltage command generation unit and the operations corresponding to the magnetic fluxes of each axis are executed, whereby the magnetic flux Even for PM motors where saturation is significant and there are many inter-interference magnetic fluxes between the axes, the current control transient response as set is realized. In particular, the voltage command value to the power converter defined on one of the dq control axes orthogonal to each other in the rotational coordinate system of the motor is the current command value and current detection value defined on the same axis. And a deviation between the current command value defined on the other axis and the detected current value.

  Next, detailed embodiments of the present invention will be described with reference to the drawings.

Embodiment 1
FIG. 1 is a block diagram showing a configuration of a voltage command generator 1 in Embodiment 1 of an AC motor control device according to the present invention. FIG. 7 is a block diagram in which a torque control system is configured using the voltage command generation unit 1 configured in FIG. As shown in FIG. 7, the control device according to the first embodiment includes a torque command generator 30 that provides a torque command τ ^* to the PM motor 33, and a torque constant for the torque command τ ^* output from the torque command generator 30. a gain 31 for giving a reciprocal gain of kt to generate a q-axis current command value i _q ^* , an i _d ^* generating unit 32 for generating a d-axis current command value i _d ^* , and i _d ^* , i _q ^* And the d-axis current detection value i _d , the q-axis current detection value i _q , and the electrical angular velocity ω ₁ of the PM motor 33 are input, and the d-axis voltage command value v _d ^* and the q-axis voltage are input to the dq coordinate inverse conversion unit 37. The voltage command generator 1 that outputs the command value v _q ^* , the position detector 34 that provides the rotor position information of the PM motor 33, and the output signal of the position detector 34 is input to the electrical angle θ _{c of the} rotor. The electrical angle calculation unit 35 for outputting the output signal and the output signal of the position detector 34 are input, and the electric power of the PM motor rotor is input. And electrical angular velocity calculating unit 36 for outputting an angular velocity ^{_{ω 1, v d *, v}} q * the three-phase by an electrical angle theta _c AC voltage command _{^{v u *, v v *,}} v dq coordinate inverse transformation to convert the _w ^* Unit 37, a PWM inverter 38 that generates a three-phase AC voltage based on a three-phase AC voltage command, a U-phase current detector 39 that detects a U-phase current _iu output from the PWM inverter 38, and a PWM inverter 38 The current detector 40 for detecting the W-phase current i _w to be output, and the detected currents i _u and i _w by the electrical angle θ _c , the component i on each of the d and q axes orthogonal to each other in the rotational coordinate system of the PM motor. _{It comprises} a dq coordinate conversion unit 41 that converts coordinates to _d and i _q .

Further, in the first embodiment, the voltage command generation unit 1 having a characteristic structure, as shown in FIG. 1, has a d-axis current deviation Δi between the d-axis current command value i _d ^* and the d-axis current detection value i _d. a subtractor 2 for calculating _d , a subtractor 3 for calculating a q-axis current deviation Δi _q between the q-axis current command value i _q ^* and the q-axis current detection value i _q, and i _d , i _q , Δi _d , enter the .delta.i _q, the d-axis magnetic flux deviation calculation unit 4 for calculating a d-axis magnetic flux deviation .DELTA..PHI _d (i _d, i _q) in accordance with (equation 28), enter the d-axis current deviation .delta.i _d, the integral value 'and integral calculator 5 for outputting, i _d' current control response angular frequency ωacr (rad / s) multiplied by i _d enter a, an armature winding resistance R multiplied gain 7 of the PM motor 33, d-axis the magnetic flux deviation _{ΔΦ d (i d, i q} ) to enter, a current control response angular frequency ωacr (rad / s) multiplied gain 8, the voltage adds the output of the gain 7 and gain 8 An adder 6 for outputting the interference control before d-axis voltage command value ^{_{v d '*, i d,}} i q, Δi d, enter the .DELTA.i _q, q-axis magnetic flux deviation according equation (29) ΔΦ _q (i _d , I _q ), q-axis magnetic flux deviation calculation unit 9, and q-axis current deviation Δi _q are input, and the integral value is multiplied by current control response angular frequency ωacr (rad / s) to output i _q ′ The calculator 10 and i _q ′ are input, the gain 12 for multiplying the armature winding resistance R of the PM motor 33 and the q-axis magnetic flux deviation ΔΦ _q (i _d , i _q ) are input, and the current control response angular frequency a gain 13 that is multiplied by ωacr (rad / s), an adder 11 that adds the outputs of the gain 12 and the gain 13 and outputs a q-axis voltage command value v _q ′ ^* before voltage non-interference control, i _q , i _d And q-axis magnetic flux calculation unit 14 for calculating q-axis magnetic flux Φ _q ( _id , i _q ) based on (Equation 25), and i _q , i _d are inputted, (Equation 2 3), d-axis magnetic flux calculation unit 15 for calculating d-axis magnetic flux Φ _d ( _id , i _q ) based on (Equation 24), and Φ _q ( _id , i _q ) multiplied by the omega ₁ q-axis magnetic flux _{Φ q (i d, i q} ) has a multiplier 16 for calculating a speed induction voltage [Phi _q omega ₁ to generate the d-axis direction, v _d ^'* from [Phi _q omega ₁ subtraction The subtractor 17 that outputs the d-axis voltage command value v _d ^* subjected to voltage non-interference control, and Φ _d ( _id , i _q ) are multiplied by ω ₁ to obtain the d-axis magnetic flux Φ _d ( _id , i _q ) is a multiplier 18 for calculating a speed induced voltage Φ _d ω ₁ generated in the opposite direction to the q axis, and a q axis voltage command obtained by subtracting Φ _d ω ₁ from v _q ′ ^* and performing voltage non-interference control. And an adder 19 for outputting a value v _q ^* .

Next, the internal configuration of the d-axis magnetic flux deviation calculation unit 4 will be described with reference to FIG. In FIG. 8, i _d, as input i _q variables, d-axis flux _{Φ d (i d, i q} ) a _{i d ∂Φ d (i d,} i q) which is partially differentiated by outputting the / ∂i _d A function operation unit that has an operation block 50, outputs constants k ₃ , k ₄ , and K ₁₀ as parameters in advance and outputs the result when i _d ≧ 0 by calculating the right side of (Equation 30). 51 and constants k ₁ , k ₂ , and K ₁₀ given as parameters in advance, and a function calculation unit 52 that outputs a result when i _d <0 by calculating the right side of (Expression 31), The outputs of the calculation unit 51 and the function calculation unit 52 are input. If i _d ≧ 0, the output of the function calculation unit 51 is adopted, and if i _d <0, the output of the function calculation unit 52 is adopted, and ∂Φ _d ( comprises i _d, i _q) / ∂i selection means 53 for outputting a _d. Also, i _d, a i _q as an input variable, d-axis flux _{Φ d (i d, i q} ) to _{i q ∂Φ d (i d,} i q) obtained by partially differentiating / ∂i function operation for outputting a _q The unit 56 has constants K ₁₀ and I _m0 as parameters in advance, and calculates the right side of (Expression 32). The multiplier _{54, ∂Φ d (i d, i} q) output from the d-axis current deviation .DELTA.i _d and calculation block 50 multiplies the / ∂i _d, as the first item the right side of (Equation 28) (∂Φ _{d (i d, i q)} / ∂i d) outputting a .DELTA.i _d. The multiplier 57 multiplies the q-axis current deviation Δi _q by ∂Φ _d (i _d , i _q ) / ∂i _q output from the function calculation unit 56, and sets (∂Φ _{d (i d, i q)} / ∂i q) outputs a .DELTA.i _q. The adder 55 corresponds to the operation of adding the first item and right second item right side of (Equation 28), adds the output of multiplier 54 and the multiplier 57, d-axis magnetic flux deviation ΔΦ _d (i _d, i _q ) is output.

Next, the internal configuration of the q-axis magnetic flux deviation calculation unit 9 will be described with reference to FIG. In FIG. 9, i _d, a i _q as an input variable, q-axis magnetic flux _{Φ q (i d, i q} ) ∂Φ was partially differentiated by _{i d q (i d, i} q) for outputting the / ∂i _d The function calculation unit 60 and the constant K20 are given as parameters in advance, and the right side of (Expression 33) is calculated. Also, i _d, a i _q as an input variable, q-axis magnetic flux _{Φ q (i d, i q} ) to _{i q ∂Φ q (i d,} i q) obtained by partially differentiating / ∂i function operation for outputting a _q A unit 63, constants k ₅ , k ₆ , and K ₂₀ are given as parameters in advance, and the right side of (Equation 34) is calculated. The multiplier _{61, ∂Φ q (i d, i} q) output from the d-axis current deviation .DELTA.i _d and function calculating unit 60 multiplies the / ∂i _d, as the first item the right side of (Equation 29) (∂ _{Φ q (i d, i q} ) for outputting the / ∂i _d) Δi _d. The multiplier 64 multiplies the q-axis current deviation Δi _q by ∂Φ _q (i _d , i _q ) / ∂i _q output from the function calculation unit 63, and sets (∂Φ _{q (i d, i q)} / ∂i q) outputs a .DELTA.i _q. The adder 62 corresponds to the operation of adding the first item and right second item right-hand side of equation (29), adds the output of multiplier 61 and the multiplier 64, q-axis magnetic flux deviation ΔΦ _q (i _d, i _q ) is output.

Next, the internal configuration of the d-axis magnetic flux calculation unit 15 will be described with reference to FIG. In FIG. 10, a function calculation unit 70 that outputs d-axis magnetic flux Φ _d (i _d , i _q ) when i _d , i _q is an input variable and i _d ≧ 0 is provided, and constants k ₃ , k ₄ , K ₁₀ , I _m0 and permanent magnet magnetic flux φ _m are given in advance as parameters, and the right side of (Equation 23) is calculated. In addition, a function calculation unit 71 is provided that outputs i _d and i _q as input variables and outputs a d-axis magnetic flux Φ _d (i _d , i _q ) when i _d <0. Constants k ₁ , k ₂ , K ₁₀ , I _m0 and permanent magnet magnetic flux φ _m are given in advance as parameters, and the right side of (Equation 24) is calculated. Each output result of the function calculation unit 70 and the function calculation unit 71 is input. If i _d ≧ 0, the output of the function calculation unit 70 is selected. If i _d <0, the output of the function calculation unit 71 is selected. It has a selection means 72 and outputs a d-axis magnetic flux Φ _d (i _d , i _q ).

Next, details of the q-axis magnetic flux calculator 14 will be described with reference to FIG. In FIG. 11, function calculation processing means 14 for executing function calculation processing for outputting q-axis magnetic flux Φ _d ( _id , i _q ) with i _d , i _q as input variables, constants k ₅ , k ₆ , K ₂₀ is given as a parameter in advance, and the right side of (Equation 25) is calculated and output.

  In the first embodiment described above, by introducing the magnetic flux calculation expression expressed by (Expression 23), (Expression 24), and (Expression 25) in the torque control system, the magnetic flux saturation is remarkable, and A current control transient response as set is also realized for a PM motor having a large amount of mutual interference magnetic flux.

[Embodiment 2]
In the first embodiment, the magnetic flux calculation expression expressed by (Equation 23), (Equation 24), and (Equation 25) is introduced in the torque control system. However, the second embodiment described below has the same configuration as the first embodiment. Another magnetic flux calculation expression expressed by (Expression 26) and (Expression 27) is introduced into the torque control system. Therefore, also in the second embodiment, FIG. 1 showing the configuration of the voltage command generation unit 1 and FIG. 7 showing the configuration of the torque control system including the voltage command generation unit 1 shown in FIG. 1 are the same.

  The difference from the first embodiment is that the d-axis magnetic flux deviation calculation unit 4, the q-axis magnetic flux deviation calculation unit 9, the q-axis magnetic flux calculation unit 14, and the d-axis magnetic flux calculation unit 15 in the first embodiment are respectively described in the d-axis. This is a point replaced with a magnetic flux deviation calculation unit 80, a q-axis magnetic flux deviation calculation unit 90, a q-axis magnetic flux calculation unit 98, and a d-axis magnetic flux calculation unit 99.

First, the internal configuration of the d-axis magnetic flux deviation calculation unit 80 will be described with reference to FIG. In FIG. 12, i _d, as input i _q variables, d-axis flux _{Φ d (i d, i q} ) a _{i d ∂Φ d (i d,} i q) which is partially differentiated by outputting the / ∂i _d The function calculation unit 81 and constants K ₁ , K ₂ , K ₃ , and I ₀ are given in advance as parameters, and the right side of (Expression 35) is calculated. An operation block 84 that outputs ∂Φ _d (i _d , i _q ) / ∂i _q obtained by partial differentiation of the d-axis magnetic flux Φ _d (i _d , i _q ) by i _q with i _d and i _q as input variables. Have. Constants K ₁ , K ₂ , K ₃ , and I ₀ are given as parameters in advance, and a function calculation unit 85 that outputs a result when i _q ≧ 0 by calculating the right side of (Equation 36), and a constant A function operation unit 86 that outputs K ₁ , K ₂ , K ₃ , and I ₀ as parameters in advance and outputs a result when i _q <0 by calculating the right side of (Expression 37); , And the output of the function calculation unit 85 is adopted if i _q ≧ 0, and the output of the function calculation unit 86 is adopted if i _q <0, and ΦΦ _d (i _d, comprising a i _q) / ∂i _q selection means 87 for outputting a.

The multiplier _{82, ∂Φ d (i d, i} q) output from the d-axis current deviation .DELTA.i _d and function calculating unit 81 multiplies the / ∂i _d, as the first item the right side of (Equation 28) (∂ _{Φ d (i d, i q} ) for outputting the / ∂i _d) Δi _d. The multiplier 88 multiplies the q-axis current deviation Δi _q by ∂Φ _d (i _d , i _q ) / ∂i _q output from the calculation block 84, and sets (∂Φ _d as the second item on the right side of (Equation 28). (I _d , i _q ) / ∂i _q ) Δi _q is output. The adder 83 corresponds to the operation of adding the first item and right second item right side of (Equation 28), adds the output of multiplier 82 and the multiplier 88, d-axis magnetic flux deviation ΔΦ _d (i _d, i _q ) is output.

Next, the internal configuration of the q-axis magnetic flux deviation calculator 90 will be described with reference to FIG. In FIG. 13, ｉΦ _q (i _d , i _q ) / ∂ i _{d obtained} by partial differentiation of the q-axis magnetic flux Φ _q (i _d , i _q ) by i _d is output using i _d and i _q as input variables. The calculation block 600 and constants K ₄ , K ₅ , K ₆ , and I ₁ are given in advance as parameters, and the result when i _d + I ₁ ≧ 0 is output by calculating the right side of (Equation 38). The function calculation unit 91 and constants K ₄ , K ₅ , K ₆ , and I ₁ are given in advance as parameters, and the result when i _d + I ₁ <0 is output by calculating the right side of (Equation 39). Function calculation unit 92, and the output of function calculation unit 91 and function calculation unit 92 are input. If i _d + I ₁ ≧ 0, the output of function calculation unit 91 is adopted, and if i _d + I ₁ <0, function calculation is performed. adopted output parts 92, a selection unit 93 for outputting _{∂Φ q (i d, i q} ) as / ∂i _d. Also, i _d, a i _q as an input variable, q-axis magnetic flux _{Φ q (i d, i q} ) to _{i q ∂Φ q (i d,} i q) obtained by partially differentiating / ∂i function operation for outputting a _q The unit 96 has constants K ₄ , K ₅ , K ₆ , and I ₁ given as parameters in advance, and calculates the right side of (Equation 40). The multiplier _{94, ∂Φ q (i d, i} q) output from the d-axis current deviation .DELTA.i _d and calculation block 500 multiplies the / ∂i _d, as the first item the right side of (Equation 29) (∂Φ _{q (i d, i q)} / ∂i d) outputting a .DELTA.i _d. The multiplier 97 multiplies the q-axis current deviation Δi _q by ∂Φ _q (i _d , i _q ) / ∂i _q output from the function calculation unit 96, and sets (∂Φ _{q (i d, i q)} / ∂i q) outputs a .DELTA.i _q. The adder 95 corresponds to the operation of adding the first item and right second item right-hand side of equation (29), adds the output of the multiplier 94 and the multiplier 97, q-axis magnetic flux deviation ΔΦ _q (i _d, i _q ) is output.

Next, details of the d-axis magnetic flux calculation unit 99 will be described with reference to FIG. In FIG. 14, a function calculation processing unit 99 that performs a function calculation process for outputting d-axis magnetic flux Φ _d ( _id , i _q ) with i _d , i _q as input variables is a constant K ₁ , K ₂ , K ₃ , I ₀ , φ ₀ are given as parameters in advance, and the right side of (Equation 26) is calculated and output.

Next, details of the q-axis magnetic flux calculation unit 98 will be described with reference to FIG. In FIG. 15, a function calculation processing unit 98 that performs a function calculation process for outputting q-axis magnetic flux Φ _q ( _id , i _q ) with i _d and i _q as input variables is a constant K ₄ , K ₅ , K ₆ and I ₁ are given in advance as parameters, and the right side of (Equation 27) is calculated and output.

  In the second embodiment described above, by introducing the magnetic flux calculation expressions expressed by (Equation 26) and (Equation 27) in the torque control system, the magnetic flux saturation is remarkable and the mutual interference magnetic flux between the axes is large. Also for the existing PM motor, the current control transient response as set is realized as in the first embodiment.

[Embodiment 3]
In the first embodiment, the d-axis current detection value i _d and the q-axis current detection value i _q are used when calculating the d-axis magnetic flux Φ _d and the q-axis magnetic flux Φ _q used for voltage non-interference control. In the third embodiment, voltage non-interference control using the d-axis current command value i _d ^* and the q-axis current command value i _q ^* is performed on the torque control system having the same configuration as that of the first embodiment. Therefore, also in the third embodiment, the overall configuration of the torque control system is FIG.

The difference from the first embodiment is that the d-axis magnetic flux Φ _d and the q-axis magnetic flux Φ _q in the voltage command generating unit 1 of the first embodiment are changed into a d-axis simulated magnetic flux Φ _{d as in the} voltage command generating unit 200 shown in FIG. It is only the point replaced with 'and q-axis simulated magnetic flux Φ _q '.

  First, the internal configuration of the voltage command generator 200, which is a feature of the third embodiment, will be described with reference to FIG. In FIG. 16, the configurations of reference numerals 2-13 and 16-18 are the same as or equivalent to those described in the first embodiment. Further, the d-axis magnetic flux deviation calculator 4 uses the d-axis magnetic flux deviation calculator 4 shown in FIG. 8, and the q-axis magnetic flux deviation calculator 9 uses the q-axis magnetic flux deviation calculator 9 shown in FIG. .

First-order lag filter 201 that receives q-axis current command value i _q ^* and outputs q-axis current simulation value i _q ′ sets its cutoff angular frequency equal to current control response angular frequency ωacr (rad / s). To do. Similarly, the first-order lag filter 202 that receives the d-axis current command value i _d ^* and outputs the d-axis current simulation value i _d ′ changes its cutoff angular frequency to the current control response angular frequency ωacr (rad / s). Set equal. Q-axis simulated magnetic flux which inputs i _d ′ and i _q ′ output from the first-order lag filters 201 and 202 and calculates the q-axis simulated magnetic flux Φ _q ′ (id _d , i _q ′) based on (Equation 41) The calculation unit 203 includes i _d ′ and i _q ′ output from the first-order lag filters 201 and 202, and the d-axis simulated magnetic flux Φ _d ′ (i _d ′) based on (Equation 42) and (Equation 43). , I _q ′) is provided. In multipliers 16 and 18, the q-axis simulated magnetic fluxes Φ _q ′ ( _id ′, i _q ′) and Φ _d ′ ( _id ′, i _q ′) are set to the electrical angular velocity ω ₁ of the PM motor rotor. By multiplying, the speed-induced voltages Φ _q ′ ω ₁ and Φ _d ′ ω ₁ generated with respect to different orthogonal axes are estimated and input to the subtracter 17 and the adder 19, respectively. Is offset by

Next, the internal configuration of the d-axis simulated magnetic flux calculation unit 204 will be described with reference to FIG. In FIG. 17, a function calculation unit 210 that outputs i _d ′, i _q ′ as input variables and outputs a d-axis simulated magnetic flux Φ _d ′ (i _d ′, i _q ′) when i _d ′ ≧ 0 is a constant. _{k 3, k 4, K 10} , previously given beforehand I _m0 and the permanent magnet flux phi _m as a parameter, was replaced with the input variable i _d equation (23), i _q from i _d ', i _q' The right side of (Expression 42) is calculated. The function calculation unit 211 that outputs i _d ′, i _q ′ as input variables and outputs a simulated d-axis magnetic flux Φ _d ′ (i _d ′, i _q ′) when i _d ′ <0 is a constant k ₁ , k _2, K _10, advance give I _m0 and the permanent magnet flux phi _m in advance as a parameter, was replaced with the input variable i _d (formula 24), i _q from i _d ', i _q' (formula 43) is calculated. Each output result of the function calculation unit 210 and the function calculation unit 211 is input. If i _d ′ ≧ 0, the output of the function calculation unit 210 is selected. If i _d ′ <0, the output of the function calculation unit 211 is output. The selection means 212 for selecting outputs the d-axis simulated magnetic flux Φ _d ′ (i _d ′, i _q ′).

Next, details of the q-axis simulated magnetic flux calculation unit 203 will be described with reference to FIG. In FIG. 18, q-axis simulated magnetic flux calculation unit 203 that performs a function calculation process that outputs i- _d ′, i _q ′ as input variables and outputs q-axis simulated magnetic flux Φ _q ′ ( _id ′, i _q ′) constant k _5, previously given to pre k _6, K ₂₀ as a parameter, calculating the right side of the input variable i _d (formula 25), i _q from i _d ', i _q' was replaced by (equation 41) And output.

In the third embodiment described above, voltage non-interference control is performed using the d-axis current command value i _d ^* and the q-axis current command value i _q ^* in the torque control system. Similarly to the first embodiment, a current control transient response as set is realized for a PM motor having a large amount of mutual interference magnetic flux between the axes.

[Embodiment 4]
In the third embodiment, a magnetic flux calculation expression expressed by (Equation 23), (Equation 24), and (Equation 25) is introduced in the torque control system in the same way as in the first embodiment. Another magnetic flux calculation expression expressed by (Expression 26) and (Expression 27) is introduced to the torque control system having the same configuration as that of Embodiment 3. Therefore, also in the fourth embodiment, FIG. 16 showing the configuration of the voltage command generation unit 200 and FIG. 7 showing the configuration of the torque control system including the voltage command generation unit 200 shown in FIG. 16 are the same.

The difference from the third embodiment is that the d-axis magnetic flux deviation calculating section 4, the q-axis magnetic flux deviation calculating section 9, the q-axis simulated magnetic flux calculating section 203, and the d-axis simulated magnetic flux calculating section 204 in the third embodiment are respectively changed to the d-axis. The magnetic flux deviation calculating unit 80, the q-axis magnetic flux deviation calculating unit 90, the q-axis simulated magnetic flux calculating unit 223, and the d-axis simulated magnetic flux calculating unit 224 are replaced. Among these, the d-axis magnetic flux deviation calculator 80 and the q-axis magnetic flux deviation calculator 90 have been described in the second embodiment. First, details of the d-axis simulated magnetic flux calculation unit 224 will be described with reference to FIG. In FIG. 19, the d-axis simulated magnetic flux calculation unit 224 that performs the function calculation process of outputting d-axis simulated magnetic flux Φ _d ′ ( _id ′, i _q ′) with i _d ′, i _q ′ as input variables is constant _{K 1, K 2, K 3} , I 0, previously given to advance phi ₀ as a parameter, it was replaced with the input variable i _d equation (26), i _q from i _d ', i _q' (formula 44) is computed and output.

Next, details of the q-axis simulated magnetic flux calculation unit 223 will be described with reference to FIG. In FIG. 20, a q-axis simulated magnetic flux calculation unit 223 that performs a function calculation process for outputting q-axis simulated magnetic flux Φ _q ′ (i _d ′, i _q ′) with i _d ′, i _q ′ as input variables is constant _{K 4, K 5, K 6} , previously given beforehand I ₁ as a parameter of the input variable i _d (formula 27), i _q from i _d ', i _q' was replaced by (equation 45) Calculate and output the right side.

  In the fourth embodiment described above, magnetic flux saturation is remarkable by introducing the magnetic flux calculation expressions expressed by (Expression 26) and (Expression 27) to the torque control system having the same configuration as that of the third embodiment. Thus, as with the third embodiment, the current control transient response as set is also realized for the PM motor having a large amount of mutual interference magnetic flux between the axes.

Here, the current control transient response simulation waveforms of the embodiments 2 and 4 described above are shown in FIG. As a model formula of the PM motor used for the simulation, a function obtained by resolving (Equation 26) and (Equation 27) with respect to i _d and i _q was used. Further, the current command value is set to i _d ^* = 0 [A], and i _q ^* is stepped from 0 [A] to 400 [A] at which magnetic flux saturation is noticeably generated at time 1.0 [s]. The shape was changed. In FIG. 21, FIG. 21 (a) shows the transient response result of the q-axis current i _q , and FIG. 21 (b) shows the transient response result of the d-axis current i _d . The waveforms 230 and 234 correspond to the prior art 3 that does not consider the magnetic flux saturation at all, and the waveforms 231 and 235 consider the magnetic flux saturation in the current feedback control but the magnetic flux interference action between the d axis and the q axis. Corresponds to the prior art 2 not considered, and the waveforms 232, 236, 233, and 237 correspond to the second and fourth embodiments, and are almost the same waveform. From the above results, in Prior Art 3, not only the rising waveform 230 of the q-axis current deviates from the ideal first-order lag waveform, but also the _d -axis current waveform 234 that should be originally maintained at i _d = 0 [A]. It can be confirmed that it fluctuates greatly from 0 [A]. On the other hand, in the conventional technique 2, the rising waveform 231 of the q-axis current is an ideal first-order lag waveform, but the d-axis current waveform 235 varies from 0 [A]. On the other hand, in the second and fourth embodiments, it is confirmed that the rising waveforms 232 and 233 of the q-axis current are ideal first-order lag waveforms, and that the d-axis current waveforms 236 and 237 maintain 0 [A]. it can. Such excellent control performance is considered to be an effect of configuring the feedback control system in consideration of the mutual interference action of the magnetic flux between the d-axis and the q-axis in the above embodiment.

[Embodiment 5]
In the fifth embodiment, FIG. 24 is used as the voltage command generation unit 300 in the torque control system shown in FIG. 7 described above. FIG. 24 is a diagram of the 2004 Annual Meeting of the Institute of Electrical Engineers of Japan, No. 1-12, I (Hei 16-8) “Stability Analysis in New Vector Control System of Permanent Magnet Synchronous Motor for High Speed”. In contrast to the “cascade type vector control system” (hereinafter referred to as prior art 4) proposed by Endo, Iwaji, Ito, et al. (Equation 23), 24), (Equation 25) or a configuration in which a magnetic flux calculation expression expressed by (Equation 26) and (Equation 27) is introduced.

First, the basic prior art 4 constituting the fifth embodiment will be described with reference to FIG. FIG. 22 is a block diagram representing the configuration principle of prior art 4. In FIG. 22, the d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* are respectively expressed in the d-axis direction and the q-axis. and applied voltage component in the direction, a motor model 266 which the result as the d-axis current detection value i _d and the q-axis current detection value i _q, d-axis direction, respectively, the inflow current component in the q-axis direction, i _d ^*, i _q ^* are each d-axis current command value and q-axis current command value, i _d ^* and i _d is subtracted from, a subtracter 262 for calculating an axis current deviation .DELTA.i _d, i _q subtracted from i _q ^* and, enter a subtracter 263 for calculating axis current deviation .DELTA.i _q, enter the .DELTA.i _d, an integrator 260 with gain ωacr for calculating a second d-axis current command value i _d ^**, the .DELTA.i _q, second q-axis current command value i _q ^** calculates the dated gain ωacr integrator 261, i _d ^**, enter the i _q ^**, v _d ^*, This is a motor inverse model 267 that realizes the inverse transfer function characteristics of the motor model 266 that outputs v _q ^* . Further, in the motor model 266, the block 250 represents a transfer characteristic for the _{d d} v _d ^* in the d axis direction when the rotor is stationary, where R is the armature winding resistance of the motor, and L _d is the d axis of the motor. Inductance, s means a Laplace operator. Similarly, this is a block 251 expressing the transfer characteristic of i _q with respect to v _q ^* in the q-axis direction in the rotor stationary state, and L _q means the q-axis inductance of the motor. i _q on by multiplying the electrical angular velocity omega ₁ and L _q of the motor rotor, the calculator 252 i _q is computed velocity induced voltage [Phi _q 'omega ₁ to generate the d-axis direction. By adding the calculated speed induced voltage Φ _q ′ ω ₁ to v _d ^* , the adder 254, i _d reflected in the d-axis direction voltage is multiplied by the electrical angular velocity ω ₁ and L _d of the motor rotor, A calculator 253 for calculating a speed induced voltage Φ _d ′ ω ₁ generated by i _d in the q-axis direction. The subtractor 255 reflects the calculated speed-induced voltage Φ _q ′ ω ₁ and the speed-induced voltage ω ₁ Φ _m caused by the permanent magnet magnetic flux φ _m from the v _q ^* to reflect them in the q-axis direction voltage. On the other hand, in the motor reverse model 267, the reverse transfer characteristic blocks 256 and 257 are arranged in series with the blocks 250 and 251 of the motor model 266 as the reverse transfer characteristic blocks of the blocks 250 and 251, respectively. Accordingly, if the influence of the speed induced voltage is ignored, the total transfer function is 1 in the block in which the motor inverse model 267 and the motor model 266 are connected in series as shown in FIG. 22, and the input and the output theoretically match. . That is, i _d ^** = i _d and i _q ^** = i _q hold. Here, if it is assumed that i _d ^** = _id and i _q ^** = i _q holds, the calculator 258 that multiplies i _q ^** by the electrical angular velocity ω ₁ and L _q of the motor rotor. The output becomes equal to Φ _q ′ ω ₁ , and is subtracted from the output of the reverse transfer characteristic block 256 by the subtractor 264, so that the speed induced voltage Φ _q that i _q is generated in the d-axis direction inside the motor model 266. ′ Ω ₁ can be offset. Similarly, the output of the calculator 259 that multiplies i _d ^** by the electrical angular velocity ω ₁ and L _d of the motor rotor is equal to Φ _d ′ ω ₁ , and this and ω ₁ Φ _m are reversed in the adder 265. By adding to the output of the transfer characteristic block 257, in the motor model 266, the speed induced voltage Φ _d ′ ω ₁ generated by i _d in the d-axis direction and the speed induced voltage ω ₁ Φ _m due to the permanent magnet magnetic flux φ _{m are} generated. Can be offset. Thus, in FIG. 22, i _d ^** = i _d and i _q ^** = i _q always hold regardless of the speed induced voltage inside the motor. Therefore, only the integrators 260 and 261 with gain ωacr exist in the closed loop of the current control system of each axis, and the closed loop characteristics thereof are the first-order lag characteristics of the gain crossing angular frequency ωacr (rad / s). It becomes.

FIG. 23 is a block diagram obtained by equivalently converting the cascade type vector controller 268 in the principle explanatory diagram of the prior art 4 shown in FIG. In FIG. 23, the configuration of reference numerals 102-105, 107-110, 112, 113, 121, 122 and i _d ^* , i _q ^* , i _d , i _q , i _d ', i _q ', ω ₁ , v ^{_{d *, v q *, v}} d '*, v q' *, Δi d, Δi q, ΔΦ d, ΔΦ q, φ m is the same and equal to the prior art 3 (Figure 2). Also, the integrator 280 receives the output of the gain 108, d-axis current 'calculates the, .PHI.i _d adder 286' simulated flux .PHI.i _d d-axis simulated magnetic flux to the sum of the permanent magnet flux [Phi _m in the d-axis direction Φ _d ′ is calculated, and the multiplier 281 multiplies Φ _d ′ by the electrical angular velocity ω ₁ of the PM motor rotor, thereby generating a speed-induced voltage Φ generated by the d-axis simulated magnetic flux Φ _d ′ in the direction opposite to the q axis. to calculate the _{d 'ω 1.} The adder 285 outputs a d-axis voltage command value v _q ^* obtained by adding Φ _d ′ ω ₁ to the q-axis voltage command value v _q ′ ^* before voltage non-interference control. As a result, the speed induced voltage Φ _d ′ ω ₁ derived from the d axis simulated magnetic flux Φ _d ′ generated in the direction opposite to the q axis inside the PM motor is canceled. The same applies to the generation of the d-axis voltage command value v _d ^* . The integrator 282 calculates the q-axis simulated magnetic flux Φ _q ′ from the output of the gain 113, and the multiplier 283 multiplies Φ _q ′ by ω _1. Thus, the speed-induced voltage Φ _q ′ ω ₁ generated by the q-axis simulated magnetic flux Φ _q ′ in the d-axis direction is calculated. The subtractor 284 outputs a d-axis voltage command value v _d ^* obtained by subtracting Φ _q ′ ω ₁ from the d-axis voltage command value v _d ′ ^* before voltage non-interference control. As a result, the speed-induced voltage Φ _q ′ ω ₁ corresponding to the q-axis simulated magnetic flux Φ _q ′ generated in the d-axis direction inside the PM motor is canceled.

For the equivalent conversion block diagram (A) of the cascade type vector controller of FIG. 23 described above, a magnetic flux calculation expression expressed by (Expression 23), (Expression 24), (Expression 25), or (Expression 26), Another magnetic flux calculation expression expressed by (Expression 27) is introduced. Configuration of the voltage command generating unit 300 is 24, the gain 104 in FIG. _{23, i d, i q,} Δi d, enter the .DELTA.i _q, d-axis magnetic flux deviation .DELTA..PHI _d (i accordance (Formula 28) _d, the d-axis magnetic flux deviation calculation unit 4 calculates a i _q), or is replaced with the d-axis magnetic flux deviation calculation unit 80, the gain 109 in FIG. 23, i _d, i _q, .DELTA.i _d, the .DELTA.i _q The q-axis magnetic flux deviation calculator 9 or the q-axis magnetic flux deviation calculator 90 that calculates the q-axis magnetic flux deviation ΔΦ _q (i _d , i _q ) according to (Equation 29) is substituted. However, in the case where the magnetic flux calculation expression expressed by (Expression 26) and (Expression 27) is introduced, the permanent magnet magnetic flux φ _m is calculated by (Expression 46).

  According to the fifth embodiment, the high stability in the high-speed rotation region when the current control is performed with the low sampling calculation period, and the magnetic flux saturation, which are the features of the cascade vector controller, are remarkable. It is possible to achieve both current control transient response accuracy for a PM motor having a large amount of mutual interference magnetic flux.

[Embodiment 6]
In the sixth embodiment, as in the fifth embodiment, the voltage command generation unit 400 in the torque control system shown in FIG. 7 described above is compared to the conventional technique 4 with (Expression 23), (Expression 24), (Expression). 25) or a magnetic flux calculation expression expressed by (Expression 26) or (Expression 27). Specifically, the above-described magnetic flux calculation formula is introduced into the equivalent conversion block diagram (B) of the cascade type vector controller of FIG. FIG. 25, which is an introduction target of the magnetic flux calculation formula, is equivalent to the equivalent conversion block diagram (A) of the cascade vector controller described in FIG. 23, but the d-axis current simulated magnetic flux Φi _d ′ and the q-axis simulated magnetic flux Φi. Only the means for calculating _q ′ is different.

FIG. 26 shows a voltage command generation unit 400 according to the sixth embodiment. The gain 104 in FIG. 25 is inputted with i _d , i _q , Δi _d , Δi _q, and d-axis magnetic flux deviation ΔΦ _d ( i _d, the d-axis magnetic flux deviation calculation unit 4 calculates a i _q), or is replaced with the d-axis magnetic flux deviation calculation unit 80, the gain 109 in FIG. _{25, i d, i q,} Δi d, Δi q enter a, the q-axis magnetic flux deviation calculation unit 9 calculates a q-axis magnetic flux deviation .DELTA..PHI _q (i _d, i _q) in accordance with equation (29), or replaced with the q-axis magnetic flux deviation calculation unit 90 further The calculation means for the d-axis simulated magnetic flux Φ _d ′ composed of the gain 310 and the adder 286 in FIG. 25 is inputted with i _d ′, i _q ′, and the q-axis simulated magnetic flux Φ _q ′ (i _d ', i _q' wherein q-axis simulated magnetic flux calculating unit 203 calculates a) or a group (formula 45) There are q-axis simulated flux _{Φ q '(i d',} i q ') is replaced with the q-axis simulated magnetic flux calculation unit 223 for calculating a gain 311 in FIG. 25, i _d' enter the, i _q ', The d-axis simulated magnetic flux calculating unit 204 that calculates the d-axis simulated magnetic flux Φ _d ′ ( _id ′, i _q ′) based on (Expression 42) and (Expression 43), or d based on (Expression 44). This is a substitute for the d-axis simulated magnetic flux calculation unit 224 that calculates the simulated axis magnetic flux Φ _d ′ (i _d ′, i _q ′).

  According to the sixth embodiment, as in the fifth embodiment, the high stability in the high-speed rotation region and the magnetic flux saturation when the current control is performed with the low sampling calculation period, which are the features of the cascade vector controller. The current control transient response accuracy can be achieved for a PM motor having a large amount of mutual interference magnetic flux between the axes.

[Embodiment 7]
FIG. 27 is a block diagram showing a configuration of an AC motor control device according to Embodiment 7 of the present invention. Figure 27 constitutes a speed control system according to position and speed sensorless, the speed command generating portion 415 to provide a speed command omega ₁ ^* to the PM motor 33, i to generate a d-axis current command value i _d ^* _d ^* The generator 32, a subtractor 410 that subtracts the electrical angular velocity ω ₁ of the PM motor 33 from ω ₁ ^* and outputs a velocity deviation ω _e , performs a proportional-integral operation on ω _e , and performs a q-axis current command value i _q ^The speed controller 411 that outputs ^* , and the i _d ^* , i _q ^{*, the} d axis current detection value i _d , the q axis current detection value i _q , and the electrical angular velocity ω ₁ of the PM motor 33 are input, and the dq coordinate reverse conversion is performed. Voltage command generation unit 1, voltage command generation unit 200, voltage command generation unit 300, or voltage command generation unit that outputs d-axis voltage command value v _d ^* and q-axis voltage command value v _q ^* to unit 37 and ^{_{400, v d *, v q}} *, i d, i q, enter the ω _1, (equation 25) or ( Is calculated from 27) q-axis magnetic flux Φ _q (i _d, and i _q), the axis error estimator for calculating an estimated axis error Δθ by using a resistance set value R of the motor (the formula 47) (formula 48) Unit 412, a PLL controller 413 that performs proportional integral calculation or proportional calculation on Δθ and outputs electric angular velocity ω ₁ , and integrator 414 that integrates ω ₁ and outputs electric angle θ _c of the motor rotor. When, v _d ^*, the v _q ^*, the electrical angle theta _c three-phase AC voltage command by ^{_{v u *, v v *,}} v and dq coordinate inverse transformation unit 37 to be converted to _w ^*, based on the three-phase AC voltage command A PWM inverter 38 that generates a three-phase AC voltage, a current detector 39 that detects a U-phase current i _u output from the PWM inverter 38, and a current detector that detects a W-phase current i _w output from the PWM inverter 38. 40, the detected current i _u, the i _w, by electrical angle theta _c, PM D perpendicular to the rotational coordinate system over data consists dq coordinate converter 41 for coordinate conversion components i _d, i _q on the q axes.

  In the seventh embodiment, a magnetic flux calculation formula is introduced into the voltage command generation unit, and the configuration shown in FIG. 1, FIG. 16, FIG. 24, or FIG. Furthermore, by introducing a magnetic flux calculation formula to the axis error estimation unit in the position / speed sensorless speed control system, even for PM motors with a large amount of inter-interference magnetic flux between the axes, the rotor can be accurately detected. Position information can be acquired and highly responsive control can be realized.

[Embodiment 8]
FIG. 28 is a block diagram showing a configuration of an AC motor control apparatus according to Embodiment 8 of the present invention. FIG. 28 shows a configuration for discriminating the presence or absence of a voltage vector generation unit characteristic of the present invention from only information of a measurement system added outside in the torque control system shown in FIG. Therefore, the configuration and symbols of the control system are basically the same as those in FIG. 7, and only the measurement system is added. However, for the PM motor connected to the PWM inverter 38, an ideal motor 420 that does not generate magnetic flux saturation or magnetic flux interference internally is prepared. As an alternative to such an ideal motor, as shown in FIG. 29, a circuit in which three series connection elements of a single resistor R and a single inductance L are star-connected may be used.

Hereinafter, the measurement system added to FIG. 7, its role, and specific means for determining the presence or absence of a voltage vector generation unit characteristic of the present invention will be shown. First, the position detector 34 that provides the rotor position information of the PM motor 33 is removed from the shaft of the PM motor 33 so that the electrical angle θ _c output from the electrical angle calculation unit 35 does not change involuntarily. Next, the torque command generator 30 generates a torque command τ ^* that is equal to or less than the rated torque of the motor 420 or the PWM rated current. At this time, the detection value of the current detector 421 that detects the newly added U-phase current i _u is observed with an oscilloscope 424 to which the detection signal is connected, and it is confirmed that some DC current is flowing. If the direct current cannot be confirmed, the shaft of the position detector 34 is turned to confirm that the direct current increases or decreases positively and negatively, and then the position detector 34 at a position where the observation current of the oscilloscope 1 becomes zero. Fix the shaft. As a result, the U-phase current i _u matches the d-axis current. Next, the output line of the alternating current source 427 that generates a specific frequency is passed through the U-phase current detector 39. As a result, a signal equal to the frequency generated by the AC current source 427 is superimposed on the current detection signal i _u (now equal to i _d ) of the U-phase current detector 39 of the torque control system. Here, if the voltage vector generator does not have the characteristic structure of the present invention, the measurement conditions are such that magnetic flux interference between the dq axes inside the motor 420 and interference terms due to speed induced voltage do not occur. Considering this, the frequency component generated by the alternating current source 427 superimposed on the d-axis current detection value i _d should not be detected in the q-axis current. Therefore, from the detected value of the current detector 422 for detecting the the V-phase current i _v, newly added, subtracted detected value of the current detector 423 for detecting the newly added W-phase current i _w by the subtractor 425 , I _q √3 is calculated. A signal proportional to the magnitude of i _q obtained in this way is observed with an oscilloscope 426, and the presence or absence of a signal equal to the frequency generated by the alternating current source 427 is confirmed. It can be said that there is a department.

[Embodiment 9]
In the first to eighth embodiments described above, the magnetic flux deviation calculation unit and the magnetic flux calculation unit are calculated using functions relating to i _q and i _d . However, even if these are given as two-dimensional table data as shown in FIG. 30, equivalent functions can be realized at higher speed. For example, instead of the function calculation unit 60 in the q-axis magnetic flux deviation calculation unit 9, table data as shown in FIG. 30 in which the calculation result of the right side of (Expression 33) is stored in advance is prepared. Then, the element closest to i _q and i _d is selected and output.

  In the ninth embodiment described above, in the torque control system, the magnetic flux saturation is remarkable, and the PM motor having a large amount of mutual interference magnetic flux between the shafts is also set as in the first to eighth embodiments. A current control transient response is realized.

[Embodiment 10]
Further, the present embodiment shown in FIG. 31 applies the torque control system of the first to sixth embodiments described above to a vehicle drive device. In this embodiment, the direct current received from the overhead line 500 via the pantograph 501 is input to the PWM inverter 38 via the power receiving filter 502. The power receiving filter 502 includes a filter reactor 502a and a filter capacitor 502b, and smoothes the ripple current from the PWM inverter 38. The master controller 505 converts the driver's notch operation into a torque command τ ^* and inputs it to the torque control system of the first to sixth embodiments. The PM motor 33 generates a torque substantially equal to the torque command τ ^* by the torque control system, and drives the wheels 503 via a gear (not shown).

[Embodiment 11]
Furthermore, the present embodiment shown in FIG. 32 applies the torque control system of Embodiments 1 to 6 described above to a rear-wheel drive electric vehicle. In the present embodiment, the DC voltage of the secondary battery 611 is input to the PWM inverter 38. The control unit (CU) 600 inputs a torque command τ ^* corresponding to the depression amount of an accelerator pedal (not shown) to the torque control system of the first to sixth embodiments. The PM motor 33 generates a torque substantially equal to the torque command τ ^* by the torque control system, and the right rear wheel 609 via the motor shaft 601, the clutch 602, the clutch output shaft 603, the differential gear 604, and the rear wheel axle 605. , The left rear wheel 610 is driven.

1,200,300,400 Voltage command generator 2,3,17,102,103,117,255,262,263,264,284,410,425 Subtractor 4,80 d-axis magnetic flux deviation calculator 5,10 Integration calculators 6, 11, 19, 55, 62, 83, 95, 119, 120, 121, 122, 254 Adders 7, 8, 12, 13, 31, 104, 107, 108, 109, 112, 113, 114, 115, 310, 311 Gain 9, 90 q-axis magnetic flux deviation calculator 14, 98 q-axis magnetic flux calculator 15, 99 d-axis magnetic flux calculator 16, 18, 54, 57, 61, 64, 82, 94, 88 , 97,116,118,281,283 multiplier 30 torque command generating unit 32 i _d ^* generation unit 33 PM motor 34 the position detector 35 electrical angle calculation unit 36 electrical angular velocity calculating section 37 dq seat Inverse converter 38 PWM inverter 39 U-phase current detector 40 W-phase current detector 41 dq coordinate converters 50, 84, 500 Arithmetic blocks 51, 52, 56, 60, 63, 70, 71, 81, 85, 86, 91, 92, 96, 210, 211 Function calculation units 53, 72, 87, 93, 212 Selection means 100 Voltage command generation unit 105, 110 of prior art 3 Integrator 201, 202 Primary delay filter 203, 223 q-axis simulated magnetic flux Calculation unit 204, 224 d-axis simulated magnetic flux calculation unit 230 q-axis current transient response waveform 231 of prior art 3 q-axis current transient response waveform 232 of prior art 2 q-axis current transient response waveform 233 q of embodiment 4 Axis current transient response waveform 234 Prior art 3 d axis current transient response waveform 235 Prior art 2 d axis current transient response waveform 236 Embodiment 2 d axis current Transient response waveform 237 Embodiment 4 d-axis current of the transient response waveform 250 rotor in a static state of the i _d the d-axis direction v _d ^* i _q related to the q-axis direction in the block 251 the rotor stationary state representing the transfer characteristic for Block 252 expressing the transfer characteristic for v _q ^* 252 Φ _q ′ ω ₁ computing unit 253 Φ _d ′ ω ₁ computing unit 256 250 reverse transfer characteristic block 257 251 reverse transfer characteristic block 258 Φ _q ′ calculator 259 [Phi _d 'calculator calculates the omega ₁ 260, 261 gain ωacr with integrator 265,285,286 adder 266 motor model 267 motor inverse model 268 cascade vector controller for calculating a omega ₁ 280, 282 , 414 Integrator 411 Speed controller 412 Axis error estimator 413 PLL controller 415 Speed command generator 420 Ideal Motor 421 U-phase current detector 422 V-phase current detector 423 W-phase current detector 424, 426 Oscilloscope 427 AC current source 501 Pantograph 502 Power receiving filter 502a Filter reactor 502b Filter capacitor 503 Wheel 504 Rail 505 Main controller 510 Overhead 600 Control Unit 601 Motor shaft 602 Clutch 603 Clutch output shaft 604 Differential gear 605 Rear wheel axle 606 Front wheel axle 607 Right front wheel 608 Left front wheel 609 Right rear wheel 610 Left rear wheel 611 Secondary battery i _d ^* d-axis current command value i _q ^* q axis current value i _d d-axis current detection value i _q q-axis current detection value omega ₁ electrical angular [Phi _d d-axis magnetic flux [Phi _q q-axis magnetic flux .PHI.i d _d-axis current magnetic flux .DELTA.i d _d-axis current deviation .DELTA.i q _q-axis current deviation ΔΦ _{d d-axis} magnetic flux deviation ΔΦ _{q q-axis} magnetic flux deviation v _d ^* Voltage decoupling control before d-axis voltage command value v _q ^'* voltage decoupling control before the q-axis voltage command value v _d ^* d-axis voltage command value v _q ^* q-axis voltage command value ωacr current control response angular frequency R Armature winding resistance τ of PM motor 33 ^* Torque command value kt Torque constant V _u ^* U phase voltage command V _v ^* V phase voltage command V _w ^* W phase voltage command i _u U phase current i _v V phase Current i _w W phase current θ _c rotor electrical angle i _d ′ d axis current simulation value i _q ′ q axis current simulation value Φ _d ′ d axis simulation magnetic flux Φ _q ′ q axis simulation magnetic flux φ _m permanent magnet magnetic flux i _d ^** second d-axis current command value i _q ^** second q-axis current command value Δθ axis error estimated value ω _e speed deviation ω ₁ ^* speed command

Claims (3)

In an AC motor control device that controls the output voltage of a power converter that drives an AC motor,
The control device uses a current command value defined on the same axis as a voltage command value to the power converter defined on one of two control axes orthogonal to each other in the rotational coordinate system of the motor. And the deviation between the current detection value and the deviation between the current command value defined on the other axis and the current detection value.
The voltage command value to the power converter defined on the one axis is defined on the same axis with respect to the deviation between the current command value and the current detection value defined on the same axis. Deviation between the current detection value and a value obtained by multiplying a function having at least one of the current detection value defined on the other axis as a variable, and the current command value defined on the other axis and the current detection value Is generated based on a sum of a current detection value defined on the same axis and a value obtained by multiplying a function having at least one of the current detection values defined on the other axis as a variable. An AC motor control device characterized by the above. In claim 1,
The function defines a coil flux linkage function of the motor defined on one of two control axes orthogonal to each other in the rotational coordinate system of the motor on the same axis or the other axis. AC motor control device characterized in that it is a function that is partially differentiated by a current that is generated. In one of claim 1 or claim 2,
A vehicle drive device comprising a control device for the AC motor.

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