JP5012229B2 - Motor drive device - Google Patents

Description

  The present invention relates to a motor drive device, and more particularly to a motor control means by V / f control of a permanent magnet motor.

Conventionally, this type of motor drive device detects the inverter circuit DC current using a shunt resistor, estimates the motor effective current Iδ from the DC current, and performs V / f control so that the motor current becomes a predetermined value. (For example, see Patent Document 1).
JP 2005-218273 A

  However, the conventional motor drive device uses a filter circuit and an integration circuit (average circuit) to detect the DC average current from the shunt resistance voltage that changes according to the switching state of the inverter circuit, so that the circuit becomes complicated and the current detection accuracy When the value is increased, there is a problem that current detection responsiveness deteriorates. Furthermore, because the V / f control of the permanent magnet motor is a sensorless control method that does not calculate and estimate the rotor position, turbulence is likely to occur. If control responsiveness is poor, turbulence is likely to occur. Therefore, detection accuracy and control responsiveness are improved. There was a trade-off issue.

  Further, since the conventional method is a method in which the motor effective current Iδ is estimated and calculated to change the drive frequency, there is a problem that when the load torque increases, the rotational speed decreases and the target rotational speed cannot be controlled.

  The present invention solves the above-mentioned conventional problems, detects a DC peak current corresponding to the motor peak current, and controls the inverter circuit output voltage according to the DC peak current. Highly efficient because the inverter circuit output voltage is controlled so that the motor current phase is almost in phase with the q-axis according to the load torque. Driving is possible.

In order to solve the above-described conventional problems, a motor driving device according to the present invention includes a DC power source, an inverter circuit that converts DC power of the DC power source into AC power, and a rotor that is driven by the inverter circuit and includes a permanent magnet. The inverter circuit is controlled by a permanent magnet synchronous motor, a load driven by the permanent magnet synchronous motor, current detection means for detecting a DC peak current of the DC power supply, and an output signal of the current detection means. Control means for driving the permanent magnet synchronous motor with a sine wave at a predetermined frequency. The control means includes frequency setting means for setting an inverter output frequency, and voltage control means for generating a voltage substantially proportional to the inverter output frequency. , The inverter according to the DC peak current signal detected by the current detecting means and the inverter output frequency. Voltage correction means for correcting the output voltage of the digital circuit, phase generation means for integrating the frequency signal of the frequency setting means to generate a phase signal, and frequency correction means for correcting the frequency signal in proportion to the DC peak current signal And inverter control means for generating a three-phase sine wave signal from the voltage control signal of the voltage control means and applying a PWM control signal to the inverter circuit, and the motor induced voltage phase and the motor current phase are substantially in phase. Thus, the inverter circuit output voltage is controlled .

  The motor driving device of the present invention detects the peak value of the inverter circuit DC current and controls the inverter circuit output voltage so that the motor current phase is substantially in phase with the motor induced voltage phase. Sinusoidal drive enables simple control program, and it can be configured with an inexpensive processor and current detection means that do not require high-speed computation, and direct response to DC peak current equal to motor peak current provides excellent control response performance and high reliability A motor drive device can be realized. Further, since the motor current phase is substantially in phase with the motor induced voltage phase, the motor can be operated with high efficiency, and stable operation is always possible even when the load torque varies.

1st invention is a permanent magnet synchronous motor comprised from the direct current power supply, the inverter circuit which converts the direct current power of the direct current power supply into alternating current power, the rotor which is driven by the inverter circuit and consists of a permanent magnet, and the permanent magnet A load driven by a synchronous motor, current detection means for detecting a DC peak current of the DC power source, and the inverter circuit is controlled by an output signal of the current detection means to control the permanent magnet synchronous motor at a predetermined frequency. The control means comprises a frequency control means for setting an inverter output frequency, a voltage control means for generating a voltage substantially proportional to the inverter output frequency, and a DC peak current detected by the current detection means. Voltage correcting means for correcting the inverter circuit output voltage according to the signal and the inverter output frequency, Phase generating means for integrating the frequency signal of the frequency setting means to generate a phase signal, frequency correcting means for correcting the frequency signal in proportion to the DC peak current signal, and three-phase from the voltage control signal of the voltage control means It is composed of inverter control means that generates a sine wave signal and applies a PWM control signal to the inverter circuit, and controls the inverter circuit output voltage so that the motor induced voltage phase and the motor current phase are substantially in phase. Sensorless sine wave drive is possible with a simple current detection means with one shunt resistor, the processor and motor control program are simplified, and an inexpensive and highly reliable motor drive device with excellent control response can be realized. .

Further, the motor current phase and the motor induced voltage phase can be controlled to be substantially in phase, and the motor current and the inverter circuit current can be reduced to enable high-efficiency operation .

(Embodiment 1)
FIG. 1 is a block diagram of a motor drive device according to the first embodiment of the present invention.

  In FIG. 1, a DC power supply 2 is configured by applying AC power from an AC power supply 1 to a DC power supply circuit composed of a rectifier circuit, and DC power is converted into three-phase AC power by a three-phase full bridge inverter circuit 3 from a permanent magnet. The motor 4 composed of the constructed rotor is driven. In the DC power supply 2, capacitors 21a and 21b are connected in series to the DC output terminal of the full-wave rectifier circuit 20, and the connection point of the capacitors 21a and 21b is connected to one terminal of the AC power supply input to constitute a voltage doubler rectifier circuit. The voltage applied to the inverter circuit 3 is increased to reduce the current, thereby reducing the inverter circuit loss. The motor 4 drives a motor load 5 such as a fan or a pump. The current detection means 6 is connected to the negative voltage side of the inverter circuit 3, and the current corresponding to the output current of the inverter circuit 3, that is, the peak current of the motor 4 or the drive corresponding to the rotating magnetic field is detected by detecting the current flowing through the inverter circuit 3. Detect current.

  The current detection means 6 is a so-called one-shunt method, and includes a shunt resistor 60 connected to the emitter terminal side of the lower arm transistor of the inverter circuit 3, an amplifier circuit for detecting a peak current flowing through the shunt resistor 60, and a peak hold. The current detection circuit 61 is composed of a circuit. A method in which the peak hold circuit is configured inside the processor and the maximum value is compared by software inside the processor is also possible.

  In the 1 shunt method, when the carrier frequency is high or the modulation degree becomes large, a current undetectable region appears. Therefore, the 3 shunt method is more suitable for detecting an instantaneous current corresponding to each phase. Although excellent, in the present invention, since the current corresponding to the peak value of the motor sine wave current is detected, the circuit configuration of the single shunt method is simpler.

  The control means 7 detects the peak value of the direct current corresponding to the peak current of the motor 4 and controls the output voltage and output frequency of the inverter circuit 3 according to the direct current peak current, and sets the inverter circuit output frequency. Frequency setting means 70, voltage control means 71 for generating a voltage substantially proportional to the inverter output frequency, and voltage correction means 72 for correcting the inverter circuit output voltage in accordance with the DC peak current signal ip detected by the current detection means 6. A phase generation means 73 for integrating the frequency signal ω of the frequency setting means 70 to generate a phase signal θ, a frequency correction means 74 for correcting the frequency signal ω in proportion to the DC current peak current signal ip, and voltage control An inverter control unit that generates a three-phase sine wave signal from the voltage control signal Vδ of the means 71 and applies a PWM control signal to the inverter circuit 3. Composed of 75.

  The output signal Vδ of the voltage control means 71 is a value obtained by multiplying the inverter output frequency ω by the induced voltage constant Ke of the motor, that is, a value corresponding to the motor induced voltage Em and the voltage correction signal ΔVδ. More demanded.

  Here, Vs is a starting voltage, which is determined according to the starting torque and moment of inertia of the load. The sine wave PWM control unit 75 generates a PWM control signal by generating a three-phase sine wave signal from the phase signal θ from the phase generation unit 73 and the output signal Vδ of the voltage control unit 71, and is obtained from Equation 2. .

  The voltage correction means 72 controls the inverter circuit output voltage according to the motor current peak value Ip, so that the motor current phase is substantially in phase with the q axis, that is, the motor induced voltage phase and the motor current phase are substantially in phase. Correct the voltage. A voltage correction signal ΔVδ is generated by a predetermined function calculation corresponding to the inverter output frequency ω and the DC current peak value ip, or by a lookup table. If the current peak value Ip, the motor coil inductance L, the coil resistance R, the motor induced voltage constant Ke, and the drive frequency ω are known, the phase can be almost exactly in phase with the q axis. Since the coil resistance R is negligible at high rotational speeds, the coil inductance L and the induced voltage constant Ke are motor parameters, and there is an advantage that fewer control parameters are required as compared with the sensorless driving method for position estimation.

  FIG. 2 is a control vector diagram of the surface magnet type synchronous motor (SPMSM) according to the present invention. The motor induced voltage vector Em, the motor applied voltage vector Va, the motor current vector I, the motor coil voltage vector ωLI, and the motor magnet The relationship between axis dq coordinates and motor applied voltage γ-δ coordinates is shown. During high speed rotation, the coil resistance is very small compared to the motor coil impedance ωL, so the voltage vector due to the coil resistance can be ignored. Iδ and Iγ are currents obtained by vector decomposition into γ-δ coordinates, and Iq is a value obtained by vector decomposition into dq coordinates. Since Id is almost zero, it is not displayed. The motor applied voltage coordinate (γ-δ coordinate) is advanced by a load angle (internal phase difference angle) δ from the dq coordinate, the motor applied voltage Va is equal to the δ-axis voltage, and only the δ-axis is controlled, so Va = Vδ , Vγ = 0, so that the reverse coordinate transformation is unnecessary. The motor applied voltage Va is set so that the motor induced voltage Em is on the q axis and the motor current I is substantially equal to the q axis current Iq at the rated load. That is, the applied voltage Va is set so that the induced voltage vector Em and the coil voltage vector ωLI are substantially perpendicular. Therefore, the motor peak current Ip is substantially equal to the q-axis current Iq. In FIG. 2, the motor current vector I is displayed with a phase β delay from the q axis. The phase (power factor angle) between the motor applied voltage Va and the current I is indicated by φ.

  When voltage control is performed on the permanent magnet motor 4, turbulence occurs and control stability is poor. Therefore, by adding frequency control, stability can be improved and turbulence can be suppressed. As shown in FIG. 1, the frequency correction means 74 feeds back a signal proportional to the peak current signal ip to the inverter output signal ω, and is composed of a proportional portion 74a and a subtracting portion 74b. Given.

  FIG. 3 is a time chart showing a startup control method of the inverter output voltage and frequency and frequency correction gain.

  So-called V / f control is performed in which the output voltage Vδ and the set frequency ω are linearly increased from the start of operation to the target rotational speed, and the frequency correction gain Kf is also increased in proportion to the frequency ω. By changing the frequency correction gain Kf according to the frequency ω, it is possible to reduce fluctuations in the motor rotation speed at the time of starting low speed, and to increase the starting current by making the motor current close to a sine wave. In the figure, Kfa indicates a case where the frequency is linearly increased in proportion to the frequency ω, and Kfb is linearly changed to the frequency correction gain Kf in a steady state after reaching a predetermined frequency ωb (not shown). Is shown.

  The startup time ts until reaching the target rotational speed can be reduced according to the load torque and the moment of inertia, thereby reducing turbulence. That is, as the moment of inertia increases, the turbulence can be reduced by increasing the startup time ts.

  FIG. 3 shows an embodiment in which the frequency correction gain is controlled according to the set frequency ω, but the voltage correction gain may be controlled according to the set frequency ω. However, as will be described in detail later, the correction voltage ΔVδ is controlled in proportion to the induced voltage, so that it is substantially corrected in proportion to the set frequency ω.

  FIG. 4 shows a PWM signal and a shunt resistance voltage waveform during two-phase modulation.

  In FIG. 4, vc is a triangular wave carrier signal, vu and vv are u-phase and v-phase modulation signals, up, vp and wp are upper arm control signals for each UVW phase, and Vsh is a shunt resistance voltage waveform. Since the w-phase lower arm transistor is forced to conduct, the w-phase modulation signal is not shown.

  As shown in FIG. 4, the pattern in which the motor peak current appears in the two-phase modulation is a section where only the upper arm of one phase is turned on (t0 to t2, t4 to t5), or the upper arm of the two phases is turned on. Appear in a certain section (t5 to t7). Unlike the three-phase modulation, the two-phase modulation is PWM-controlled only for the two phases, so that the section where the peak current appears is widened, so that the peak current can be easily detected.

  FIG. 5 shows a phase in which two modulation signal waveforms of each phase of UVW and each phase current appear in the shunt resistor. The interval from 0 to 1 / 3π is the W phase current Iw and the V phase current Iv, the interval from 1 / 3π to 2 / 3π is the U phase current Iu and the V phase current Iv, and the interval from 2 / 3π to π is the U phase. A phase current Iu, a W-phase current Iw, and each phase current appear sequentially. As shown by the arrows in the figure, the section where the current peak value appears is delayed by 30 degrees from the phase at which the voltage from the neutral point of each phase reaches its peak. The peak value of each phase current appears. That is, the interval 0 to 1 / 3π is the peak value of Iw, the interval 1 / 3π to 2 / 3π is the peak value of Iv, the interval 2 / 3π to π is the peak value of Iu, and the peak value is 6 times in one period. Appears. When the current phase is delayed by 30 degrees from the voltage phase, it is easy to detect the peak current. However, when the current phase is delayed by 60 degrees, the pulse width is narrowed, indicating that current detection becomes difficult. However, in the case of IPMSM, the power factor angle φ between the voltage phase and the current phase is small, so that it is easy to detect the current peak value. In the case of SPMSM, the degree of advance is slight and almost the induced voltage phase. When the power factor angle φ is larger than 30 degrees, it is very rare and practically no problem occurs.

  The one-shunt current detection method and the method constituted by the voltage amplifier and the peak hold circuit have the feature that not only the hardware configuration is simple, but also the processor software is light and simple. Also, the A / D conversion timing for detecting the current may be the valley or peak (t0, t3, t6 in FIG. 4) of the carrier signal in which all the switching transistors of the inverter circuit are on or off, and the current detection is simple. In addition, it has a strong feature against noise.

  The waveform at the time of the two-phase modulation has been described above. The two-phase modulation is basically the same in the three-phase modulation except that the pulse width at which the current peak value is widened.

  As described above, in the first embodiment, the DC current peak value corresponding to the peak value of the motor current is detected, and the inverter circuit output voltage is corrected and controlled according to the DC current peak value. In addition, since it can be controlled without reverse conversion and without high-speed A / D conversion means or high-speed calculation means, it can be controlled by an inexpensive processor and simple control program, or motor drive that can be driven sensorless sine wave with a dedicated IC A device can be realized. In addition, a DC peak current corresponding to the motor peak current may be detected by a simple and inexpensive current sensor such as a single shunt method, and the vector sum of the motor induced voltage and the motor coil voltage according to the peak current is the inverter output voltage. The motor current phase and the induced voltage phase can be made almost equal by controlling the triangle so that it is almost a right triangle with the vector, so that the q-axis current tracking operation is possible, the maximum efficiency operation is possible according to the load torque, and the load torque The motor current automatically becomes an optimum value according to the above.

  As is apparent from the vector diagram of FIG. 2, the motor current phase β from the q axis can be controlled by adjusting the motor applied voltage Va. When the motor applied voltage Va is set small, the lead angle control is performed, and when it is set large, the delay angle is set. If the delay angle control is performed, the control becomes stable, but the voltage is likely to be saturated. Therefore, the lead angle control is generally used for high speed operation. If the advance angle control is performed, the turbulence tends to occur and becomes unstable. Therefore, the stabilization control is performed by frequency feedback.

  Further, the motor drive system according to the present invention is very simple and can be realized by a dedicated IC without using a processor. Since the structure is simple, the chip size can be reduced, so that it can be integrated with the power semiconductor, that is, it can be made into one chip. Therefore, it is easy to realize a motor control 1-chip intelligent power module (IPM), A sinusoidal drive permanent magnet motor without a position sensor, which has been necessary in the past, can be easily realized by incorporating it in the motor.

(Embodiment 2)
Hereinafter, a second embodiment of the present invention will be described with reference to a control block diagram shown in FIG.

  FIG. 6 shows a block diagram of the control means of the motor drive apparatus in Embodiment 2 of the present invention.

  The block diagram of the control means shown in FIG. 6 is a part of the control means 7 shown in FIG.

  The voltage control means 71 includes a V / f control section 71a and a voltage addition section 71b. The V / f control section 71a has a voltage proportional to the output signal ω of the frequency setting means 70, that is, an induced voltage constant Ke for ω. The multiplied voltage Vf is output to make the inverter circuit output voltage ratio to the drive frequency constant. The voltage adder 71 b adds the correction voltage ΔVδ and the starting voltage Vs (not shown) to the voltage Vf to output the voltage Vδ, and adds the voltage signal Vδ to the inverter control means 75. The output signal ip of the current detection means 6 is applied to the voltage correction means 72. The voltage correction means 72 obtains the voltage coefficient Ki corresponding to the current peak signal ip by the current function unit 72a, and multiplies the voltage Vf obtained by multiplying the inverter frequency ω by the induced voltage constant Ke and the voltage correction coefficient Ki via the low-pass filter 72b. In addition to the device 72c, a voltage correction signal ΔVδ is generated. The low-pass filter 72b suppresses fluctuations in the current peak value, and the cut-off frequency is preferably about 50 to 100 Hz.

  When the induced voltage vector Em and the motor coil voltage vector ωLI are perpendicular, the inverter output voltage Va can be obtained from Equation 4.

  Here, L is a coil inductance, Ke is an induced voltage constant, I is a motor current, and is equal to the peak current Ip. An approximate expression of Expression 4 is expressed by Expression 5.

  From Equation 5, it can be seen that the correction voltage ΔVδ may be obtained by multiplying the induced voltage Em (Ke · ω) by the voltage correction coefficient Ki. Further, the voltage correction coefficient Ki is almost equal to a value obtained by multiplying the square of the current I by the current coefficient ki, and it can be seen that the voltage correction coefficient Ki is a function corresponding to the current Ip.

  FIG. 7 shows the relationship between the detected current and the voltage correction coefficient according to the present invention, and shows the relationship between the induced voltage for equalizing the q-axis phase, the inverter output voltage ratio (Va / Em), and the voltage correction coefficient Ki. Here, the alternate long and short dash line indicates a value with a voltage ratio of 1. A value obtained by subtracting 1 from the voltage ratio (Va / Em) indicates the voltage correction coefficient Ki.

  When the motor is changed, it is only necessary to change the induced voltage constant Ke and the current function unit 72a, so that tuning becomes easy. If the current function unit 72a is configured by a calculation unit, only the current constant ki needs to be changed. In order to reduce the calculation, a lookup table is used, and the motor can be changed by simply changing the table. Compared to the conventional sensorless sine wave drive method, almost no computation is required, so that it can be realized by a simple 8-bit or 16-bit microcomputer or a dedicated IC. In the case of a dedicated IC, the induced voltage constant Ke and the current constant ki can be changed by a voltage signal of an external terminal, a resistance, or the like, so that the motor can be changed.

  The output signal ω of the frequency setting means 70 and the output signal ip of the current detection means 6 are applied to the frequency correction means 74. The frequency correction means 74 includes a proportional unit 74a for calculating a signal proportional to the signal ip, a subtracting unit 74b for the frequency setting signal ω, a frequency proportional calculating unit 74c for the frequency setting signal ω, and an output signal (Kf · ip) of the proportional unit 74a. ) And the output signal (K · ω) of the frequency proportional calculation unit 74c, the signal Δω0 from the multiplication unit 74d is added to the subtraction unit 74b via the frequency limiting unit 74e. The proportional constant Kf of the proportional unit 74a is set to about 1 to 10, and the proportional constant K of the frequency proportional calculation unit 74c is 1 in which the output signal Δω0 from the multiplication unit 74d is almost zero at startup and becomes Kf · ip at steady state. Choose a smaller value. The output signal ω1 of the frequency correction unit 74 is applied to the phase signal generation unit 73, and the phase signal θ generates the three-phase sine wave signals vu, vv, vw in addition to the sine wave generation unit 75a of the inverter control unit 75, and PWM control. Three-phase PWM signals up, un, vp, vn, wp, wn are generated via means 75b. As shown in FIG. 5, the PWM control means 75b includes a carrier signal generation unit, a signal comparison unit, a dead time insertion unit (none of which are shown), and the details are omitted.

  As described above in the second embodiment, the voltage correction unit 72 is provided with the voltage multiplier 72c, and the frequency correction unit 74 is provided with the frequency proportional calculation unit 74c. Since it is a sine wave, there is no waveform distortion, and since the rate of fluctuation due to frequency control is small during high-speed rotation, only irregularity can be prevented.

  As described above, according to the present invention, in the V / f control in which the inverter output voltage that is substantially proportional to the driving frequency of the permanent magnet motor is applied, the peak value of the direct current corresponding to the motor peak current is detected and the direct current is detected. The inverter output voltage is controlled by applying a correction voltage according to the peak current, and is a simple current detection means by a single shunt method and is driven by a sine wave without coordinate conversion and coordinate reverse conversion. With a bit processor and a simple control program, sensorless sine wave drive is possible, and a low-cost, low-noise, high-efficiency, high-reliability motor drive device can be realized with a small number of parts.

  Furthermore, since it has excellent current detection accuracy and detection response, and also has excellent control response, it does not step out due to load fluctuations and operates stably from no load to rated load. Further, since the motor current phase is controlled to be substantially in phase with the q axis, the maximum efficiency operation can always be performed by the q axis tracking operation according to the load torque, and the loss of the motor and the inverter circuit can be reduced. In addition, the sensorless sine wave drive based on V / f control without estimating the rotor position is advantageous in that there are few motor parameters and control parameters, excellent robustness, and almost no tuning man-hours. In particular, when a plurality of motors are driven simultaneously by a single processor, since the motor control program and current detection are simplified, the burden on the processor is lightened. It can be applied to the simultaneous sine wave drive system, and an inexpensive and highly reliable multiple motor simultaneous drive device can be realized.

As described above, the motor drive device of the present invention is a motor drive device of the present invention, in which a permanent magnet motor is driven by a sensorless sine wave by an inverter circuit that converts DC power into AC power, and the peak value of the motor current or the rotating magnetic field The inverter circuit output voltage and the motor drive frequency are controlled so that the motor current corresponding to is set to a set value, and thus can be applied to most motor drive devices that drive permanent magnet motors. The present invention can be applied to a motor driving device for a washing pump, a motor driving device for a washing machine, a motor driving device for a vacuum cleaner, a fan motor driving device such as a ventilation fan or a combustor, and a heat pump motor driving device for an air conditioner or a refrigerator. Furthermore, the present invention can also be applied to a multiple motor simultaneous drive system such as a heat pump washer / dryer or an air conditioner.

  Further, the sensorless sine wave driving method according to the present invention is very simple and can be realized by a dedicated integrated circuit without a processor, and can be embodied as a power module for sine wave driving in which a power semiconductor and a control IC are integrated. . If a power module for sine wave driving is realized, it becomes easy to mount inside the motor, and if a direct current is applied, it becomes easy to downsize the permanent magnet motor module driven by sine wave.

1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. Motor control vector diagram of the motor drive device Time chart showing start-up control of the motor drive device Shunt resistance voltage waveform and current detection timing diagram of the motor drive device Current detection timing chart during two-phase modulation of the motor drive device The block diagram of the control means of the motor drive unit in Embodiment 2 of this invention Relationship diagram between current and voltage correction coefficient of control means of the motor drive device

2 DC power supply 3 Inverter circuit 4 Motor 5 Motor load 6 Current detection means 7 Control means 70 Frequency setting means 71 Voltage control means 72 Voltage correction means 74 Frequency correction means

Claims (1)

A DC power source, an inverter circuit that converts DC power of the DC power source into AC power, a permanent magnet synchronous motor that is driven by the inverter circuit and is made of a permanent magnet, and driven by the permanent magnet synchronous motor A load, current detection means for detecting a DC peak current of the DC power supply, and control means for controlling the inverter circuit by an output signal of the current detection means to drive the permanent magnet synchronous motor with a sine wave at a predetermined frequency. The control means includes frequency setting means for setting an inverter output frequency, voltage control means for generating a voltage substantially proportional to the inverter output frequency, a DC peak current signal detected by the current detection means, and the inverter output frequency. Voltage correction means for correcting the inverter circuit output voltage in accordance with the frequency setting means A phase generation means for generating a phase signal by integrating the frequency signal, a frequency correction means for correcting the frequency signal in proportion to the DC peak current signal, and a three-phase sine wave signal from the voltage control signal of the voltage control means And an inverter control means for applying a PWM control signal to the inverter circuit and controlling the inverter circuit output voltage so that the motor induced voltage phase and the motor current phase are substantially in phase .

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