JP4983457B2 - Motor drive device - Google Patents

Description

  The present invention relates to a motor drive device, and more particularly to a motor control means by V / f control of a permanent magnet motor.

Conventionally, this type of motor drive device detects the inverter circuit DC current using a shunt resistor, estimates the motor effective current Iδ from the DC current, and performs V / f control so that the motor current becomes a predetermined value. (For example, see Patent Document 1).
JP 2005-218273 A

  However, the conventional motor drive device uses a filter circuit and an integration circuit (average circuit) to detect the DC average current from the shunt resistance voltage that changes according to the switching state of the inverter circuit, so that the circuit becomes complicated and the current detection accuracy When the value is increased, there is a problem that current detection responsiveness deteriorates. Furthermore, because the V / f control of the permanent magnet motor is a sensorless control method that does not calculate and estimate the rotor position, turbulence is likely to occur. If control responsiveness is poor, turbulence is likely to occur. Therefore, detection accuracy and control responsiveness are improved. There was a trade-off issue.

  Further, since the conventional method is a method in which the motor effective current Iδ is estimated and calculated to change the drive frequency, there is a problem that when the load torque increases, the rotational speed decreases and the target rotational speed cannot be controlled.

  The present invention solves the above-mentioned conventional problems, detects a DC peak current corresponding to the motor peak current, and controls the inverter circuit output voltage according to the DC peak current. The inverter circuit output voltage is controlled so that the motor current phase is almost in phase with the q-axis according to the load torque, because the current detection accuracy and detection response are excellent. Efficient operation is possible.

In order to solve the above-described conventional problems, a motor driving device according to the present invention includes a DC power source, an inverter circuit that converts DC power of the DC power source into AC power, and a rotor that is driven by the inverter circuit and includes a permanent magnet. A permanent magnet synchronous motor configured; a load driven by the permanent magnet synchronous motor; current detection means for detecting a DC peak current of the DC power supply; frequency setting means for setting the inverter circuit output frequency; A current change for detecting a change in the output signal of the current detection means, comprising a control means for controlling the output voltage of the inverter circuit by an output signal of the frequency setting means and the current detection means to drive the motor in a sine wave at a predetermined frequency. The detection circuit corrects the inverter circuit output frequency .

  The motor drive device of the present invention detects the peak value of the inverter circuit DC current and the change in the current, and controls the inverter circuit output voltage and output frequency so that the motor current phase is substantially in phase with the motor induced voltage phase. Yes, sensorless sine wave drive without coordinate conversion simplifies the control program, can be configured with an inexpensive processor and current detection means that do not require high-speed computation, and control response performance because it directly detects DC peak current equal to motor peak current Excellent and highly reliable motor driving device can be realized. Furthermore, since the motor current phase is almost in phase with the motor induced voltage phase, high-efficiency operation of the motor is possible, and the inverter frequency is corrected by the current change signal. Is possible.

1st invention is a permanent magnet synchronous motor comprised from the direct current power supply, the inverter circuit which converts the direct current power of the direct current power supply into alternating current power, the rotor which is driven by the inverter circuit and consists of a permanent magnet, and the permanent magnet By a load driven by a synchronous motor, current detection means for detecting a DC peak current of the DC power supply, frequency setting means for setting the inverter circuit output frequency , output signals of the frequency setting means and the current detection means It comprises control means for controlling the inverter circuit output voltage to drive the motor with a sine wave at a predetermined frequency, and corrects the inverter circuit output frequency by current change detection means for detecting a change in the output signal of the current detection means. The sensorless sine wave drive is possible with a simple current detection means with one shunt resistor. , And the processor and the motor control program is simplified and can achieve high motor driving device reliable inexpensive high-efficiency engine speed fluctuation is excellent in less control responsiveness and stability.

According to a second aspect of the invention, the control means in the first aspect of the invention is a current change detection for detecting a change in the output signal of the frequency setting means for setting the inverter circuit output frequency and the current detection means for detecting the DC peak current of the DC power supply And a frequency correcting means for correcting the output frequency of the inverter circuit in accordance with an output signal of the current change detecting means, wherein the current change detecting means is constituted by a high pass filter for cutting a direct current or low frequency signal, and the current is obtained by so as to control and stabilization control of the inverter circuit output frequency according to the change detection signal, the phase difference of b over motor magnet axis and the inverter circuit output voltage axis, i.e., the internal phase angle corresponding to the load torque The current detection means output signal changes in the same way. Stable rotation can be obtained even in the high load click fluctuation suppressing hunting, it can be realized with less motor driving apparatus of the engine speed fluctuation.

In addition, the current change signal that is an AC component can be detected by a high-pass filter, and the internal phase difference angle can be instantaneously controlled according to the load torque by changing the inverter output frequency based on the current change signal. Stabilization control is also possible.

(Embodiment 1)
FIG. 1 is a block diagram of a motor drive device according to the first embodiment of the present invention.

  In FIG. 1, a DC power supply 2 is configured by applying AC power from an AC power supply 1 to a DC power supply circuit composed of a rectifier circuit, and DC power is converted into three-phase AC power by a three-phase full bridge inverter circuit 3 from a permanent magnet. The motor 4 composed of the constructed rotor is driven. In the DC power supply 2, capacitors 21a and 21b are connected in series to the DC output terminal of the full-wave rectifier circuit 20, and the connection point of the capacitors 21a and 21b is connected to one terminal of the AC power supply input to constitute a voltage doubler rectifier circuit. The voltage applied to the inverter circuit 3 is increased to reduce the current, thereby reducing the inverter circuit loss. The motor 4 drives a motor load 5 such as a fan or a pump. By connecting the current detection means 6 to the negative voltage side of the inverter circuit 3 and detecting the peak current flowing through the inverter circuit 3, it corresponds to the output current of the inverter circuit 3, that is, the peak current of the motor 4 or the rotating magnetic field. Drive current is detected.

  The current detection means 6 is a so-called one-shunt method, and includes a shunt resistor 60 connected to the emitter terminal side of the lower arm transistor of the inverter circuit 3, an amplifier circuit for detecting a peak current flowing through the shunt resistor 60, and a peak hold. The current detection circuit 61 is composed of a circuit. A method in which the peak hold circuit is configured inside the processor and the maximum value is compared by software inside the processor is also possible.

  In the 1 shunt method, when the carrier frequency is high or the modulation degree becomes large, a current undetectable region appears. Therefore, the 3 shunt method is more suitable for detecting an instantaneous current corresponding to each phase. Although excellent, in the present invention, only the current corresponding to the peak value of the motor sine wave current needs to be detected, so the circuit configuration is simpler with the one shunt method.

  The control means 7 detects the peak value of the direct current corresponding to the peak current of the motor 4 and controls the output voltage and output frequency of the inverter circuit 3 according to the direct current peak current, and sets the inverter circuit output frequency. Frequency setting means 70, voltage control means 71 for generating a voltage substantially proportional to the inverter output frequency, and voltage correction means 72 for correcting the inverter circuit output voltage in accordance with the DC peak current signal ip detected by the current detection means 6. And the current change signal Δip of the DC current peak current signal ip, that is, the current change detection means 73 for detecting the AC component signal, and the output frequency signal ω of the frequency setting means 70 by the current change signal Δip of the current change detection means 73. The frequency correction means 74 for correction and the angular frequency signal ω1 of the frequency correction means 74 are integrated to generate the phase signal θ. Phase generation means 75 and a sine wave PWM control means 76 for generating a PWM signal by generating a three-phase sine wave signal from the output signal Vδ of the voltage control means 71 and the phase signal θ of the phase generation means 75. The output signal of the sine wave PWM control means 76 is applied to the control terminal (gate signal terminal) of the power switching means (not shown) of the inverter circuit 3.

  The voltage control means 71 includes a V / f control unit 71a that generates a voltage proportional to the inverter frequency ω and a voltage addition unit 71b. The output signal of the V / f control unit 71a is applied to the voltage correction unit 72, and the voltage The correction unit 72 generates a voltage correction signal ΔVδ from the current signal ip of the current detection unit 6 and the output signal of the V / f control unit 71a, and adds the voltage correction signal ΔVδ to the voltage addition unit 71b. The output signal Vδ of the voltage control means 71 is obtained by setting the voltage correction signal ΔVδ to a value Vf (= Ke × ω) obtained by multiplying the inverter output frequency ω by the induced voltage constant Ke of the motor, that is, a value corresponding to the motor induced voltage Em. The added value is obtained from Equation 1.

  Here, Vs is a starting voltage, which is determined according to the starting torque and moment of inertia of the load. The sine wave PWM control means 76 generates a sine wave signal obtained from Equation 2 from the voltage control signal Vδ and the phase signal θ, and uses the sine wave signals vu, vv, vw obtained from Equation 2 as carrier signals (triangular wave signals, Alternatively, a PWM signal is generated in comparison with a sawtooth wave signal.

  The voltage correction means 72 controls the inverter circuit output voltage according to the motor current peak value Ip, so that the motor current phase is substantially in phase with the q axis, that is, the motor induced voltage phase and the motor current phase are substantially in phase. Correct the voltage. A voltage correction signal ΔVδ is generated by a predetermined function calculation corresponding to the inverter output frequency ω and the DC current peak value ip, or by a lookup table. If the current peak value Ip, the motor coil inductance L, the coil resistance R, the motor induced voltage constant Ke, and the drive frequency ω are known, the phase can be almost exactly in phase with the q axis. Since the coil resistance R is negligible at high rotational speeds, the coil inductance L and the induced voltage constant Ke are motor parameters, and the V / f control method according to the present invention has fewer control parameters than the sensorless drive method for position estimation. There are good features.

  FIG. 2 is a control vector diagram of the surface magnet type synchronous motor (SPMSM) according to the present invention. The motor induced voltage vector Em, the motor applied voltage vector Vδ, the motor current vector I, the motor coil voltage vector ωLI, and the magnet axis of the motor The relationship between the dq coordinate and the motor applied voltage γ-δ coordinate is shown. During high speed rotation, the coil resistance is very small compared to the motor coil impedance ωL, so the voltage vector due to the coil resistance can be ignored. Iδ and Iγ are currents obtained by vector decomposition into γ-δ coordinates, and Iq is a value obtained by vector decomposition into dq coordinates. Since Id is almost zero, it is not displayed. The motor applied voltage coordinate (γ-δ coordinate) is advanced by a load angle (internal phase difference angle) δ from the dq coordinate, the motor applied voltage Va is equal to the δ-axis voltage, and only the δ-axis is controlled, so Va = Vδ , Vγ = 0, so that the reverse coordinate transformation is unnecessary. The motor applied voltage Vδ is set so that the motor induced voltage Em is on the q axis and the motor current I is substantially equal to the q axis current Iq at the rated load. That is, the applied voltage Vδ is set so that the induced voltage vector Em and the coil voltage vector ωLI are substantially perpendicular. Therefore, the motor peak current Ip is substantially equal to the q-axis current Iq. The phase (power factor angle) φ of the motor applied voltage Va (= Vδ) and the current I is equal to the load angle δ.

  Since voltage regulation of the permanent magnet motor 4 causes turbulence and poor control stability, it has been known for some time that stability can be improved and turbulence can be suppressed by adding frequency control, which is also described in Document 1. Yes. As shown in FIG. 1, the frequency correction means 74 feeds back a signal proportional to the peak current change signal Δip to the inverter output signal ω1, and the output signal ω1 is given by Equation 3. Here, Kf is a proportionality constant.

  FIG. 3 shows a startup control method of the inverter output voltage and frequency, and the frequency correction gain.

  So-called V / f control is performed in which the output voltage Vδ and the set frequency ω are linearly increased from the start of operation to the target rotational speed, and the frequency correction gain Kf is also increased in proportion to the frequency ω. By changing the frequency correction gain Kf according to the frequency ω, it is possible to reduce fluctuations in the motor rotation speed at the time of starting low speed, and to increase the starting current by making the motor current close to a sine wave. In the figure, Kfa indicates a case where the frequency is linearly increased in proportion to the frequency ω, and Kfb is linearly changed to the frequency correction gain Kf in a steady state after reaching a predetermined frequency ωb (not shown). Is shown.

  The startup time ts until reaching the target rotational speed can be reduced according to the load torque and the moment of inertia, thereby reducing turbulence. That is, if the starting time ts is lengthened as the moment of inertia is larger, the starting torque can be reduced and turbulence can be suppressed.

  FIG. 3 shows an embodiment in which the frequency correction gain is controlled according to the set frequency ω, but the voltage correction gain may be controlled according to the set frequency ω. However, since the correction voltage ΔVδ for voltage correction is corrected in proportion to the V / f control voltage Vf (= Ke × ω), the voltage is corrected in proportion to the frequency.

  FIG. 4 shows a PWM signal and a shunt resistance voltage waveform during two-phase modulation.

  In FIG. 4, vc is a triangular wave carrier signal, vu and vv are u-phase and v-phase modulation signals, up, vp and wp are upper arm control signals for each UVW phase, and Vsh is a shunt resistance voltage waveform. Since the w-phase lower arm transistor is forced to conduct, the w-phase modulation signal is not shown.

  The pattern in which the motor peak current appears in the two-phase modulation is a section where only the upper arm of one phase is turned on (t0 to t2, t4 to t5) as shown in FIG. 4, or the upper arm of the two phases is turned on. It appears in the section (t5 to t7). Unlike the three-phase modulation, the two-phase modulation is PWM-controlled only for the two phases, so that the section where the peak current appears is widened, so that the peak current can be easily detected.

  FIG. 5 shows a phase in which two modulation signal waveforms of each phase of UVW and each phase current appear in the shunt resistor. The interval from 0 to 1 / 3π is the W phase current Iw and the V phase current Iv, the interval from 1 / 3π to 2 / 3π is the U phase current Iu and the V phase current Iv, and the interval from 2 / 3π to π is the U phase. A phase current Iu, a W-phase current Iw, and each phase current appear sequentially. As shown by the arrows in the figure, the section where the current peak value appears is delayed by 30 degrees from the phase at which the voltage from the neutral point of each phase reaches its peak. The peak value of each phase current appears. That is, the interval 0 to 1 / 3π is the peak value of Iw, the interval 1 / 3π to 2 / 3π is the peak value of Iv, the interval 2 / 3π to π is the peak value of Iu, and the peak value is 6 times in one period. Appears. When the current phase is delayed by 30 degrees from the voltage phase, it is easy to detect the peak current. However, when the current phase is delayed by 60 degrees, the pulse width is narrowed, indicating that current detection becomes difficult. However, in the case of IPMSM, the power factor angle φ between the voltage phase and the current phase is small, so that it is easy to detect the current peak value. In the case of SPMSM, the degree of advance is slight and almost the induced voltage phase. When the power factor angle φ is larger than 30 degrees, it is very rare and practically no problem occurs.

  The one-shunt current detection method and the peak-hold circuit method, which is composed of a voltage amplifier and a peak-hold circuit, as will be described later, are not only simple in hardware configuration but also simple in the processor software. There is. Also, the A / D conversion timing for detecting the current may be the valley or peak (t0, t3, t6 in FIG. 4) of the carrier signal in which all the switching transistors of the inverter circuit are on or off, and the current detection is simple. In addition, it has a strong feature against noise.

  The waveform at the time of the two-phase modulation has been described above. The two-phase modulation is basically the same in the three-phase modulation except that the pulse width at which the current peak value is widened.

  FIG. 6 shows a specific embodiment of the current detection circuit 61 that detects the peak current.

  The non-inverting input terminal of the high-speed operational amplifier 610 is connected to the shunt resistor 60, the annotated terminal of the diode 611 is connected to the output terminal of the high-speed operational amplifier 610, and the feedback resistor 612 is inverted from the cathode terminal of the diode 611. A non-inverting amplifier is configured by connecting to the input terminal and connecting a grounding resistor 613 between the inverting input terminal and the ground. A peak hold capacitor 614 is connected between the cathode terminal of the diode 611 and the ground to peak-hold the current signal. The feedback resistor 612 and the ground resistor 613 serve as a discharge resistor. The discharge time constant composed of the discharge resistor and the peak hold capacitor 614 can be set from several times to several tens of times the period of the inverter driving frequency because the peak current can be detected six times in one period of the inverter driving frequency. As the discharge time constant is shortened, the response performance becomes faster, but the high frequency component increases, and the frequency correction signal component, which will be described later, increases and the inverter drive frequency fluctuation increases. As described above, the feedback amplifier that amplifies the signal peak value can be configured easily and inexpensively, the current signal corresponding to the motor peak current can be converted into a DC voltage signal, the gain is determined by the feedback resistor 612 and the ground resistor 613, and the current The output signal peak value vip of the detection circuit 61 is not affected by the temperature characteristics of the diode. Converting the motor peak current into a DC signal by a simple peak hold circuit is the same as coordinate conversion, which means that a control method that does not require coordinate conversion calculation by a processor can be realized.

  As described above, in the first embodiment according to the present invention, the DC current peak value corresponding to the peak value of the motor current is detected, and the inverter circuit output voltage is corrected and controlled according to the DC current peak value. Since the motor current peak value can be directly detected, coordinate conversion is unnecessary, and the motor applied voltage peak value is controlled, so that coordinate reverse conversion is unnecessary.

  In addition, the current change detection means and the frequency correction means can prevent turbulence due to torque fluctuations, and the current phase can always be close to the q-axis, so that even when the load torque increases, the motor current can be reduced almost without reducing the drive frequency. Increases the optimum operation according to the load torque.

  Furthermore, since control is possible without high-speed A / D conversion means and high-speed calculation means, an inexpensive processor and a simple control program, or a motor drive device capable of sensorless sine wave drive with a dedicated IC can be realized. In addition, a DC peak current corresponding to the motor peak current may be detected by a simple and inexpensive current sensor such as a single shunt method, and the vector sum of the motor induced voltage and the motor coil voltage according to the peak current is the inverter output voltage. The motor current phase and induced voltage phase can be made approximately equal by controlling the vector so that it is almost a right triangle with the vector, so load torque current tracking operation is possible, maximum efficiency operation is possible according to load torque, and load torque Accordingly, the motor current is automatically controlled to the optimum value.

  As is apparent from the vector diagram of FIG. 2, the motor current phase β from the q axis can be controlled by adjusting the motor applied voltage Va (= Vδ). When the motor applied voltage Va is set small, the lead angle control is performed, and when it is set large, the delay angle is set. If the delay angle control is performed, the control becomes stable, but the voltage is likely to be saturated. Therefore, the lead angle control is generally used for high speed operation. If the advance angle control is performed, the turbulence tends to occur and becomes unstable. Therefore, the stabilization control is performed by frequency feedback.

  Further, the motor drive system according to the present invention is very simple and can be realized by a dedicated IC without using a processor. Since the structure is simple, the chip size can be reduced, so that it can be integrated with the power semiconductor, that is, it can be made into one chip. Therefore, it is easy to realize a motor control 1-chip intelligent power module (IPM), A sinusoidal drive permanent magnet motor without a position sensor, which has been necessary in the past, can be easily realized by incorporating it in the motor.

(Embodiment 2)
Hereinafter, a second embodiment of the present invention will be described with reference to a control block diagram shown in FIG.

  FIG. 7 shows a block diagram of the control means of the motor drive apparatus in Embodiment 2 of the present invention.

  The block diagram of the control means shown in FIG. 7 is obtained by adding detailed blocks to the block diagram of the control means 7 shown in FIG. 1, and only the change adding unit will be described below.

  The voltage control means 71 comprises a V / f control section 71a and a voltage addition section 71b. The V / f control section 71a multiplies the voltage proportional to the output signal ω of the frequency setting means 70, that is, ω by an induced voltage constant Ke. The voltage Vf is output to make the inverter circuit output voltage ratio to the drive frequency constant. The voltage adder 71 b adds the correction voltage ΔVδ and the starting voltage Vs (not shown) to the voltage Vf to output the voltage Vδ, and adds the voltage signal Vδ to the inverter control means 75. The output signal ip of the current detection means 6 is applied to the voltage correction means 72. The voltage correction means 72 obtains the voltage coefficient Ki corresponding to the current peak signal ip by the current function unit 72a, and multiplies the voltage Vf obtained by multiplying the inverter frequency ω by the induced voltage constant Ke and the voltage correction coefficient Ki via the low-pass filter 72b. In addition to the device 72c, a voltage correction signal ΔVδ is generated. The low-pass filter 72b suppresses fluctuations in the current peak value, and the cut-off frequency is preferably about 50 to 100 Hz.

  When the induced voltage vector Em and the motor coil voltage vector ωLI are perpendicular, the inverter output voltage Va can be obtained from Equation 4.

  Here, L is a coil inductance, Ke is an induced voltage constant, I is a motor current, and is equal to the peak current Ip. When the current is 1 A or less, the approximate expression of Expression 4 is expressed by Expression 5.

  From Equation 5, it can be seen that the correction voltage ΔVδ may be obtained by multiplying the induced voltage Em (= Ke · ω) by the voltage correction coefficient Ki. Further, the voltage correction coefficient Ki is almost equal to a value obtained by multiplying the square of the current I by the current coefficient ki, and it can be seen that the voltage correction coefficient Ki is a function corresponding to the current Ip.

  Since the approximate expression changes when the current is large, the approximate expression is set according to the motor and the load.

  FIG. 8 shows the relationship between the detected current and the voltage correction coefficient according to the present invention, and shows the relationship between the induced voltage, the inverter output voltage ratio (Va / Em), and the voltage correction coefficient Ki for equalizing the q-axis phase. Here, the alternate long and short dash line indicates a value with a voltage ratio of 1. A value obtained by subtracting 1 from the voltage ratio (Va / Em) indicates the voltage correction coefficient Ki.

  When the motor is changed, it is only necessary to change the induced voltage constant Ke and the current function unit 72a, so that tuning becomes easy. If the current function unit 72a is configured by a calculation unit, only the current constant ki needs to be changed. In order to reduce the calculation, a lookup table is used, and the motor can be changed by simply changing the table. Compared to the conventional sensorless sine wave drive method, almost no computation is required, so that it can be realized by a simple 8-bit or 16-bit microcomputer or a dedicated IC. In the case of a dedicated IC, the induced voltage constant Ke and the current constant ki can be changed by a voltage signal of an external terminal, a resistance, or the like, so that the motor can be changed.

  The output signal ip of the current detection circuit 6 is applied to the current change detection means 73, and the current change detection means 73 detects the change signal Δip from the average value of the current signal ip and applies it to the frequency correction means 74. When the load torque increases instantaneously, the load angle increases, the motor peak current ip increases, the current phase advances, and the current change signal Δip increases. When Δip increases, the frequency correction means 74 decreases the inverter frequency ω1, so that the motor current phase approaches the q-axis, so that the torque can be increased and stabilized. On the contrary, when the load is lightened, the current phase is delayed and the load angle is reduced, so that the current change signal Δip is lowered and there is almost no change in the inverter drive frequency. However, since the inverter correction voltage decreases and the motor applied voltage decreases due to the instantaneous decrease in the current signal ip, the motor current decreases and the motor rotates at an appropriate current delay phase. When the current phase is delayed, the stability of the synchronous motor increases and the turbulence decreases.

  FIG. 9 shows a block diagram of the current change detecting means 73, comprising an averaging means 730 for averaging the current signal ip, a comparison means for detecting an error signal between the output signal ipav of the averaging means 730 and the current signal ip, A current change signal Δip is output. The current change signal Δip is obtained by subtracting an average signal (DC signal) ipav from the current signal ip and consequently becomes equal to the AC signal component. The current change detection means shown in FIG. 9 has an advantage that it can be controlled even in the case of a load having a large moment of inertia because it is easy to detect a low frequency current change.

  The current change detection means 73 can be configured by a high-pass filter such as a capacitor that extracts an AC signal component from the current signal ip.

  FIG. 10 shows the frequency output voltage characteristics of the current change detection means constituted by a high-pass filter. The cut-off frequency fc, which is 3 dB lower than the output voltage Vo, is the natural vibration of the electromechanical system expressed by Equation 6. It may be set substantially equal to the number ωn or lower than the natural frequency ωn. This is because, in V / f control, the vibration frequency of the control system with respect to torque fluctuation is substantially equal to the natural frequency ωn of the electromechanical system.

  In Equation 6, Pf is the number of pole pairs, Ψm is the number of flux linkages, J is the moment of inertia, L is the coil inductance, and δ0 and the load angle at the operating point. It can be seen from Equation 6 that the natural frequency decreases as the moment of inertia and the coil inductance increase, so that a high-pass filter that cuts only the DC component is required. It can also be seen that when the moment of inertia is small, the natural frequency is high, so that control with excellent high-speed response is possible.

  The frequency correction means 74 corrects the inverter angular frequency using the current change signal Δip, and a proportional unit 740 that calculates a signal proportional to the signal ip, a frequency proportional calculation unit 741 of the frequency setting signal ω, and an output signal of the proportional unit 740 The signal Δω0 from the multiplication unit 742 that calculates the product of (Kf · Δip) and the output signal (K · ω) of the frequency proportional calculation unit 741 is added to the subtraction unit 744 via the frequency limiting unit 743. The proportional constant Kf of the proportional unit 740 is set to about 5 to 20, and the proportional constant K of the frequency proportional calculation unit 741 is that the output signal Δω0 from the multiplication unit 742 is almost zero at start-up, and the correction frequency is 0.5 at steady state. Choose a small value that will be around a few percent. Kf shown in FIG. 3 is equivalent to the output signal of the multiplication unit 742 shown in the block diagram of FIG.

  The output signal ω1 of the frequency correction unit 74 is applied to the phase signal generation unit 73, and the phase signal θ generates the three-phase sine wave signals vu, vv, vw in addition to the sine wave generation unit 76a of the sine wave PWM control unit 76, Three-phase PWM signals up, un, vp, vn, wp, wn are generated via the PWM control means 76b. As shown in FIG. 5, the PWM control unit 76b includes a carrier signal generation unit, a signal comparison unit, a dead time insertion unit (all not shown), and the like, but details thereof are omitted.

  As described above in the second embodiment, even when an instantaneous load fluctuation is generated by the voltage correction unit and the frequency correction unit, the motor current phase is controlled to be in phase with the q axis, so that the stabilization control is performed. It becomes possible. In particular, when the load torque increases instantaneously, the rotor rotational frequency decreases relatively, the motor applied voltage advances, the load angle increases, and the motor peak current increases as the load angle increases, so the motor peak current increases. Then, the feedback control that reduces the inverter drive frequency according to the change of the peak current prevents the increase and turbulence of the load angle, prevents the motor current phase from advancing and delaying from the q axis, and always adjusts the current phase to the q axis. So it is stabilized.

  In addition, a frequency limiter 743 is provided in the frequency correction means for correcting the drive frequency by the current change signal, and even if torque fluctuation occurs, it can be stabilized by motor applied voltage correction and frequency correction, and the fluctuation of the inverter drive frequency is minimized. A motor drive device that can be limited can be realized.

  Further, by providing a voltage multiplier 72c in the voltage correction means 72 and providing a frequency proportional calculation unit 741 in the frequency correction means 74, the PWM control signal becomes a sine wave during low speed rotation such as when the motor is started, so there is no waveform distortion. In high-speed rotation, only the turbulence can be prevented because the fluctuation rate by frequency control is small.

  Also, by providing a current limiter on the input side of the voltage correction means or a voltage correction limiter on the output side to limit the voltage correction amount due to the current change, it is possible to suppress the vibration of the control system and improve the excessive control response performance. .

  As described above, according to the present invention, in the V / f control in which the inverter output voltage that is substantially proportional to the driving frequency of the permanent magnet motor is applied, the peak value of the direct current corresponding to the motor peak current is detected and the direct current is detected. The inverter output voltage is controlled by adding a correction voltage according to the peak current, and can be driven by a sine wave without simple current detection means and coordinate conversion and coordinate reverse conversion using a single shunt method. A sensorless sine wave drive that always adjusts the current phase to the q-axis by a processor and a simple control program is possible, and a low-cost, low-noise, high-efficiency, high-reliability motor drive device can be realized with a small number of parts.

  In addition, by detecting a change in the DC current peak value or an AC component and correcting the inverter drive frequency, current phase fluctuation due to torque fluctuation can be reduced and turbulence can be prevented. In particular, the current advance angle and turbulence from the q-axis due to an increase in load torque can be prevented, and the motor applied voltage can be controlled to the optimum value according to the torque, so that the current phase is always close to the q-axis for maximum efficiency operation. Heat generation of inverter circuit power components can be prevented.

  Further, even if the load torque fluctuates, the inverter drive frequency change is small, so that a motor drive device with excellent rotational speed stability accuracy can be realized.

  Furthermore, since it has excellent current detection accuracy and detection response, and also has excellent control response, it does not step out due to load fluctuations and operates stably from no load to rated load. Further, since the motor current phase is controlled to be substantially in phase with the q axis, the maximum efficiency operation can always be performed by the q axis tracking operation according to the load torque, and the loss of the motor and the inverter circuit can be reduced. In addition, the sensorless sine wave drive based on V / f control without estimating the rotor position is advantageous in that there are few motor parameters and control parameters, excellent robustness, and almost no tuning man-hours. In particular, when a plurality of motors are driven simultaneously by a single processor, since the motor control program and current detection are simplified, the burden on the processor is lightened. It can be applied to the simultaneous sine wave drive system, and an inexpensive and highly reliable multiple motor simultaneous drive device can be realized.

  As described above, the motor drive device of the present invention detects the motor current corresponding to the peak value of the motor current or the rotating magnetic field by driving the permanent magnet motor with the inverter circuit that converts the DC power into the AC power. The inverter circuit output voltage and the motor drive frequency are controlled so that they become the set values. Therefore, it can be applied to almost all motor drive devices that drive permanent magnet motors, and spindle motors and pump motors with little rotational speed fluctuations. Suitable for fan motor drive device, dishwasher washing pump drive device, washing machine motor drive device, vacuum cleaner motor drive device, fan motor drive device such as ventilation fan and combustor, air conditioner and refrigerator It can be applied to a heat pump motor drive device. Furthermore, the present invention can also be applied to a multiple motor simultaneous drive system such as a heat pump washer / dryer or an air conditioner.

  Further, the sensorless sine wave driving method according to the present invention is very simple and can be realized by a dedicated integrated circuit without a processor, and can be embodied as a power module for sine wave driving in which a power semiconductor and a control IC are integrated. . If a power module for sine wave driving is realized, it becomes easy to mount inside the motor, and if a direct current is applied, it becomes easy to downsize the permanent magnet motor module driven by sine wave.

1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. Motor control vector diagram of the motor drive device Time chart showing start-up control of the motor drive device Shunt resistance voltage waveform and current detection timing diagram of the motor drive device Current detection timing chart during two-phase modulation of the motor drive device Circuit diagram of current detection means of the motor drive device The block diagram of the control means of the motor drive unit in Embodiment 2 of this invention Relationship diagram between current and voltage correction coefficient of control means of the motor drive device Block diagram of current change detection means of the motor drive device Frequency output voltage characteristic diagram of current change detection means of the motor drive device

2 DC power supply 3 Inverter circuit 4 Motor 5 Motor load 6 Current detection means 7 Control means 70 Frequency setting means 71 Voltage control means 72 Voltage correction means 73 Current change detection means 74 Frequency correction means

Claims (2)

A DC power source, an inverter circuit that converts DC power of the DC power source into AC power, a permanent magnet synchronous motor that is driven by the inverter circuit and is made of a permanent magnet, and driven by the permanent magnet synchronous motor Load, current detection means for detecting a DC peak current of the DC power supply, frequency setting means for setting the inverter circuit output frequency, and output voltage of the inverter circuit according to output signals of the frequency setting means and the current detection means A motor drive device comprising control means for controlling and driving the motor with a sine wave at a predetermined frequency, wherein the inverter circuit output frequency is corrected by a current change detection means for detecting a change in an output signal of the current detection means . The control means includes frequency setting means for setting the inverter circuit output frequency, current change detection means for detecting a change in the output signal of the current detection means for detecting the DC peak current of the DC power supply, and an output signal of the current change detection means The current change detection means is composed of a high-pass filter that cuts a direct current or low frequency signal, and the inverter circuit output frequency according to the current change detection signal. The motor driving device according to claim 1, wherein the motor is controlled and stabilized .

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