EP2184803A1 - Coplanar differential bi-strip delay line, higher-order differential filter and filtering antenna furnished with such a line - Google Patents

Abstract

The line (30) has conducting strips (32, 34) arranged on same surface (36) of a dielectric substrate, where the line is configured in the form of printed circuit board. The board presents structural discontinuities formed by portions (32B, 32C, 34B, 34C) of the strips. The discontinuities generate impedance jump and capacitive coupling between the strips for reproducing predetermined quarter-wave phase difference. An interdigitated capacitor for capacitive coupling is formed by conductive fingers (32D, 34D). The fingers are respectively connected to the strip through their ends. An independent claim is also included for a differential dipole filtering antenna comprising a higher-order differential filter.

Description

The present invention relates to a coplanar differential bi-ribbon delay line. It also relates to a higher order differential filter and a filter antenna equipped with such a bi-ribbon delay line.

Radio frequency transmit / receive systems powered by differential electrical signals are very attractive for current and future wireless communications systems, especially for autonomous communicating object concepts. A differential supply is a supply of two signals of equal amplitude in phase opposition. It helps reduce, or even eliminate, unwanted "common mode" noise in transmit and receive systems.

In the field of mobile telephony for example, when a non-differential system is used, a significant degradation of the radiation performance is indeed observed when the operator holds a handset equipped with such a system. This degradation is caused by the variation, due to the hand of the operator, of the distribution of the current on the chassis of the combined used as ground plane. The use of a differential power supply makes the system symmetrical and thus reduces the current concentration on the handset case, thus making the handset less sensitive to common mode noise introduced by the operator's hand.

In the field of antennas, a non-differential power supply causes the radiation of an undesired cross component due to the common mode flowing on the non-symmetrical power cables. The use of a differential power supply eliminates the cross-radiation of the measurement cables and thus makes it possible to obtain reproducible measurements independent of the measurement context as well as perfectly symmetrical radiation diagrams.

In the field of active components, the "push-pull" power amplifiers whose structure is differential have several advantages, such as the doubling of the output power and the elimination of higher order harmonics. In reception, the low noise differential amplifiers offer several perspectives in terms of reduction of the noise factor. Also, the use of a differential structure prevents unwanted triggering of the oscillators by common mode noise.

A differential bi-ribbon delay line may be useful for connecting two differential devices, such as for example two filter devices, so as to form a higher order filter. In the particular case of connecting two filtering devices, the differential bi-ribbon delay line must have the characteristics of a quarter-wave phase shift line (Π / 2) to be used as an impedance inverter.

More generally, a differential bi-band delay line may be useful in a large number of applications requiring the connection of differential devices, including as a phase-shifter. For example, in a power application of an antenna array, where several different antennas are powered by one or more sources, at least one phase shifter of this type can be advantageously envisaged.

However, more and more differential devices such as filtering devices or dipole antennas are designed in differential CPS ("CoPlanar Stripline" technology, for "coplanar band line"). Indeed, the differential CPS technology makes it possible to benefit from the advantages of the differential structures while allowing a simple coplanar integration with discrete elements: it is not necessary to create connections of type via to connect the elements between them. In addition, the absence of ground plane makes it possible to envisage a simple and less disturbing connection with, for example, a differential antenna.

It is therefore advantageous to also use this technology to produce a differential bi-ribbon delay line, in particular a quarter-wave line. According to this technique, a bi-ribbon line for propagation of a differential signal comprises two rectilinear conductive strips arranged in parallel on the same face of a dielectric substrate and each comprising a first and a second end. The first two ends of the two conductive strips form two conductors of a first bi-ribbon port connecting to a first external differential device. The two second ends of the two conductive strips form two conductors of a second bi-ribbon port connecting to a second external differential device.

Thus, a differential bi-band delay line designed in this way can optimally connect to external devices designed in differential CPS technology. The delay that it induces and its impedance are directly related to its length, the spacing between its two conductive strips and their width.

For example, the document "Broadband and compact coupled coplanar stripline filters with impedance steps", by Ning Yang et al, IEEE Transactions on Microwave Theory and Techniques, vol. 55, No. 12, December 2007 , describes the realization of a differential CPS filter, in particular with reference to the figure 12 . This compact topology makes it possible to achieve high bandwidths with high out-of-band rejection for filters of order 2, 3 or 4. Unfortunately, the interposition of a quarter-wave delay line in differential CPS technology between two devices Filtration methods such as that illustrated in the aforementioned document, although necessary to obtain a higher order filter with good rejection properties, substantially increases the size of the complete device, mainly because of its length.

It may thus be desired to design, in differential CPS technology, a bi-ribbon delay line having a better compactness while maintaining the same performance in terms of phase shift and impedance matching as a bi-ribbon propagation line. predetermined phase shift delay.

The subject of the invention is therefore a coplanar differential bi-ribbon delay line, comprising two conductive strips disposed on the same face of a dielectric substrate and each comprising a first and a second end, the first two ends of the two conductive strips forming two conductors of a first bi-ribbon port connecting to a first external differential device, the two second ends of the two conductive strips forming two conductors of a second bi-ribbon port connecting to a second external differential device, this bi-ribbon line wafer being further shaped in the form of a printed circuit for presenting structural discontinuities generating at least one impedance jump and at least one capacitive coupling with interdigitated capacitance between its two conducting strips so as to reproduce a predetermined phase shift , the interdigital capacitance being formed by at least one pair of fingers ductors connected respectively by one of their ends to the two conductive strips.

On crée donc un déphasage qui, dans le cas d'une ligne de propagation, est normalement fonction de sa longueur.The printed circuit of the L, C type thus created has, thanks to its discontinuities (impedance jump and capacitive coupling), an inductance L and a capacitance C such that it can reproduce the phase shift characteristics of a propagation line. classic delay. Indeed, the phase shift φ of this circuit can be expressed as a function of L and C in the following way: $\phi =2\pi \sqrt{\mathit{LC}}\mathrm{.}$

A phase shift is thus created which, in the case of a propagation line, is normally a function of its length.

Optionally, at least one of the structural discontinuities comprises a variation of the distance between the two conductive strips for performing an impedance jump.

Optionally also, a first discontinuity of increasing the distance between the two conductive strips and a second discontinuity of reduction of the distance between the two conductive strips form an area of the substrate in which the bi-ribbon line has a spacing between its conductive strips greater than the spacing between the two conductors of each of its bi-ribbon connection ports.

Optionally also, the interdigitated capacitance is formed in the area of the substrate in which the bi-ribbon line has a greater spacing between its conductive strips, the pair of conductive fingers extending laterally inwardly of this area from two conductive strips.

Optionally also, the structure discontinuities are generating at least one impedance jump and at least one capacitive coupling between its two conductive strips so as to reproduce a quarter-wave phase shift.

The subject of the invention is also a higher order differential filter comprising two coplanar coupled resonator differential filtering devices and a bi-band transmission line of a differential signal as defined above, this bi-ribbon line being connected, via its first bi-ribbon port, to one of the two filtering devices and, via its second bi-ribbon port, to the other of the two filtering devices.

Optionally, each of the two coplanar coupled resonator differential filtering devices comprises a pair of coupled resonators disposed on the same face of a dielectric substrate, each resonator comprising two conductive strips positioned symmetrically with respect to a plane perpendicular to the face on which the resonator is arranged, these two conductive strips being respectively connected to two conductors of a differential bi-ribbon port of the corresponding differential filtering device, each conducting band of each resonator being further folded back on itself so as to form a capacitive coupling between its two ends.

Thus, the folding of each conductive strip on itself makes it possible to envisage a smaller filter size, for geometrical reasons. Furthermore, the fact that this refolding is designed to form a capacitive coupling between the two ends of each conductive strip creates at least one additional frequency transmission zero ensuring high bandwidth and out-of-band rejection performance. filtering device. Finally, the capacitive coupling by folding also generating a magnetic coupling, the size of each conductive strip can be further reduced while ensuring the same filtering function of the assembly.

Finally, the subject of the invention is also a differential dipole filter antenna comprising at least one higher order differential filter as defined above.

Optionally, a differential dipole filter antenna according to the invention may comprise a radiating structure shaped to integrate in its external dimensions said differential filter of higher order.

• la figure 1 représente schématiquement la structure générale d'une ligne bi-ruban différentielle de l'état de la technique en technologie CPS,
• la figure 2 représente un circuit électrique équivalent de la ligne bi-ruban de la figure 1,
• la figure 3 représente schématiquement la structure générale d'une ligne à retard bi-ruban différentielle selon un mode de réalisation de l'invention,
• la figure 4 représente schématiquement la structure générale d'un premier exemple de dispositif de filtrage pour la réalisation d'un filtre d'ordre supérieur selon l'invention,
• la figure 5 représente un schéma électrique équivalent du dispositif de filtrage de la figure 4,
• la figure 6 illustre la caractéristique d'une réponse fréquentielle en transmission et en réflexion du dispositif de filtrage de la figure 4,
• la figure 7 représente schématiquement la structure générale d'un deuxième exemple de dispositif de filtrage pour la réalisation d'un filtre d'ordre supérieur selon l'invention,
• la figure 8 représente schématiquement la structure générale d'un troisième exemple de dispositif de filtrage pour la réalisation d'un filtre d'ordre supérieur selon l'invention,
• la figure 9 représente schématiquement la structure générale d'un ensemble de filtrage et d'adaptation d'impédances à deux filtres tels que celui de la figure 8, selon un mode de réalisation de l'invention,
• la figure 10 représente schématiquement la structure générale d'un filtre d'ordre supérieur selon un premier mode de réalisation de l'invention,
• la figure 11 représente schématiquement la structure générale d'un filtre d'ordre supérieur selon un second mode de réalisation de l'invention,
• la figure 12 illustre la caractéristique d'une réponse fréquentielle en transmission et en réflexion du filtre de la figure 11,
• les figures 13, 14 et 15 représentent schématiquement trois modes de réalisation d'antennes filtrantes selon l'invention.

The invention will be better understood with the aid of the description which follows, given solely by way of example and with reference to the appended drawings in which:

• the figure 1 schematically represents the general structure of a differential bi-ribbon line of the state of the art in CPS technology,
• the figure 2 represents an equivalent electrical circuit of the bi-ribbon line of the figure 1 ,
• the figure 3 schematically represents the general structure of a differential bi-ribbon delay line according to one embodiment of the invention,
• the figure 4 schematically represents the general structure of a first exemplary filtering device for producing a higher order filter according to the invention,
• the figure 5 represents an equivalent electrical diagram of the filtering device of the figure 4 ,
• the figure 6 illustrates the characteristic of a frequency response in transmission and reflection of the filtering device of the figure 4 ,
• the figure 7 schematically represents the general structure of a second exemplary filtering device for producing a higher order filter according to the invention,
• the figure 8 schematically represents the general structure of a third example of a filtering device for producing a higher order filter according to the invention,
• the figure 9 schematically represents the general structure of a filter and impedance matching set with two filters such as that of the figure 8 according to one embodiment of the invention,
• the figure 10 schematically represents the general structure of a higher order filter according to a first embodiment of the invention,
• the figure 11 schematically represents the general structure of a higher order filter according to a second embodiment of the invention,
• the figure 12 illustrates the characteristic of a frequency response in transmission and reflection of the filter of the figure 11 ,
• the figures 13 , 14 and 15 schematically represent three embodiments of filter antennas according to the invention.

The coplanar differential bi-ribbon delay line 10 shown in FIG. figure 1 comprises two conductive strips 12 and 14 disposed on the same flat face 16 of a dielectric substrate.

The conductive strip 12 comprises a first end E1 and a second end S1. Similarly, the second conductive strip 14 comprises a first end E2 and a second end S2.

The first two ends E1 and E2 of the two conductive strips 12 and 14 respectively form two conductors of a first bi-ribbon port 18 for connection to a first external differential device (not shown) and the two second ends S1 and S2 of the two bands. Conductors respectively form two conductors of a second bi-ribbon port 20 for connection to a second external differential device (not shown).

The two conductive strips 12 and 14 are rectilinear. They are also parallel and symmetrical with respect to a plane P perpendicular to the plane face 16 and forming a virtual electric ground plane of the differential bi-ribbon line. They are of a width w and distant from each other by a distance s, these two parameters w and s defining the characteristic impedance of the bi-ribbon line 10.

They are also of length I, this length I defining the phase shift generated by the bi-ribbon line on a differential signal that it propagates and therefore its impedance matching. This is why, for a predetermined phase shift, for example a quarter-wave phase shift, a certain length of this bi-ribbon propagation line is necessary, which generates an additional bulk of the device in which the bi-ribbon line 10 is integrated.

An equivalent electric circuit of this bi-ribbon line 10 is represented on the figure 2 . This electrical circuit comprises two conducting wires 22 and 24 between which a capacitor C is arranged in parallel. Each portion of conducting wire 22 or 24, between one of the terminals of the capacitor C and one of the ends E1, E2, S1 and S2 of the circuit further comprises an inductance L. This electric circuit model produces a bi-ribbon delay line, of predetermined phase shift and obtained by given values of the capacitance C and inductances L.

The same electric circuit with discrete elements L and C can be realized using a bi-ribbon line 30 such as that shown on FIG. figure 3 according to one embodiment of the invention. This bi-ribbon line 30 can therefore be modeled by the same electrical circuit as the bi-ribbon line 10.

Like the bi-ribbon line 10, it comprises two conductive strips 32 and 34 disposed on the same plane face 36 of a dielectric substrate. But unlike the bi-ribbon line 10, the two conductive strips 32 and 34 are shaped as a printed circuit having discontinuities in structure.

The conductive strip 32 comprises a first end E'1 and a second end S'1. Similarly, the second conductive strip 34 comprises a first end E'2 and a second end S'2.

The first two ends E'1 and E'2 of the two conductive strips 32 and 34 respectively form two conductors of a first bi-ribbon port 38 for connection to a first external differential device (not shown) and the two second ends S ' 1 and S'2 of the two conductive strips respectively form two conductors of a second bi-ribbon port 40 for connection to a second external differential device (not shown).

The capacitive coupling and the impedance jumps of the bi-ribbon line 30, conferring on it a predetermined phase shift, are directly generated by structural discontinuities themselves generating an inductance and a capacitance. More specifically, these structural discontinuities comprise, on the one hand, linearity failures of the conductive strips 32 and 34 and, on the other hand, additional conductive branch formations extending from the conductive strips 32 and 34.

The breaks in linearity make it possible to vary the distance between the two conductive strips for achieving at least one impedance jump.

Thus, the first conductive strip 32 has several linearity breaks allowing a portion 32A of this conductive strip 32 to be further from the plane of symmetry P than the portions E'1 and S'1 forming the ends of this conductive strip 32 , while maintaining the portions E'1, S'1 and 32A parallel to the plane of symmetry P. These linearity breaks are made by a portion 32B of the conductive strip 32, extending laterally and orthogonally to the plane P of one end of the portion E'1 towards one end of the portion 32A, and by a portion 32C of the conductive strip 32, extending laterally and orthogonally at the plane P of the other end of the portion 32A towards one end of the portion S'1.

By symmetry, the second conductive strip 34 has a plurality of linearity breaks enabling a portion 34A of this conductive strip 34 to be further from the plane of symmetry P than the portions E '2 and S'2 forming the ends of this conductive strip. 34, while maintaining the portions E'2, S'2 and 34A parallel to the plane of symmetry P. These linearity breaks are made by a portion 34B of the conductive strip 34, extending laterally and orthogonally to the plane P d one end of the portion E'2 to one end of the portion 34A, and a portion 34C of the conductive strip 34, extending laterally and orthogonally to the plane P of the other end of the portion 34A towards one end of the serving S'2.

Therefore, the bi-ribbon line 30 has a first structure discontinuity, increasing the distance between its two conductive strips 32 and 34, made by the portions 32B and 34B, for the realization of a first impedance jump. by increasing this impedance. Indeed, the impedance increases with the distance between the two conductive strips.

It also has a second structure discontinuity, reducing the distance between its two conductive strips 32 and 34, made by the portions 32C and 34C, for performing a second impedance jump by reducing this impedance.

These two discontinuities of structure create a rectangular zone, essentially delimited by the portions 32B, 32A, 32C, 34C, 34A and 34B, in which the bi-ribbon line 30 has a spacing between its conducting strips 32 and 34 greater than the spacing between the two conductors E'1, E'2 and S'1, S'2 of each of its bi-ribbon connection ports 38 and 40.

The formations of additional conductive branches extending from the conductive strips 32 and 34 make it possible to create at least one interdigitated capacitor for carrying out the capacitive coupling between the two conductive strips 32 and 34.

More specifically, in the example of figure 3 , an interdigitated capacitance is formed by two conductive fingers 32D and 34D extending parallel to each other and orthogonal to the plane P, facing each other over at least a part of their length. The conductive finger 32D consists of a portion of a band rectilinear conductor whose one end is secured to the portion 32A of the first conductive strip 32 and the other end remains free, while the conductive finger 34D consists of a rectilinear conductive strip portion whose end is secured to the portion 34A of the second conductive strip 34 and the other end remains free.

The pair of conductive fingers thus extends laterally inwardly of the rectangular zone defined above from the portions 32A and 34A of the two conductive strips 32 and 34, which makes it possible to take advantage of the zone of the substrate in which the bi-line -tape 30 has a greater spacing between its conductive strips 32 and 34 to form the interdigitated capacitance.

In a variant, it is possible to create several parallel interdigitated capacitances in the rectangular zone defined above. This makes it possible to increase the capacity of the printed circuit formed by the bi-ribbon line 30 without changing its inductance. In other words, it is an additional parameter for adjusting the characteristic impedance of the bi-ribbon line 30 with a given phase shift. It will be noted, however, that the addition of interdigitated capacitances increases the length and therefore the size of the bi-ribbon line, which is not always desirable.

In concrete terms, it is simple for a person skilled in the art to adjust the dimensions of the aforementioned elements of the bi-ribbon line 30, so as to obtain a predetermined phase shift delay line by adjustment, in particular, of its capacitive coupling and of his impedance jumps.

The length I 'of the bi-ribbon line 30 thus produced is substantially smaller than the length I of a bi-ribbon line 10 of the state of the art with an identical equivalent electrical circuit, thanks to the structural discontinuities. As a result, a bi-ribbon line according to the invention has a better compactness while retaining the same characteristics as a bi-ribbon line of the state of the art.

In practice, it is possible in particular to design a quarter-wave line according to the invention to connect, with a better compactness, two differential filtering devices with coplanar coupled resonators and thus achieve a higher order filter in CPS technology.

A higher order differential filter according to the invention therefore comprises at least two differential coplanar coupled resonator filtering devices and at least one differential bi-ribbon line according to the invention, for example that described with reference to FIG. figure 3 this bi-ribbon line being connected via its first bi-ribbon port 38, to one of the two filtering devices and, via its second bi-ribbon port 40, to the other of the two filtering devices.

Each of the two filtering devices can for example be designed according to the example illustrated by the figure 12 of the document "Broadband and compact coupled coplanar stripline filters with impedance steps", by Ning Yang et al, IEEE Transactions on Microwave Theory and Techniques, vol. 55, No. 12, December 2007 .

However, the compactness of the filtering devices to which the differential bi-ribbon line is connected could also be advantageously improved. Combined with the improved compactness of the bi-ribbon line according to the invention, it would then allow to consider a higher order filter even more compact.

We will now describe in detail and with reference to Figures 4 to 8 several examples of differential filtering devices with compact resonators improved compactness, particularly suitable for the realization of higher order filters including at least one bi-ribbon line according to the invention.

The differential filter device 50 with coupled resonators shown in FIG. figure 4 comprises at least one pair of resonators 52 and 54, coupled to one another by capacitive coupling and arranged on the same plane face 56 of a dielectric substrate.

The first resonator 52, consisting of a bi-ribbon line portion, is connected to two conductors E "1 and E" 2 of a bi-ribbon connection port to a transmission line of a differential signal. These two conductors E "1 and E" 2 of the bi-ribbon port are symmetrical with respect to a plane P 'perpendicular to the plane face 56 and forming a virtual electric ground plane. They are of a width w and distant from each other by a distance s, these two parameters s and w defining the impedance of the bi-ribbon port.

Similarly, the second resonator 54, also consisting of a bi-ribbon line portion, is connected to two conductors S "1 and S" 2 of a bi-ribbon connection port to a transmission line of a differential signal. These two conductors S "1 and S" 2 of the bi-ribbon port are also symmetrical with respect to the virtual electrical ground plane P '.

The two resonators 52 and 54 are themselves symmetrical with respect to an axis normal to the plane P 'situated on the plane face 56. Therefore, the filtering device 50 is symmetrical between its differential input and output so that These can be totally reversed. Thus, in the following description of the embodiment shown on the figure 4 , the two conductors E "1 and E" 2 will be chosen by convention as the dual-input port of the filtering device 50, for receiving an unfiltered differential signal. The two conductors S "1 and S" 2 will be chosen by convention as being the dual-band output port of the filtering device 50, for the supply of the filtered differential signal.

More specifically, the first resonator 52 comprises two conductive strips identified by their references LE1 and LE2. These two conductive strips LE1 and LE2 are positioned symmetrically with respect to the virtual electrical ground plane P '. They are respectively connected to the two conductors E "1 and E" 2 of the input port. The second resonator 54 comprises two conductive strips identified by their references LS1 and LS2. These two conductive strips LS1 and LS2 are also positioned symmetrically with respect to the virtual electrical ground plane P '. They are respectively connected to the two conductors S "1 and S" 2 of the output port.

The capacitive coupling of the two resonators 52 and 54 is ensured by the arrangement in opposite but non-contact of their respective pairs of conductive strips. Thus, the conductive strips LE1 and LS1, located on the same side with respect to the virtual electrical ground plane P ', are arranged facing each other at a distance e from one another. Similarly, the conductive strips LE2 and LS2, situated on the other side with respect to the virtual electrical ground plane P ', are arranged facing each other at the same distance e from each other.

This distance e between the two resonators 52 and 54 mainly influences the bandwidth of the filtering device 50 and has a side effect on its characteristic impedance. The more e decreases, that is to say the more capacitive coupling is strong between the two resonators, the wider the bandwidth. This also has the effect of increasing the impedance. More precisely, the bandwidth is widened by the appearance of two distinct reflection zeros within this bandwidth, corresponding to two distinct resonance frequencies, when e is small enough to achieve the capacitive coupling between the two resonators. The lower the distance e, the more the two reflection zeros created move away from each other, thus widening the bandwidth. However, if they are too far apart, they can cause the separation of the enlarged bandwidth into two distinct bandwidths by reappearance of a significant reflection between the two zeros, which goes against the desired effect. Therefore, the distance e must be small enough to increase the bandwidth but also sufficiently important not to generate unwanted reflection within the bandwidth.

In a conventional way, for a good functioning of the resonators of a filtering device with coupled resonators, each conductive strip must be of length λ / 4, where λ is the apparent wavelength, for a substrate considered, corresponding to the frequency high operating filter device. Thus, if the conductive strips were arranged linearly in the extension of the input and output ports of the filtering device 50, the assembly would reach a length close to λ / 2: in practice, for a frequency of 3 GHz, one would obtain by example a length close to 3 cm.

But in fact, the conductive strips LE1, LE2, LS1 and LS2 are advantageously folded back on themselves so as to locally form additional capacitive and magnetic couplings between their two ends. The size of the filtering device 50 is thus reduced for at least two reasons: the collapses geometrically generate a size reduction of the assembly, but moreover, thanks to the capacitive and magnetic couplings, the size of each conductive strip can be further reduced. while ensuring a good functioning of the resonators. This capacitive and magnetic coupling further generates a feedback between the input and the output of each conductive strip, so as to create one or more additional transmission zeros at frequencies higher than the upper limit of the bandwidth of the filter device 50 The high band rejection is thus improved.

In the embodiment illustrated on the figure 4 the four conductive strips are of generally annular shape, their ends being folded inside this annular general shape over a portion of predetermined length thereof.

For a good operation of the filtering device 50, the folding of the ends of each conductive strip is located on a portion of this conductive strip disposed vis-à-vis the other conductive strip of the same resonator. Thus, the folds of ends of the conductive strips LE1 and LE2 are arranged vis-à-vis on both sides of the plane of symmetry P 'and close thereto.

More specifically, the conductive strip LE1 is generally rectangular in shape and consists of rectilinear conductive segments. A first segment LE1 ₁ having a first free end of the conductive strip LE1 extends inwardly of the rectangle formed by the conductive strip over a length L in a direction orthogonal to the virtual ground plane P '. A second segment LE1 ₂ , connected to this first segment at right angles, constitutes a part of the rectangle side parallel to the virtual ground plane P 'and close to it. A third segment LE1 ₃ , connected to this second segment at right angles, constitutes the side of the rectangle orthogonal to the virtual ground plane P 'and connected to the conductor E "1 of the input port A fourth segment LE1 ₄ , connected to this third segment at right angles, constitutes the side of the rectangle parallel to the virtual ground plane P 'and close to an outer edge of the substrate A fifth segment LE1 ₅ , connected to this fourth segment at right angles, constitutes the side of the orthogonal rectangle to the virtual ground plane P 'and opposite the side LE1 _3. A sixth segment LE1 ₆ , connected to this fifth segment at right angles, constitutes as the second segment LE1 ₂ a portion of the side of the rectangle parallel to the virtual ground plane P' Finally, a seventh segment LE1 ₇ having the second free end of the conductive strip LE1, connected to the sixth segment at right angles, extends inwardly of the rectangle over a long period of time. ur L in a direction orthogonal to the virtual ground plane P ', that is to say parallel to the segment LE1 ₁ and vis-à-vis it over the entire length L of folding.

The segments LE1 ₁ and LE1 ₇ are spaced a constant distance e _S over their entire length which ensures their capacitive coupling.

The conductive strip LE1 can also be seen as consisting of a folded main conductive strip connected at one of its ends to the conductor E "1, this main conductive strip comprising the segments LE1 ₁ , LE1 ₂ and the part of the segment LE1 ₃ located between the segment LE1 ₂ and the conductor E "1, and a stub-type branch folded on the main conductive strip, this stub-type branch comprising the other part of the segment LE1 ₃ , and the segments LE1 ₄ to LE1 ₇ . The "stub" type branch is then considered to be placed at the junction between the main conducting strip and the conductor E "1. It should theoretically have a total length of λ / 4, but the capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself can reduce this length, especially 10 to 20% on the derivation in "stub".

It is also interesting to note that a sufficiently small size of the segment LE1 ₄ makes it possible to bring the segments LE1 ₃ and LE1 ₅ closer together, but also the segments LE1 ₃ and LE1 ₁ , or the segments LE1 ₅ and LE1 ₇ , so as to multiply the number of capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself. These multiple couplings improve the operation of the filtering device 50.

The coupling length L between the two folded ends, ie the two segments LE1 ₁ and LE1 ₇ , mainly influences the bandwidth of the filtering device 50, but also has a side effect on the high band rejection. The more it increases, the lower the bandwidth but the higher the band rejection is improved.

The distance e _S between the two folded ends mainly influences the high-band rejection of the filtering device 50: the smaller it is, the higher the rejection in the high band. It should be noted, however, that this distance can not be less than a limit imposed by the precision of the etching of the conductive strip LE1 on the substrate.

The conductive strip LE2 consists, like the conductive strip LE1, of seven conductive segments LE2 ₁ to LE2 ₇ disposed on the plane face 56 of the substrate symmetrically to the seven segments LE1 ₁ to LE1 ₇ with respect to the virtual ground plane P ' . The two conductive strips LE1 and LE2 are spaced a constant distance e ₁ , corresponding to the distance separating the segments LE1 ₂ and LE1 ₆ , on the one hand, the segments LE2 ₂ and LE2 ₆ , on the other hand.

This distance e ₁ mainly influences the impedance of the first resonator 52, that is to say the input impedance of the filtering device 50, but also has a side effect on the bandwidth of the filtering device 50. More it increases, the more the impedance increases and less markedly, the more the bandwidth is reduced.

Since the two resonators 52 and 54 are symmetrical with respect to an axis normal to the virtual ground plane P 'situated on the plane face 56, the conductive strips LS1 and LS2 each consist, like the conductive strips LE1 and LE2, of seven conducting segments. LS1 ₁ to LS1 ₇ and LS2 ₁ to LS2 ₇ respectively, printed on the flat face 56 of the substrate symmetrically to the segments of the conductive strips LE1 and LE2 with respect to this axis. By symmetry also, the two conductive strips LS1 and LS2 are spaced a constant distance e ₂ equal to e ₁ , corresponding to the distance separating the segments LS1 ₂ and LS1 ₆ , on the one hand, of the segments LS2 ₂ and LS2 ₆ , on the other hand.

This distance e ₂ also mainly influences the impedance of the second resonator 54, that is to say the output impedance of the filtering device 50, but also a side effect on the bandwidth of the filter device 50. The more it increases, the more the impedance increases and less markedly, the lower the bandwidth is reduced.

The distance e separating the two resonators 52 and 54 corresponds to the distance separating the segments LE1 ₅ and LE2 ₅ , on the one hand, from the segments LS1 ₅ and LS2 ₅ , on the other hand. The capacitive coupling between the two resonators 52 and 54 is thus established over the entire length of the segments LE1 ₅ and LE2 ₅ , on the one hand, and the segments LS1 ₅ and LS2 ₅ , on the other hand.

In a topology such as that illustrated on the figure 4 where the length of the rectangle formed by any one of the conductive strips is approximately twice its width and where the length L folds is half the length of the rectangle therein, obtains dimensions of the rectangle formed by each conductive strip close to λ / 30 by λ / 60, ie dimensions of the filtering device 50 close to λ / 15 by λ / 30. These dimensions make it possible to achieve a much better compactness than those of existing devices.

The figure 5 schematically presents an equivalent electric circuit of the filtering device 50 previously described.

In this circuit, a first inverter 60 represents an impedance jump, from Z ₀ to Z ₁ , at the input of the filtering device 50. The impedance Z ₀ is determined by the parameters s and w of the conductors E "1 and E. 2 of the input port, while the impedance Z ₁ is determined in particular by the distance e ₁ between the conductive strips LE 1 and LE 2.

A second inverter 62 represents the corresponding impedance jump, from Z ₁ to Z ₀ , at the output of the filtering device 50.

The first and second coupled resonators 52 and 54 are each represented by an LC circuit with capacitance C and inductance L in parallel. These two LC circuits are connected, on the one hand, respectively to the first and second inverters 60 and 62 and, on the other hand, to ground.

Finally, the folding of the conductive strips LE1, LE2, LS1 and LS2 creates additional couplings, inside each resonator but also between the resonators, which can be represented by a feedback circuit LC 64, with capacitance C1 and inductance L1. parallel, connected, on the one hand, to the junction 66 between the first resonator 52 and the first inverter 60 and, on the other hand, to the junction 68 between the second resonator 54 and the second inverter 62. This LC feedback circuit 64 improves the high band rejection of the filtering device 50 by adding one or more transmission zeros in the high frequencies.

The graphic shown on the figure 6 represents the characteristic of a frequency response in transmission and reflection of the filtering device described above.

The reflection coefficient S ₁₁ of this frequency response shows a bandwidth of -10 dB (generally accepted definition of the bandwidth in reflection) of between about 3.2 and 4.4 GHz. As indicated above, the bandwidth is widened by the presence of two distinct reflection zeros within this bandwidth, these two zeros being due to the presence of the two coupled resonators remote from e in the filtering device 50. However we can see clearly figure 6 if they are too far apart, the portion of curve S ₁₁ situated between these two reflection zeros can go back up to -10 dB, which generates a separation of the enlarged bandwidth into two distinct bandwidths. Therefore, the distance e should not be too small not to cause reflection greater than -10 dB in the extended bandwidth.

The transmission coefficient S ₂₁ of the frequency response shows a bandwidth of -3 dB (generally accepted definition of the bandwidth in transmission), between about 2.7 and 4.5 GHz, as well as two transmission zeros at about 5.1 and 6.9 GHz.

One of these two out-of-band transmission zeros is due to the coupling between the two resonators of the filter device 50 over the entire length of their portions LE1 ₅ , LE2 ₅ on the one hand and LS1 ₅ , LS2 ₅ on the other hand . The other of these two transmission zeros is due to the additional intra-resonator couplings created by the folding of the conductive strips on themselves. These two transmission zeros cause a high rejection of the high band filter and an asymmetry of the frequency response due to the low band mean rejection. But this asymmetry can be advantageous, especially for a direct integration application of the filtering device 50 in a differential antenna. Indeed, such antennas generally have high resonances low frequency and therefore equivalent to high-pass filters, which compensates for the asymmetry of the filter device 50 by improving its low band rejection.

A second example of differential filtering device with improved compactness is shown schematically on the figure 7 . This device 50 'comprises a pair of resonators 52' and 54 ', coupled to each other by capacitive coupling and disposed on the same plane face 56 of a dielectric substrate. These two resonators are similar to those, 52 and 54, of the device of the figure 4 .

On the other hand, in this second example, the two resonators 52 'and 54' are not symmetrical with respect to an axis normal to the plane P 'situated on the plane face 56. Indeed, the distance e ₁ separating the two conductive strips LE1 and LE2 of the first resonator 52 'is distinct from the distance e ₂ between the two conductive strips LS1 and LS2 of the second resonator 52'. In the illustrated example, the distance e ₂ is greater than the distance e ₁ .

However, the capacitive coupling between the two resonators 52 'and 54' is not broken so far. Indeed, due to the folding of the conductive strips on themselves, they remain in vis-à-vis at least a portion of their length, more specifically at least a portion of the lengths LE1 ₅ and LS1 _5, d the one hand, and lengths LS2 and LE2 _{5 5,} on the other hand. In comparison with the existing one, it would not be possible, for example, to conceive of such a difference between the distances e ₁ and e ₂ in the filtering device described with reference to FIG. figure 12 of the above-mentioned "Broadband and compact coplanar stripline filters with impedance steps" document, because in this document it is the free ends of the conductive strips which are arranged opposite each other so that even a slight offset between they would break the capacitive coupling between the two resonators.

Since these distances e ₁ and e ₂ make it possible to adjust respectively the input and output impedances of the filtering device 50 ', it is thus possible to design a bandpass filtering device which also fulfills an adaptation function of impedances between the circuits to which it is intended to be connected. In the example shown on the figure 7 the distance e ₁ thus generates an input impedance Z ₁ smaller than the output impedance Z ₂ generated by the distance e ₂ .

This second example allows the direct integration of a filtering device according to the invention with differential antennas and differential active circuits of different impedances. Note, however, that such a direct integration with a single filter device works all the better that the difference between the impedances Z ₁ and Z ₂ is small.

Alternatively, a set of several filtering devices according to the second example of the invention added in series can be used to facilitate the impedance matching between very different impedance circuits.

Such a set with two filtering devices is for example represented diagrammatically on the figure 8 .

In this assembly, an amplifier 70 is connected to the input of a first filtering device 72, via the input port 74 of this first filtering device. Since the impedance of the amplifier 70 has a value Z ₁ , the first filtering device 72 is designed, by adjusting the distance between the folded conductive strips of its first resonator, to present a conjugate value input impedance Z ₁ * thus ensuring a maximum power transfer between the first filtering device 72 and the amplifier 70.

An antenna 76 is connected to the output of a second filtering device 78 via the output port 80 of this second filtering device. Since the impedance of the antenna 76 has a value Z ₂ , the second filtering device 78 is designed, by adjusting the distance between the folded conductive strips of its second resonator, to present a conjugate value output impedance Z ₂ * thus ensuring maximum power transfer between the second filter device 78 and the antenna 76.

Finally, the two filtering devices 72 and 78 are advantageously connected to each other via a quarter-wave line 82 according to the invention fulfilling an inverter function, the output of the first filtering device 72 and the input of the second device. filtering 78 being designed, by adjusting the distance between the folded conductive strips of the second resonator of the first filtering device 72 and the distance between the folded conductive strips of the first resonator of the second filtering device 78, to present the same impedance Z ₀ . This same impedance Z ₀ ensures the adaptation of impedances and can be chosen so as to ensure the best possible rejection.

Thus, the adaptation of the impedances Z ₁ and Z ₂ which can be very different is via an intermediate impedance Z ₀ through the assembly comprising the two asymmetric filtering devices 72 and 78 and the quarter wave line 82 .

The presence of the quarter wave line 82 between the two filtering devices 72 and 78 also makes it possible to improve overall the performance of the higher order filter thus constituted, in terms of bandwidth.

A third example of differential filtering device with improved compactness is shown schematically on the figure 9 . This filtering device 50 "comprises a pair of resonators 52" and 54 ", coupled together by capacitive coupling and disposed on the same plane face 56 of a dielectric substrate.

In this third example, the two resonators 52 "and 54" are symmetrical with respect to an axis normal to the plane P 'situated on the plane face 56. distance e ₁ between the two conductive strips LE1 and LE2 of the first resonator 52 "is equal to the distance e ₂ between the two conductive strips LS1 and LS2 of the second resonator 54". As a variant, these two distances could be different, as in the second example, for the filtering device to further fulfill an impedance matching function.

On the other hand, this third example is distinguished from the first and second examples by the general shape of the folded conductive strips.

Indeed, in this example, the four conductive strips are of generally annular shape, their ends being folded inside this annular general shape over a portion of predetermined length thereof, but they are more precisely of generally square shape. . In addition, each of them has additional folding on at least a portion of the sides of the square general shape.

For example, the conductive strip LE1 comprises three additional folds LE1 ₈ , LE1 ₉ and LE1 ₁₀ in the three sides of the square general shape not having the folding of its two ends. To improve the compactness of the assembly, the three additional folds are directed towards the inside of the square general shape. They are for example in the form of niche. By symmetry, the conductive strips LE2, LS1 and LS2 comprise the same additional folds, referenced LE2 ₈ , LE2 ₉ and LE2 ₁₀ for the conductive strip LE2; LS1 ₈ , LS1 ₉ and LS1 ₁₀ for the conductive strip LS1; LS2 ₈ , LS2 ₉ and LS2 ₁₀ for the conductive strip LS2.

In this example, the overall square shape of each conductive strip LE1, LE2, LS1 and LS2 implies a generally square shape of the filtering device 50 ", so the compactness of the latter is optimal.

In addition, the additional folds create additional capacitive and magnetic couplings that can further improve the performance of the filter device 50 ".

As indicated above, the length L of the folding of the two ends of each conductive strip within its square general shape can be adjusted to adjust the bandwidth of the filter device 50 ".

In this square topology, for example, the dimensions of the filter device 50 "are obtained close to λ / 20 per side.

Note that an improved compactness filter device is not limited to the examples described above. Other geometric shapes are possible for such a filtering device, from the moment they provide for a folding of each conductive strip of each resonator on itself so as to form a capacitive coupling between its two ends.

This filter device with improved compactness is particularly suitable for the design, with a bi-ribbon line according to the invention, of a smaller order of higher order filter.

For example, as shown on the figure 10 a higher order differential filter 90 etched on a substrate 92 has two coplanar coupled resonator differential filtering devices 94 and 96 in accordance with the first example shown in FIG. figure 4 . It further comprises a differential bi-ribbon line 98 conforming to that shown in FIG. figure 3 connected, via one of its two bi-ribbon ports, to one of the two differential filtering devices and, via its other bi-ribbon port, to the other of the two differential filtering devices.

For example also, as illustrated on the figure 11 a higher order differential filter 100 etched on a substrate 102 comprises two coplanar coupled resonator differential filtering devices 104 and 106 in accordance with the third example illustrated in FIG. figure 9 . It further comprises a differential bi-ribbon line 108 conforming to that shown in FIG. figure 3 connected, via one of its two bi-ribbon ports, to one of the two differential filtering devices and, via its other bi-ribbon port, to the other of the two differential filtering devices.

Specifically, this higher order filter is for example designed to operate in a high frequency band allocated to Ultra Wide Band communications, according to the European ULB standard, or even between 6 and 9 GHz. The substrate 102 is for example a high permittivity substrate (εr = 10). The dimensions of this higher order filter 100 with improved compactness are then 6 mm long by 3.5 mm wide.

The graphic shown on the figure 12 represents the characteristic of a frequency response in transmission and in reflection of the higher-order filter illustrated on the figure 11 .

The reflection coefficient S ₁₁ of this frequency response shows a bandwidth of -10 dB (generally accepted definition of the bandwidth in reflection) of between about 6 and 9 GHz and has four reflection zeros in the bandwidth.

The transmission coefficient S ₂₁ of this frequency response shows a bandwidth of -3 dB (generally accepted definition of the bandwidth in transmission), also between about 6 and 9 GHz, and a transmission zero at about 9.8 GHz.

This transmission zero causes a high rejection of the high band filter and an asymmetry of the frequency response due to the low band mean rejection. Rejections of the order of 50 dB in the high band and 30 dB in the low band are obtained. However, as indicated above, this asymmetry may be advantageous, especially for a direct integration application of this filter 100 in a differential antenna.

The Figures 13 to 15 schematically illustrate three examples of differential filter dipole antennas each advantageously incorporating a differential filter of higher order with improved compactness such as that illustrated in FIG. figure 11 .

The filtering dipole antenna 110 shown in FIG. figure 13 comprises on the one hand a radiating electric dipole 112 and on the other hand a higher order differential filter 100 such as that described with reference to FIG. figure 11 . The electric dipole 112 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of elliptical shape. This type of dipole is very wide bandwidth. The relative bandwidth defined by the relationship Δf / f ₀ , where Δf is the width of the bandwidth and f ₀ the central operating frequency of the antenna, may exceed 100%.

The two arms of the dipole 112 are directly connected to the two conductors of the output port of the filter 100. The two conductors of the input port of the filter 100 are for their part to be supplied with a differential signal.

The filtering dipole antenna 120 shown on the figure 14 comprises on the one hand a radiating electric dipole 122 and on the other hand a higher order differential filter 100 such as that described with reference to FIG. figure 11 . The electric dipole 122 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of "butterfly" shape. More specifically, the radiating structure of the dipole has a thin portion, in a central zone of the antenna comprising the connection to the filter 100, which widens outwardly of the antenna on both sides of the dipole. This type of radiating dipole is medium bandwidth. Its relative bandwidth .DELTA.f / f ₀ is about 20%.

As before, the two arms of the dipole 122 are directly connected to the two conductors of the output port of the filter 100. However, the conductors of the input port of the filter 100 are intended to be fed with a differential signal.

Finally, the filtering dipole antenna 130 represented on the figure 15 comprises on the one hand a radiating electric dipole 132 and on the other hand a differential filter of higher order 100 such as that described with reference to FIG. figure 11 . The electric dipole 132 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of "butterfly" shape. However, it differs from the electric dipole 122 in particular in that the two broad ends of its radiating structure, oriented towards the outside of the antenna, are shaped to integrate in their external dimensions (ie greater length and greater width) the filter This results in a further gain in compactness of the filter antenna 130 relative to the filter antenna 120.

Moreover, as before, the two arms of the dipole 132 are directly connected to the two conductors of the output port of the filter 100. The two conductors of the input port of the filter 100 are for their part to be supplied with a differential signal.

A number of constant filtering devices, a differential dipole filter antenna according to the invention is smaller than a conventional corresponding antenna, in particular due to the better compactness of the differential bi-ribbon line used. Alternatively, at a constant overall size, a differential dipole filter antenna according to the invention is more efficient because it may comprise a larger number of filtering devices to achieve an even higher order filtering, thus more efficient in terms of bandwidth.

It clearly appears that a differential bi-ribbon delay line such as that described previously with reference to FIG. figure 3 can achieve a much better compactness than the known differential bi-ribbon lines realized in CPS technology, while retaining their characteristics.

Given the frequency bands in which it can operate when associated with filtering devices made in CPS technology, it is particularly suitable for new radio communication protocols that require very wide bandwidths. Its compactness also makes it advantageous for communicating miniature objects.

The coplanar structure of this differential bi-ribbon delay line also facilitates its realization in hybrid technology and its integration in monolithic technology with structures comprising discrete elements mounted on area. In particular, it is simple to conceive of it as an element of a higher order filter in integration with a differential dipole antenna with a broadband coplanar radiating structure, as has been illustrated by several examples, by chemical or mechanical etching on substrates with low or high permittivity depending on the applications and desired performance.

A higher order filter according to the invention can also find applications in the millimeter frequency band where its small size and its high performance allow it to be integrated in monolithic technology with antennas and active circuits.

Finally, it will be noted that other applications than those presented above are also conceivable for a bi-ribbon line according to the invention. In particular, a bi-ribbon line according to the invention can be used as a phase-shifter, for example in a power supply application of an antenna array where several different antennas with different phase-shifts are fed by the same source. In this case, the antennas can be connected to each other by bi-ribbon lines according to the invention.

Claims (9)

Coplanar differential bi-ribbon delay line (30), comprising two conductive strips (32, 34) arranged on the same face (36) of a dielectric substrate and each comprising a first and a second end (E'1, E ' 2, S'1, S'2), the first two ends (E'1, E'2) of the two conductive strips forming two conductors of a first bi-ribbon port (38) for connection to a first external differential device , the two second ends (S'1, S'2) of the two conductive strips forming two conductors of a second bi-ribbon port (40) for connection to a second external differential device, characterized in that it is conformed under printed circuit form for presenting structure discontinuities (32B, 32C, 32D, 34B, 34C, 34D) generating at least one impedance jump and at least one capacitive coupling with interdigitated capacitance between its two conductive strips ( 32, 34) so as to reproduce a predetermined phase shift, the interdigitated capacitance being formed by at least one pair of conductive fingers (32D, 34D) respectively connected at one end to the two conductive strips. A coplanar differential bi-ribbon delay line (30) according to claim 1, wherein at least one of the structural discontinuities (32B, 32C, 32D, 34B, 34C, 34D) comprises a variation of the distance between the two bands conductors (32, 34) for performing an impedance jump. A coplanar differential bi-ribbon delay line (30) according to claim 2, wherein a first gap (32B, 34B) for increasing the distance between the two conductive strips (32, 34) and a second gap (32C, 34C ) of reducing the distance between the two conductive strips (32, 34) form an area of the substrate in which the bi-ribbon line has a spacing between its conductive strips greater than the spacing between the two conductors (E'1, E 2, S'1, S'2) of each of its bi-ribbon connection ports (38, 40). A coplanar differential bi-ribbon delay line (30) according to claim 3, wherein the interdigitated capacitance is formed in the area of the substrate in which the bi-ribbon line has a larger gap between its conductive strips (32, 34), the pair of conductive fingers (32D, 34D) extending laterally inwardly of this region from the two conductive strips. A coplanar differential bi-ribbon delay line (30) according to any one of claims 1 to 4, wherein the structural discontinuities (32B, 32C, 32D, 34B, 34C, 34D) are generating at least one impedance jump and at least one capacitive coupling between its two conductive strips (32, 34) so as to reproduce a quarter-phase phase shift. 'wave. A higher order differential filter (90; 100) having two coplanar coupled resonator differential filtering devices (94,96; 104,106) and a coplanar differential bi-band delay line (98; 108) according to any one of claims 1 to 5, this bi-ribbon line being connected, via its first bi-ribbon port, to one of the two filtering devices and, via its second bi-ribbon port, to the other of the two filtering devices . A higher order differential filter (90; 100) according to claim 6, wherein each of the two coplanar coupled resonator differential filtering devices (94, 96; 104, 106) comprises a pair of coupled resonators (52, 54; ', 54') arranged on the same face (56) of a dielectric substrate, each resonator having two conductive strips (LE1, LE2, LS1, LS2) positioned symmetrically with respect to a plane perpendicular to the face (56) on which is disposed the resonator (52, 54; 52 ', 54'), these two conductive strips (LE1, LE2, LS1, LS2) being respectively connected to two conductors (E "1, E" 2, S "1, S "2) of a differential bi-ribbon port of the corresponding differential filtering device, each conductive strip (LE1, LE2, LS1, LS2) of each resonator (52, 54; 52 ', 54') being further folded over itself to form a capacitive coupling between its two ends. Differential filter dipole antenna (110; 120; 130) having at least one higher order differential filter (90; 100) according to claim 6 or 7. A differential dipole filter antenna (130) according to claim 8, comprising a radiating structure (132) shaped to integrate in said outer dimensions said higher order differential filter (90; 100).

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