WO2013117049A1 - Self-oscillatory flyback converter - Google Patents

Self-oscillatory flyback converter Download PDF

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Publication number
WO2013117049A1
WO2013117049A1 PCT/CN2012/074151 CN2012074151W WO2013117049A1 WO 2013117049 A1 WO2013117049 A1 WO 2013117049A1 CN 2012074151 W CN2012074151 W CN 2012074151W WO 2013117049 A1 WO2013117049 A1 WO 2013117049A1
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Prior art keywords
circuit
current source
constant current
self
flyback converter
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PCT/CN2012/074151
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French (fr)
Chinese (zh)
Inventor
郭国文
尹向阳
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广州金升阳科技有限公司
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Publication of WO2013117049A1 publication Critical patent/WO2013117049A1/en

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    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of dc power input into dc power output
    • H02M3/22Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/338Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement

Definitions

  • the present invention relates to a self-oscillating flyback converter, and more particularly to a self-oscillating flyback converter with a constant current source soft start circuit.
  • the Ringing Choke Converter is favored by designers because of its low design cost and strong market competitiveness.
  • its output characteristics depend largely on discrete components.
  • the consistency of the device when the consistency of the transformer, main power tube, control transistor, optocoupler and other components is better, the output performance is more stable. Therefore, reducing the stringent requirements of the self-oscillating flyback converter circuit (RCC) for discrete device uniformity and improving its output voltage stability have become the focus of designers.
  • designers often use offset compensation circuits (control the base of the triode to add turn-off compensation), and the stability of the output voltage is improved to some extent.
  • the scheme is difficult to debug, and the consistency of the product is still poor, especially in the output. The effect is more obvious when the power is increased and the output capacitive load is increased.
  • the self-oscillating flyback converter mainly includes a filtering part and a soft start part. , MOS tube, transformer, pulse frequency modulation section (PFM), auxiliary power supply, isolated optocoupler, regulated output loop section.
  • PFM pulse frequency modulation section
  • the input power is connected to the output circuit part through the transformer, the soft start part is connected to the gate of the MOS tube, the gate of the MOS tube is also connected to the pulse frequency modulation part, the auxiliary power supply is connected to the pulse frequency modulation part, the pulse frequency modulation part and the regulated output circuit part are
  • the indirect reference amplification section and the isolated optocoupler form a voltage negative feedback loop.
  • the soft start circuit of the flyback converter improves the stability of the output voltage at the start of the product.
  • the working principle is as follows: After the input is powered on, the voltage of the soft start circuit rises, the MOS transistor turns on, and the output voltage gradually rises. Before entering the steady state, the soft start circuit continues to provide driving energy for the MOS tube, and the voltage of the soft start circuit is gradually reduced. When the system is working normally, since the charging time of the soft start circuit is much longer than the discharge time, the soft start circuit is always kept at a lower potential. (When starting up, the soft-start voltage waveform is shown in Figure 7).
  • this soft-start circuit At steady state, the capacitor C9 in the soft-start circuit discharges much faster than the charging speed, and its ground-to-ground voltage is close to IV (see Figure 7), so the MOS tube drive is completely dependent on the feedback winding energy.
  • the amount of soft-start circuit is insufficient for MOS tube compensation during normal operation.
  • the output performance of the product is easily affected by the transformer process (feedback and input coupling degree), MOS tube conduction threshold, and output external capacitive load.
  • the charging speed is increased, due to the wide input voltage range and large variation of the output load, over-compensation is likely to occur, resulting in shortcomings such as power-on startup current limit and product output short-circuit power consumption.
  • FIG. 2 is a circuit schematic diagram of an embodiment disclosed in the Chinese Patent Application Publication No. CN101997423A, which includes an input filter circuit 11, a soft start circuit 12, a pulse frequency modulation circuit 13, a main power circuit 14, an auxiliary power supply circuit 15, and an output filter.
  • the input filter circuit 11 is composed of a filter capacitor C0, a filter capacitor C1 and a filter inductor L0 to form a ⁇ -type filter circuit.
  • Capacitor CO is connected to the input end of the power supply, and the other end is grounded.
  • the inductor L0 is connected to the input end of the power supply, and the other end is connected to the same end of the transformer.
  • the soft start circuit 12 includes: a voltage dividing resistor R10, a resistor R13, a resistor R14, and a starting capacitor C9.
  • the resistor R10, the resistor R13, and the resistor R14 are connected in series, one end of the series circuit is connected to the power input end, and one end is grounded.
  • One end of the capacitor C9 is connected to the series junction of the resistor R10 and the resistor R13, and the other end is grounded.
  • the working principle is that when the input voltage is connected, the current charges the capacitor C9 through the resistor R10, after the time ⁇ -/ ⁇ ⁇ - ⁇
  • the Vin capacitor voltage reaches the MOS threshold voltage to achieve the soft-start function.
  • the MOS transistor When the MOS transistor is turned off, the feedback winding is reversed, and the potential of the same name is negative. Therefore, the potential of the MOS transistor is also negative.
  • the capacitor C9 is quickly discharged through the resistor R13.
  • the MOS transistor When the MOS transistor is turned on, the input voltage is charged to the capacitor C9 through the resistor R10, because the resistance of the resistor R10 is far greater than the resistance of the resistor R13 (when the value of R10 is decreased, the power consumption of the product output short circuit increases, and the starting current increases, Table 2 exemplifies the shortcomings caused by the decrease in the value of R10. Therefore, the discharge speed is much faster than the charging speed.
  • the pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, an NPN transistor TR2, a PNP transistor TR3, and a positive feedback winding P3.
  • the capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6. One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2.
  • the capacitor C5 is connected in parallel with the resistor R9.
  • One end of the parallel circuit is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3.
  • the capacitor C12 is connected in parallel with the resistor R111.
  • One end of the parallel circuit is connected to the emitter of the transistor TR3, and the other end is connected to the collector of the transistor TR3.
  • Its working principle is: positive feedback winding P3, capacitor C6, resistor Rll branch pass and
  • the main power tube TR1 is coupled to the primary and secondary sides to form a self-excited oscillation circuit, and the control switch tube is turned on and off.
  • the current loop R5 branch and the voltage loop optocoupler branch are regulated by the double-tube drive control circuits TR2 and TR3.
  • the duty cycle makes the product output normal.
  • the main power circuit 14 includes: a transformer primary winding P1, an output winding P2, a MOS transistor TR1, and an output rectifier diode D1 for realizing conversion, transmission, and input and output isolation of power.
  • the auxiliary power supply 15 includes: a positive feedback winding P3 and a diode D3. The working principle is that when the feedback winding has the same name and the diode D3 is turned on, the optical coupling is energized.
  • the output filter circuit 16, including the filter capacitor C3, can also be used in other existing filter circuits, which can be selected according to the relevant technical manual.
  • the isolated optocoupler comprises: an optocoupler OC1, which mainly performs signal transmission and input/output isolation.
  • the error amplifying ADJ circuit 17 includes: a sampling circuit and a signal comparison amplifying circuit.
  • the working principle is as follows: When the output voltage is high, the sampling circuit collects the signal and adjusts the primary current of the optocoupler after the signal is compared with the amplifier circuit, that is, the duty cycle of the product is adjusted by the voltage loop.
  • a sampling current flows through the sampling circuit, the error amplification, the isolation optocoupler, and the pulse frequency modulation PFM circuit to perform negative feedback control on the main power tube in the main power circuit; a soft connection is connected to the output end of the input filter circuit.
  • the startup circuit is connected to the pulse frequency modulation PFM at the other end of the soft start circuit to implement a soft start function of the power supply.
  • Figure 12 compares the normal operation of the power converter with the model PWB4805D and power of 3W and the output voltage waveform with 1000 capacitors, where CH1 is the normal working waveform. , the rise time is about 0.5ms, CH2 is the working waveform with lOOO f capacitor, the rise time is about 3.7ms, the voltage loop is out of control when the output voltage is not formally established. If the MOS tube drive can not get enough compensation, the feedback winding Easy to stop vibration, showing poor start.
  • the starting circuit parameters take into account the key performances such as short-circuit power consumption and starting current.
  • the circuit principle is shown in Figure 2.
  • Table 2 shows the average voltage of capacitor C9 under different load and input voltage conditions when R10 is 332 K ⁇ .
  • the starting current and short-circuit power consumption of the product are small, and the maximum short-circuit power consumption is only 0.792W.
  • Table 2 that when R10 takes a large value, the Vc9 voltage varies widely, and the MOS tube The compensation strength of the drive is unstable, and the compensation is the worst at light load high voltage, and the voltage is only 1.02V.
  • Table 3 shows the average voltage value of C9 under different load and voltage conditions after R10 takes 50 K ⁇ . The table shows that after R10 is reduced, the voltage of capacitor C9 varies greatly under various load conditions, and the maximum voltage is 12.8V. The minimum voltage is only 4.39V, and its compensation is too strong, resulting in large starting current and short circuit power consumption.
  • the junction capacitance of the MOS transistor is about 300 PF.
  • the capacitance is small, and the resistance R11 is only 100 ⁇ . Therefore, when the MOS transistor is turned off, the feedback winding is reversed, and the MOS tube Ciss stores energy on the one hand through the PNP transistor TR3, and On the one hand, the driving branches R11 and C6 are released to the same name end of the feedback winding, and the Vgs potential of the MOS transistor is rapidly pulled down.
  • FIG. 7 shows the voltage waveform on the C9 capacitor (high voltage 72V, output load 0.06A). From this figure, it can be seen that after the MOS transistor is turned on, the capacitor voltage begins to drop. After the steady state, the capacitor voltage is only IV, and its corresponding drive.
  • the charging time constant R10 is increased and the value of R13 is increased, the disadvantages are the same as those described under the non-steady state condition.
  • FIG. 3 shows the output power-down waveform of the PWB4805D-3W product under high-voltage and light-load conditions, after 4KV electrostatic interference at the input end. This waveform shows that when the input terminal is static, the output of the product is easy to be powered down, and when it is serious, it interferes with the restart of the customer system.
  • the design range of the input voltage of the self-oscillating flyback converter and the range of variation of the load is expanding.
  • a self-oscillating flyback converter wherein a DC input signal sequentially passes through an input filter circuit, a main power circuit and an output filter circuit, and outputs a DC signal, the main power circuit includes a main power tube and a main transformer; and the output DC signal passes through
  • the voltage stabilizing circuit, the isolated optocoupler and the drive control circuit perform negative feedback control on the main power tube to achieve a stable output, and further include a soft start circuit with a constant current source connected to the output end of the input filter circuit and Between the drive control circuits; when the self-oscillating flyback converter is in an unsteady state after power-on, the soft start circuit is charged to the soft start circuit with a constant current source through a constant current source, and the soft start voltage rises to the MOS tube threshold After the value is turned on, the MOS tube is turned on.
  • the soft starting capacitor continuously supplies driving compensation to the MOS tube to achieve normal starting; the self-oscillating flyback converter enters the steady state.
  • the constant current source charging speed is balanced with the soft start capacitor discharging speed, and the soft start continues to provide driving compensation to the MOS tube. Steady work properly.
  • the soft start circuit with a constant current source includes a constant current source, a first voltage dividing resistor, a second voltage dividing resistor, and a starting capacitor; the anode of the constant current source is connected to the self-excited a power supply input end of the oscillating flyback converter, wherein the cathode of the constant current source is sequentially connected to a power reference terminal of the self-oscillating flyback converter through the first voltage dividing resistor and the second voltage dividing resistor, the starting capacitor And the first voltage dividing resistor And a series branch of the second voltage dividing resistor is connected in parallel, and a connection point of the first voltage dividing resistor and the second voltage dividing resistor is connected to a gate of the main power tube.
  • the soft start circuit with a constant current source further includes a current limiting resistor, and the anode of the constant current source is connected to the power input end of the self-oscillating flyback converter through the current limiting resistor.
  • the soft start circuit with a constant current source further includes a current limiting resistor connected between a cathode of the constant current source and a connection point between the first voltage dividing resistor and the starting capacitor .
  • the constant current source (D1A) is a single constant current source, or a constant current source in parallel, or a constant current source formed by a combination of a constant current source and a triode, a Zener, and a resistor. , or a constant current source composed of a triode and a resistor.
  • the present invention has the following advantages:
  • One of the advantages of the present invention is to improve the starting capability of the inverter product and the capacitive load capacity, so that the product can start normally and work stably under full load and capacitive load.
  • Table 5 lists the original scheme of the power converter of the type WRF4815P and the power of 6W compared with the capacitive load capacity after adopting the technical scheme of the present invention.
  • the second advantage of the invention is: improving the output voltage stability when the product is lightly loaded (below 10% load), reducing the low energy of the feedback winding under light load (small current at light load, weak coupling energy) Hidden dangers; at the same time, it makes it possible to design a larger rated load current for the self-oscillating flyback converter.
  • the third advantage of the invention is that the process requirements of the transformer are reduced (the feedback and input coupling coefficient requirements are reduced), the requirement for the consistency of the switch tube threshold is reduced, and the product productivity is improved.
  • the technique of the present invention provides suitable compensation for the MOS tube, so that the MOS tube can be reliably driven, and the consistency requirement for the MOS tube conduction threshold is reduced.
  • the fourth advantage of the invention is: improving the anti-interference ability of the product, and effectively solving the output power-down phenomenon.
  • the power converter product with the model PWB4805D and power of 3W is used to supply power to the single-chip microcomputer.
  • the input terminal is 4KV static
  • the single-chip microcomputer is reset, and the power converter output is powered down.
  • the input terminal is 4KV. Static electricity, the output of the inverter is normal, and the MCU works normally.
  • Advantage 5 of the present invention The adaptability of the self-oscillating flyback converter over a wide input voltage range (4:1) is improved. Due to the start of the constant current source, the starting speed and compensation intensity of the product under low voltage, nominal and high voltage are effectively controlled, so that the MOS tube drive is reasonably compensated under the conditions of full input voltage and full load range and large capacitive load. Solved short-circuit power consumption, starting current and other issues. Table 6 lists the short-circuit power consumption and startup performance of a PWB4805D, 3W power converter with a constant current source of 0.5mA. Table 6
  • FIG. 1 is a schematic block diagram of a prior art self-oscillating flyback converter
  • FIG. 2 is a circuit schematic diagram of a self-oscillating flyback converter in the prior art
  • FIG. 3 is a schematic block diagram of a self-oscillating flyback converter of the present invention.
  • FIG. 4 is a schematic circuit diagram of a first embodiment of the present invention.
  • FIG. 5 is a schematic circuit diagram of a portion of a soft start circuit with a constant current source according to an embodiment of the present invention
  • FIG. 6 is a circuit schematic diagram of three soft start circuit portions with a constant current source according to an embodiment of the present invention
  • FIG. 7 is a self-oscillation A voltage waveform diagram of the starting capacitor C9 in the soft start circuit of the flyback converter
  • Figure 8 is a waveform diagram of Vgs driving at low voltage and full load of a power converter of the type PWB4805D and power of 3W;
  • Figure 9 is a waveform diagram of Vgs driving at a light load and high voltage of a power converter of model PWB4805D and power of 3W;
  • Figure 10 is a graph showing the normal temperature threshold of a MOS tube of the type IRFR220;
  • Fig. 11 is a waveform diagram of the output voltage when the power converter of the model PWB4805D and the power of 3W is input with 4KV static electricity;
  • Figure 12 shows the waveform of the output voltage rise of the PWB4805D, 3W power converter without capacitive load and 1000 ⁇ f capacitive load;
  • Figure 13 is a circuit schematic diagram of a parallel form of a constant current source
  • FIG. 14 is a circuit schematic diagram of a composite constant current source composed of a constant current source, a resistor, a Zener tube, and a triode;
  • FIG. 15 is a circuit schematic diagram of a constant current source formed by a PNP transistor and a resistor;
  • Figure 16 is a circuit schematic of a single tube drive control circuit. detailed description
  • the DC input signal sequentially passes through the input filter circuit, the main power circuit and the output filter circuit, and then outputs a DC signal
  • the main power circuit includes a main power tube and a main transformer
  • the output DC signal is sequentially subjected to negative feedback control of the main power tube through the voltage stabilization circuit, the isolation optocoupler and the drive control circuit to achieve stable output, and further includes a soft start circuit with a constant current source, the soft start circuit being connected Between the output end of the input filter circuit and the drive control circuit; the self-oscillating flyback converter is charged to a soft start circuit with a constant current source through a constant current source when it is in an unsteady state, soft start After the voltage rises to the MOS tube threshold, the MOS transistor is turned on.
  • the soft start capacitor continuously supplies driving compensation to the MOS tube to achieve normal startup; After the converter enters the steady state, the constant current source charging speed and the soft start capacitor discharge speed are balanced, and the soft start continues to the MOS tube. For drive compensation, achieve steady state normal operation.
  • the pulse frequency modulation circuit 13 is used as the drive control circuit of the converter.
  • the implementation circuit of the embodiment mainly includes the following components: an input filter circuit 11, a soft start circuit with a constant current source, and a pulse frequency. Modulation circuit 13, main power circuit 14, auxiliary power supply 15, output filter circuit 16.
  • the voltage stabilizing circuit 17, wherein the circuit structure of the input filter circuit 11, the pulse frequency modulation circuit 13, the main power circuit 14, the auxiliary power supply 15, the output filter circuit 16, and the voltage stabilization circuit 17 is the same as that of the circuit shown in FIG.
  • the circuit structure is the same.
  • the difference between this embodiment and the circuit shown in FIG. 2 is that the circuit composition of the constant current source soft start circuit 12 of the circuit shown in FIG. 2 is different from that of the circuit shown in FIG. 2:
  • the main power tube in the main power circuit 14 uses the MOS tube TR1, and the main transformer uses the transformer T1.
  • the transformer T1 includes the primary winding PI, the output winding P2, and the positive feedback winding P3.
  • the input filter circuit 11 includes a filter capacitor C0, a filter capacitor C1 and a filter inductor L0, and the structure thereof is a well-known IT type filter circuit principle structure, which will not be described in detail herein.
  • the soft start circuit with constant current source 12 includes a constant current source D1A, a first voltage dividing resistor R13, a second voltage dividing resistor R14, and a starting capacitor C9.
  • the anode of the constant current source D1A is connected to the power input end, and the cathode of the constant current source D1A is sequentially connected to the power reference end of the self-oscillating flyback converter through the first voltage dividing resistor R13 and the second voltage dividing resistor R14.
  • the starting capacitor C9 is connected in parallel with the series branch of the first voltage dividing resistor R13 and the second voltage dividing resistor R14, and the connection point of the first voltage dividing resistor R13 and the second voltage dividing resistor R14 is opposite to the main power tube
  • the gates are connected. The working principle of this circuit is described in detail below:
  • the voltage of the starting capacitor C9 reaches the threshold voltage of the MOS transistor TR1, and the soft start function is realized.
  • the starting capacitor C9 passes the first partial voltage.
  • the resistor R13 supplies energy to the MOS transistor TR1.
  • the constant current source D1A supplies energy to the starting capacitor C9 in time, and the appropriate constant current source D1A can be selected to meet the energy balance of the starting capacitor C9 charging and discharging, due to the energy compensation of the starting capacitor C9.
  • the positive feedback winding P3 requires only a small amount of energy to complete the self-oscillation process, thereby avoiding the occurrence of intermittent oscillations and starting the product normally.
  • constant current source D1A adopts constant current source of 0.5mA
  • the voltage of starting capacitor C9 under different input voltage and load conditions is compared with Table 2 and Table 4.
  • the voltage on the starting capacitor C9 is basically stable, and the variation range is only 2.4V, which avoids the MOS tube driving over-compensation and under-compensation caused by the difference in the value of the resistor R10 in the original scheme.
  • the pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, a NPN transistor TR2, and a PNP transistor TR3.
  • the capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6.
  • One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2.
  • the capacitor C5 is connected in parallel with the resistor R9.
  • One end of the parallel current is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3.
  • Capacitor C12 is connected in parallel with resistor R111.
  • the working principle is as follows:
  • the positive feedback winding P3, the capacitor C6, and the resistor R11 branch are coupled with the primary and secondary sides of the main power transformer T1 to form a self-excited oscillation circuit, and the control MOS transistor TR1 is turned on and off; and the current loop resistor R5 branch
  • the voltage loop optocoupler OC1 branch adjusts the on-duty of the switch tube by the double-tube drive control circuit of the transistor TR2 and the transistor TR3, so that the product output is normal.
  • Auxiliary power supply 15, including positive feedback winding P3, diode D3, works on the principle that the feedback winding has the same name as the timing diode D3 is conducting, providing energy for the optocoupler.
  • the output filter circuit 16, including the capacitor C3, can also be used with other existing filter circuits, which can be selected according to the relevant technical manual.
  • the voltage stabilizing circuit 17 mainly includes a voltage regulator ADJ, which is connected to the main power circuit, the auxiliary power source 15 and the pulse frequency modulation circuit 13 through the optocoupler OC1, and will not be described herein.
  • FIG. 6 is a portion of a soft start circuit with a constant current source according to a third embodiment of the present invention, which differs from the circuit shown in the second embodiment only in limiting the maximum current in the constant current source soft start circuit of the embodiment.
  • the access position of the flow resistor R1A changes, and the current limiting resistor R1A is connected between the cathode of the constant current source D1A and the connection point of the first voltage dividing resistor R13 and the starting capacitor C9.
  • the above constant current source may have multiple compositions, which may be a parallel form of a constant current source in the prior art, as shown in FIG. 13; a constant current source and a triode, a Zener, and a resistor in the prior art
  • the drive control circuit of the converter is as shown in FIG. 16, by the transistor TR2, the resistor R6.
  • the single-tube drive control circuit composed of the capacitor C5 and the resistor R9 replaces the pulse frequency modulation circuit and the like in the above embodiments.

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Abstract

A self-oscillatory flyback converter is disclosed. The direct-current input signals, from a power input terminal of the self-oscillatory flyback converter input, turns into direct-current output signals after passing through an input filter circuit (11), a master power circuit (14) and an output filter circuit (16) in turn. The master power circuit (14) includes a master power tube (TR1) and a master transformer (T1).The direct-current output signals controls the master power tube (TR1) with negative feedback to stable the output via a voltage-stabilizing circuit (17), an isolating optical coupler (OC1) and a drive control circuit (13). The self-oscillatory flyback converter also includes a soft-start circuit (12) with a constant current source. The soft start circuit (12) is connected between the output terminal of the input filter circuit (11) and the drive control circuit (13). The self-oscillatory flyback converter achieves the soft-start function while providing suitable drive compensation in the startup phase and steady state phase of the product, improving the capacity of resisting disturbance and stability, and expanding the design range of the input voltage of the self-oscillatory flyback converter and the variation range of the load.

Description

一种自激振荡反激变换器 技术领域  Self-oscillating flyback converter
本发明涉及一种自激振荡反激变换器,特别涉及一种带恒流源软启动电路的 自激振荡反激变换器。 背景技术  The present invention relates to a self-oscillating flyback converter, and more particularly to a self-oscillating flyback converter with a constant current source soft start circuit. Background technique
自振荡反激变换器 (Ringing Choke Converter) 因其设计成本较低, 具有较 强的市场竞争力而备受设计者的青睐; 然而其输出特性的好坏, 很大程度上取决 于各分立元器件的一致性, 当变压器、 主功率管、 控制三极管、 光耦等元器件的 一致性越好, 其输出性能也越稳定。 因此降低自激振荡反激变换电路(RCC)对 分立器件一致性的苛刻要求,提高其输出电压稳定性成为设计者考虑的重点。 目 前设计者多采用偏置补偿电路 (控制三极管基极添加关断补偿), 其输出电压的 稳定性得到一定程度的提高, 然而该方案调试难度大, 产品的一致性依然较差, 尤其在输出功率增加、 输出容性负载增大时影响更为明显。  The Ringing Choke Converter is favored by designers because of its low design cost and strong market competitiveness. However, its output characteristics depend largely on discrete components. The consistency of the device, when the consistency of the transformer, main power tube, control transistor, optocoupler and other components is better, the output performance is more stable. Therefore, reducing the stringent requirements of the self-oscillating flyback converter circuit (RCC) for discrete device uniformity and improving its output voltage stability have become the focus of designers. At present, designers often use offset compensation circuits (control the base of the triode to add turn-off compensation), and the stability of the output voltage is improved to some extent. However, the scheme is difficult to debug, and the consistency of the product is still poor, especially in the output. The effect is more obvious when the power is increased and the output capacitive load is increased.
中国发明专利申请公开说明书 CN101997423A 中公开了一种自激振荡反激 变换器(RCC)较为理想的电路形式, 如图 1所示, 该自激振荡反激变换器主要 包括滤波部分、 软启动部分、 MOS管、 变压器、 脉冲频率调制部分(PFM)、 辅 助电源、 隔离光耦、 稳压输出回路部分。 输入电量经变压器连接输出回路部分, 软启动部分连接 MOS管的栅极, MOS管的栅极还接脉冲频率调制部分,辅助电 源连接脉冲频率调制部分,脉冲频率调制部分和稳压输出回路部分之间接基准放 大部分、 隔离光耦, 形成电压负反馈回路。  A preferred circuit form of a self-oscillating flyback converter (RCC) is disclosed in the Chinese Patent Application Publication No. CN101997423A. As shown in FIG. 1, the self-oscillating flyback converter mainly includes a filtering part and a soft start part. , MOS tube, transformer, pulse frequency modulation section (PFM), auxiliary power supply, isolated optocoupler, regulated output loop section. The input power is connected to the output circuit part through the transformer, the soft start part is connected to the gate of the MOS tube, the gate of the MOS tube is also connected to the pulse frequency modulation part, the auxiliary power supply is connected to the pulse frequency modulation part, the pulse frequency modulation part and the regulated output circuit part are The indirect reference amplification section and the isolated optocoupler form a voltage negative feedback loop.
上述反激变换器的软启动电路,提高了产品启动时输出电压的稳定性, 其工 作原理为: 输入上电后, 软启动电路电压升高, MOS 管导通, 输出电压逐渐上 升, 在系统进入稳态前, 软启动电路继续为 MOS管提供驱动能量, 同时软启动 电路电压逐渐降低; 当系统正常工作后, 由于软启动电路充电时间远大于放电时 间, 软启动电路始终保持在较低电位 (开机启动时, 软启动电压波形见图 7)。 该软启动电路的缺点在于: 稳态时, 软启动电路中电容 C9放电速度远快于充电 速度, 其对地电压接近 IV (见图 7), 因此 MOS管驱动完全依赖于反馈绕组能 量, 软启动电路正常工作时对 MOS管补偿不足, 产品输出性能容易受变压器工 艺(反馈与输入耦合程度)、 MOS管导通门限、 输出外接容性负载等影响。 当增 大充电速度时, 受输入电压范围宽和输出负载变化大的影响, 容易出现过补偿, 导致电源开机启动限流和产品输出短路功耗大等缺点。 The soft start circuit of the flyback converter improves the stability of the output voltage at the start of the product. The working principle is as follows: After the input is powered on, the voltage of the soft start circuit rises, the MOS transistor turns on, and the output voltage gradually rises. Before entering the steady state, the soft start circuit continues to provide driving energy for the MOS tube, and the voltage of the soft start circuit is gradually reduced. When the system is working normally, since the charging time of the soft start circuit is much longer than the discharge time, the soft start circuit is always kept at a lower potential. (When starting up, the soft-start voltage waveform is shown in Figure 7). The disadvantages of this soft-start circuit are: At steady state, the capacitor C9 in the soft-start circuit discharges much faster than the charging speed, and its ground-to-ground voltage is close to IV (see Figure 7), so the MOS tube drive is completely dependent on the feedback winding energy. The amount of soft-start circuit is insufficient for MOS tube compensation during normal operation. The output performance of the product is easily affected by the transformer process (feedback and input coupling degree), MOS tube conduction threshold, and output external capacitive load. When the charging speed is increased, due to the wide input voltage range and large variation of the output load, over-compensation is likely to occur, resulting in shortcomings such as power-on startup current limit and product output short-circuit power consumption.
图 2是中国发明专利申请公开说明书 CN101997423A中公开的一个实施例的 电路原理图, 包括输入滤波电路 11、 软启动电路 12、 脉冲频率调制电路 13、 主 功率电路 14、 辅助电源电路 15、 输出滤波电路 16、 隔离光耦 OCl、 误差放大 ADJ电路 17。 输入滤波电路 11由滤波电容 C0、 滤波电容 C1和滤波电感 L0组 成, 形成了 π型滤波电路。 电容 CO接电源输入端, 另一端接地, 电感 L0接电 源输入端, 另一端接变压器同名端, 电容 C1的一端接变压器同名端, 另一端接 地。 也可采用其它现有滤波电路, 具体可按有关技术手册选定。 软启动电路 12 包括:分压电阻 R10、 电阻 R13、 电阻 R14和启动电容 C9。 电阻 R10、 电阻 R13、 电阻 R14依次串联, 串联电路的一端接电源输入端, 一端接地。 电容 C9的一端 接入电阻 R10与电阻 R13的串联结点, 另一端接地。 其工作原理为当接入输入 电压时, 电流经电阻 R10对电容 C9充电, 经过时间^ -/^^ ^^-^^后  2 is a circuit schematic diagram of an embodiment disclosed in the Chinese Patent Application Publication No. CN101997423A, which includes an input filter circuit 11, a soft start circuit 12, a pulse frequency modulation circuit 13, a main power circuit 14, an auxiliary power supply circuit 15, and an output filter. The circuit 16, the isolated optocoupler OCl, and the error amplifying ADJ circuit 17. The input filter circuit 11 is composed of a filter capacitor C0, a filter capacitor C1 and a filter inductor L0 to form a π-type filter circuit. Capacitor CO is connected to the input end of the power supply, and the other end is grounded. The inductor L0 is connected to the input end of the power supply, and the other end is connected to the same end of the transformer. One end of the capacitor C1 is connected to the same name end of the transformer, and the other end is grounded. Other existing filter circuits can also be used, which can be selected in accordance with the relevant technical manual. The soft start circuit 12 includes: a voltage dividing resistor R10, a resistor R13, a resistor R14, and a starting capacitor C9. The resistor R10, the resistor R13, and the resistor R14 are connected in series, one end of the series circuit is connected to the power input end, and one end is grounded. One end of the capacitor C9 is connected to the series junction of the resistor R10 and the resistor R13, and the other end is grounded. The working principle is that when the input voltage is connected, the current charges the capacitor C9 through the resistor R10, after the time ^ -/^^ ^^-^^
Vin 电容电压达到 MOS门限电压, 实现开机软启动功能; 当 MOS管关断时, 反馈 绕组反向, 同名端电位为负, 因此 MOS管电位亦为负, 此时电容 C9通过电阻 R13快速放电, 当 MOS管导通时, 输入电压通过电阻 R10向电容 C9充电, 由 于电阻 R10的阻值远远大于电阻 R13的阻值 (R10取值减小时, 产品输出短路 功耗增加, 启动电流增大, 表二例举了 R10取值减小时带来的不足), 因此放电 速度远快于充电速度, 经过一段时间后电容 C9能量释放完毕。 脉冲频率调制电 路 13包括: 电阻 R6、 电阻 R9、 电阻 Rll、 电阻 R111,电容 C5、 电容 C6、 电容 C12, NPN型三极管 TR2、 PNP型三极管 TR3、 正反馈绕组 P3。 电容 C6与电阻 Rll、 电阻 Rlll、 电阻 R6依次串联, 串联电路的一端接入正反馈绕组 P3的同 名端, 另一端接入三极管 TR2的集电极。 电容 C5与电阻 R9并联, 并联电路的 一端接入三极管 TR2的基极, 另一端接入三极管 TR3的集电极。 电容 C12与电 阻 R111 并联, 并联电路的一端接入三极管 TR3 的发射极, 另一端接入三极管 TR3的集电极。 其工作原理为: 正反馈绕组 P3、 电容 C6、 电阻 Rll支路通过与 主功率管 TR1 原副边耦合, 构成自激振荡回路, 控制开关管导通关断; 同时电 流环 R5支路、 电压环光耦支路通过双管驱动控制电路 TR2、 TR3调节开关管导 通占空比, 使产品输出正常。 主功率电路 14包括: 变压器原边绕组 Pl、 输出绕 组 P2, MOS管 TR1,输出整流二极管 Dl, 实现电源能量的转换、 传递以及输入 与输出隔离。 辅助电源 15包括: 正反馈绕组 P3、 二极管 D3, 其工作原理为, 反馈绕组同名端为正时, 二极管 D3导通, 为光耦提供能量。 输出滤波电路 16, 包括滤波电容 C3, 也可采用其它现有滤波电路, 具体可按有关技术手册选定。 隔离光耦包括: 光耦 OC1 , 其主要完成信号的传递和输入输出隔离作用。 误差 放大 ADJ电路 17包括: 取样电路、 信号比较放大电路。 其工作原理为: 输出电 压漂高时,取样电路采集信号经信号比较放大电路后调节光耦原边电流, 即通过 电压环调节产品的占空比。在电源的输出端有一取样电流流经取样电路、误差放 大、 隔离光耦、 脉冲频率调制 PFM电路后对主功率电路中的主功率管进行负反 馈控制; 在输入滤波电路的输出端连接有一软启动电路, 该软启动电路另一端与 脉冲频率调制 PFM连接以实现电源的软启动功能。 The Vin capacitor voltage reaches the MOS threshold voltage to achieve the soft-start function. When the MOS transistor is turned off, the feedback winding is reversed, and the potential of the same name is negative. Therefore, the potential of the MOS transistor is also negative. At this time, the capacitor C9 is quickly discharged through the resistor R13. When the MOS transistor is turned on, the input voltage is charged to the capacitor C9 through the resistor R10, because the resistance of the resistor R10 is far greater than the resistance of the resistor R13 (when the value of R10 is decreased, the power consumption of the product output short circuit increases, and the starting current increases, Table 2 exemplifies the shortcomings caused by the decrease in the value of R10. Therefore, the discharge speed is much faster than the charging speed. After a period of time, the energy of the capacitor C9 is released. The pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, an NPN transistor TR2, a PNP transistor TR3, and a positive feedback winding P3. The capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6. One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2. The capacitor C5 is connected in parallel with the resistor R9. One end of the parallel circuit is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3. The capacitor C12 is connected in parallel with the resistor R111. One end of the parallel circuit is connected to the emitter of the transistor TR3, and the other end is connected to the collector of the transistor TR3. Its working principle is: positive feedback winding P3, capacitor C6, resistor Rll branch pass and The main power tube TR1 is coupled to the primary and secondary sides to form a self-excited oscillation circuit, and the control switch tube is turned on and off. At the same time, the current loop R5 branch and the voltage loop optocoupler branch are regulated by the double-tube drive control circuits TR2 and TR3. The duty cycle makes the product output normal. The main power circuit 14 includes: a transformer primary winding P1, an output winding P2, a MOS transistor TR1, and an output rectifier diode D1 for realizing conversion, transmission, and input and output isolation of power. The auxiliary power supply 15 includes: a positive feedback winding P3 and a diode D3. The working principle is that when the feedback winding has the same name and the diode D3 is turned on, the optical coupling is energized. The output filter circuit 16, including the filter capacitor C3, can also be used in other existing filter circuits, which can be selected according to the relevant technical manual. The isolated optocoupler comprises: an optocoupler OC1, which mainly performs signal transmission and input/output isolation. The error amplifying ADJ circuit 17 includes: a sampling circuit and a signal comparison amplifying circuit. The working principle is as follows: When the output voltage is high, the sampling circuit collects the signal and adjusts the primary current of the optocoupler after the signal is compared with the amplifier circuit, that is, the duty cycle of the product is adjusted by the voltage loop. At the output end of the power supply, a sampling current flows through the sampling circuit, the error amplification, the isolation optocoupler, and the pulse frequency modulation PFM circuit to perform negative feedback control on the main power tube in the main power circuit; a soft connection is connected to the output end of the input filter circuit. The startup circuit is connected to the pulse frequency modulation PFM at the other end of the soft start circuit to implement a soft start function of the power supply.
上述电路的缺点在于:  The disadvantages of the above circuits are:
1 ) 产品启动阶段, 带满载、 尤其是大容性负载时, 不能进入稳态, 产品输出异 常; 上述电路开机后, 需要经过一段时间振荡, 产品才能进入稳态; 在非稳态时 期, 反馈绕组能量较弱, 此时依赖电容 C9补偿能量, 然而电容 C9电压从 MOS 管导通第一周期后开始下降(电容 C9开机电压波形见图 7, MOS管导通后其电 压逐渐下降), 提供能量越来越弱, 因此该电路在满负载、 轻负载、 尤其是大容 性负载时, 容易振荡, 表现为输出电压偏低。 当输出带容性负载时, 产品输出电 压上升上升时间变长, 图 12对比了型号为 PWB4805D、 功率为 3W的电源变换 器正常工作和带 1000 电容时的输出电压波形, 其中 CH1为正常工作波形, 上升时间约为 0.5ms, CH2为带 lOOO f 电容的工作波形, 上升时间约为 3.7ms, 输出电压未正式建立时电压环处于失控状态, 此时 MOS管驱动若不能得到足够 补偿, 反馈绕组容易停振, 表现为启动不良。 以下详细介绍该阶段过程: 以现有 PWB4805D, 功率为 3W的电源变换器为例, 其启动电路参数兼顾了短路功耗、 启动电流等关键性能, 电路原理见图 2, 该软启动电路 12中各参数取值如下: R10=332 K Ω ,R13=3.3 K Ω ,R14=150 K Ω ,C9=1 μ f, 控制电路 13中反馈绕组驱动 支路 Rll=100 Ω, C6=4700PF,主功率电路 14部分,其变压器 T1各参数为: Np=25, Ns=7, Nf=8。 产品在 MOS 管导通的初始阶段, 反馈绕组电压较小, 该阶段 G 极电位几乎为零, 因此非稳态时期放电时间常数可近似为 T2 R13*C9 3.3ms, 又充电时间常数 Tl R14*C9=332ms, 所以非稳态时期, MOS管导通后电容 C9 电位逐渐下降, 图 7显示了 (测试条件: 轻载 Io=0.06A, 高压 72V测试) 电容 C9上的电压波形, MOS管导通后, 电容电压逐渐下降。 上述参数 R10取值为 332 K Ω , 兼顾了产品的开机启动电流和输出短路功耗。 表一例举了输入电压 18V、 48V、 72V, 满载 Io=0.6A时的电源开机启动最小限流值和各点条件下的输 出短路功耗。 1) During the start-up phase of the product, with full load, especially for large capacitive loads, it cannot enter the steady state, and the output of the product is abnormal. After the above circuit is turned on, it needs to oscillate for a period of time before the product can enter the steady state; during the non-steady state, the feedback The winding energy is weak. At this time, the capacitor C9 is used to compensate the energy. However, the voltage of the capacitor C9 starts to decrease after the first period of the MOS transistor is turned on (the waveform of the capacitor C9 is shown in Figure 7, and the voltage of the MOS transistor gradually decreases after being turned on). The energy is getting weaker and weaker, so the circuit is easy to oscillate at full load, light load, especially large capacitive load, and the output voltage is low. When the output has a capacitive load, the product output voltage rises and the rise time becomes longer. Figure 12 compares the normal operation of the power converter with the model PWB4805D and power of 3W and the output voltage waveform with 1000 capacitors, where CH1 is the normal working waveform. , the rise time is about 0.5ms, CH2 is the working waveform with lOOO f capacitor, the rise time is about 3.7ms, the voltage loop is out of control when the output voltage is not formally established. If the MOS tube drive can not get enough compensation, the feedback winding Easy to stop vibration, showing poor start. The following describes the phase of the process in detail: Taking the existing PWB4805D power converter with power of 3W as an example, the starting circuit parameters take into account the key performances such as short-circuit power consumption and starting current. The circuit principle is shown in Figure 2. The soft-start circuit 12 The values of each parameter are as follows: R10=332 K Ω , R13=3.3 K Ω , R14=150 K Ω , C9=1 μ f, feedback winding drive in control circuit 13 The branch Rll=100 Ω, C6=4700PF, the main power circuit 14 part, the parameters of the transformer T1 are: Np=25, Ns=7, Nf=8. In the initial stage of MOS tube conduction, the feedback winding voltage is small, and the G-pole potential is almost zero in this stage. Therefore, the discharge time constant in the unsteady period can be approximated as T2 R13*C9 3.3ms, and the charging time constant Tl R14* C9=332ms, so the non-steady state period, after the MOS transistor is turned on, the potential of the capacitor C9 gradually decreases. Figure 7 shows (test conditions: light load Io=0.06A, high voltage 72V test) voltage waveform on capacitor C9, MOS tube guide After the pass, the capacitor voltage gradually decreases. The above parameter R10 takes 332 K Ω, taking into account the product's startup start current and output short-circuit power consumption. The table gives an example of the minimum current limit value of the power-on startup and the output short-circuit power consumption at each point when the input voltage is 18V, 48V, 72V, full load Io=0.6A.
表一 Table I
Figure imgf000006_0001
Figure imgf000006_0001
表二例举了 R10取 332 K Ω时, 电容 C9在不同负载和输入电压条件下的平 均电压。  Table 2 shows the average voltage of capacitor C9 under different load and input voltage conditions when R10 is 332 K Ω.
表二 Table II
Figure imgf000006_0002
Figure imgf000006_0002
从表一可以看出,产品启动电流、短路功耗均较小,最大短路功耗仅 0.792W; 从表二可以看出, 当 R10取值较大时, Vc9电压变化范围大, 对 MOS管驱动的 补偿强度不稳定, 轻载高压时补偿最差, 电压仅 1.02V。 当 R10取值变小时, 其 充电时间常数 T1=R10*C9减小, 电容 C9的能量补偿增强, 由于减小 R10, 其补 偿强度不容易控制,尤其在输入电压、负载变化范围大的产品难以兼顾各点的补 偿强度, 其带来的缺点是: 1 ) 开机启动电流变大 2) 短路功耗增大, 尤其在高 压时短路功耗增加明显, 严重时产品被烧毁。表三例举了将 R10由 332 K Ω改为 50 Κ Ω后, 产品的输出短路功耗和启动电流。 表四例举了该条件下电容 C9的平 均电压。 As can be seen from Table 1, the starting current and short-circuit power consumption of the product are small, and the maximum short-circuit power consumption is only 0.792W. It can be seen from Table 2 that when R10 takes a large value, the Vc9 voltage varies widely, and the MOS tube The compensation strength of the drive is unstable, and the compensation is the worst at light load high voltage, and the voltage is only 1.02V. When the value of R10 becomes smaller, the charging time constant T1=R10*C9 decreases, and the energy compensation of the capacitor C9 is enhanced. Since the R10 is reduced, the compensation intensity is not easy to control, especially in products with large input voltage and load variation range. Taking into account the compensation strength of each point, the disadvantages are: 1) The startup current becomes larger 2) The short-circuit power consumption increases, especially at high When the voltage is short, the power consumption of the short circuit increases significantly. In severe cases, the product is burned. Table 3 shows the output short-circuit power consumption and starting current of the product after changing R10 from 332 K Ω to 50 Ω Ω. Table 4 exemplifies the average voltage of the capacitor C9 under this condition.
表三 Table 3
Figure imgf000007_0001
Figure imgf000007_0001
表四 Table 4
Figure imgf000007_0002
Figure imgf000007_0002
从表三可以看出: 1 ) 短路功耗均较原方案增加, 高压时达 3.96W, 长时间 短路将损坏产品 2) 启动电流增加明显, 尤其高压时启动电流较原方案增加近 1.7倍, 此时可能导致客户使用时启动限流。 表四为 R10取值 50 K Ω后, 不同负 载和电压条件下, C9的平均电压值, 该表显示减小 R10后, 电容 C9在各负载 条件下电压变化较大, 最大电压为 12.8V, 最小电压仅 4.39V, 其补偿过强, 导 致产品启动电流大和短路功耗大。 在原方案基础上, 将 R13 由 3.3 K Ω改为 7.5 Κ Ω后,标称满载时产品开机启动不良,因此放电时间常数允许变化较小。综上, 改变 R10取值, 其对 MOS管驱动补偿稳定性差, 容易出现过补偿和欠补偿。  It can be seen from Table 3: 1) The short-circuit power consumption is increased compared with the original scheme, and the high voltage is 3.96W. The long-time short-circuit will damage the product. 2) The starting current increases obviously, especially when the high-voltage starting current is nearly 1.7 times higher than the original scheme. At this point, the customer may start the current limit when using it. Table 4 shows the average voltage value of C9 under different load and voltage conditions after R10 takes 50 K Ω. The table shows that after R10 is reduced, the voltage of capacitor C9 varies greatly under various load conditions, and the maximum voltage is 12.8V. The minimum voltage is only 4.39V, and its compensation is too strong, resulting in large starting current and short circuit power consumption. On the basis of the original scheme, after changing R13 from 3.3 K Ω to 7.5 Κ Ω, the product starts to start poorly at the full load, so the discharge time constant is allowed to change little. In summary, changing the value of R10, its compensation stability for MOS tube drive is poor, and over-compensation and under-compensation are prone to occur.
2) 稳态时, 高压轻载条件下, 软启动电路中电容 C9放电速度快于充电速 度, 其对地电压接近 IV (见图 7, 轻载 Io=0.06A, 高压 72V测试), 因此 MOS 管驱动几乎完全依赖于反馈绕组能量, 此时产品输出性能的优劣取决于两点: a、 反馈绕组与原边绕组的耦合性能; b、输出负载。 当反馈绕组与输入耦合变差时, 其耦合驱动能量变差, 特别是输入电压跳变条件下, 驱动能量不足, 容易导致产 品振荡; 当输出负载变小时 (10%负载以下), 反馈绕组能量大大下降, 其驱动 能量(耦合电流)也降低, 产品驱动进入间歇式振荡, 表现为输出电压偏低。 以 下详细介绍高压轻载时, 产品的工作过程: 以现有 PWB4805D、功率为 3W的电 源变换器为例, 其启动电路参数兼顾了短路功耗、 启动电流等关键性能, 电路原 理见图 2, 该软启动电路 12 中各参数取值如下: R10=332 Κ Ω ,R13=3.3 K Ω ,R14=150 Κ Ω ,C9=1 μ f, 控制电路 13 中反馈绕组驱动支路 Rll=100 Ω, C6=4700PF, 主功率电路 14部分, 其变压器 T1各参数为: Np=25, Ns=7, Nf=8。 稳态条件下, 当反馈绕组反向时 (对应 MOS 管关断) 同名端对地电压2) At steady state, under high-voltage and light-load conditions, the capacitor C9 in the soft-start circuit discharges faster than the charging speed, and its ground voltage is close to IV (see Figure 7, light-load Io=0.06A, high-voltage 72V test), so MOS The tube drive depends almost entirely on the feedback winding energy. At this time, the output performance of the product depends on two points: a. The coupling performance of the feedback winding and the primary winding; b. Output load. When the feedback winding and the input coupling deteriorate, the coupled drive energy deteriorates, especially under the condition of input voltage jump, the driving energy is insufficient, which easily leads to product oscillation; When the output load becomes small (10% load or less), the feedback winding energy Greatly reduced, its drive The energy (coupling current) is also reduced, and the product is driven into intermittent oscillation, which is manifested by a low output voltage. The following is a detailed description of the working process of the product under high voltage and light load: Take the existing PWB4805D power converter with power of 3W as an example. The starting circuit parameters take into account the key performances such as short-circuit power consumption and starting current. The circuit principle is shown in Figure 2. The values of the parameters in the soft start circuit 12 are as follows: R10=332 Κ Ω , R13=3.3 K Ω , R14=150 Κ Ω , C9=1 μ f, the feedback winding drive branch Rll=100 Ω in the control circuit 13 C6=4700PF, the main power circuit 14 part, its transformer T1 parameters are: Np=25, Ns=7, Nf=8. Under steady-state conditions, when the feedback winding is reversed (corresponding to the MOS transistor is turned off), the same name terminal-to-ground voltage
V *N fV *N f
V f = -— - ~ ^ =-4.375V,由于电容 C6和 MOS管 GS结电容 Ciss (采用 IRFR220 f P Ns V f = - - - ~ ^ = -4.375V, due to capacitor C6 and MOS tube GS junction capacitance Ciss (using IRFR220 f P Ns
MOS管其结电容 Ciss约为 300PF) 容值小, 且电阻 R11取值仅为 100 Ω, 因此 MOS管关断时, 反馈绕组反向, MOS管 Ciss储存能量一方面通过 PNP三极管 TR3释放,另一方面通过驱动支路 R11、C6向反馈绕组同名端释放,表现为 MOS 管 Vgs电位被迅速拉低, 实测波形见图 8 (测试条件: Vin=18V, Io=0.6A), 从 波形可以看出, MOS管关断阶段 Vgs«-0.64V。 因此关断阶段电容 C9通过电阻 R13向 G极放电; 产品开关频率越高, 占空比越小, 即单位时间内关断时间长, 因此高频条件 C9放电时间长, 其自身电位也较低, 图 7测试了 C9电容上的电 压波形 (高压 72V, 输出负载 0.06A), 从该图可以看出 MOS管导通后, 电容电 压开始下降, 稳态后电容电压只有 IV, 其对应的驱动波形见图 9, 其中 T« 1.07 μ s、有效导通时间 Τοη^Ο.01 μ s (驱动电压大于开启门限为有效高电平,即 Vgs〉 3.3V), D=0.0093, 即该条件下 MOS管 99.06%的时间处于关断状态, 因此有效 放电时间常数 T2 R13*C9= 3.3ms,又其充电时间常数 Tl=R10*C9=332ms,故 T2 < <T1。 当减小充电时间常数 R10、 增加 R13值, 其带来的缺点同非稳态条件 下所述一致。  The junction capacitance of the MOS transistor is about 300 PF. The capacitance is small, and the resistance R11 is only 100 Ω. Therefore, when the MOS transistor is turned off, the feedback winding is reversed, and the MOS tube Ciss stores energy on the one hand through the PNP transistor TR3, and On the one hand, the driving branches R11 and C6 are released to the same name end of the feedback winding, and the Vgs potential of the MOS transistor is rapidly pulled down. The measured waveform is shown in Fig. 8 (test condition: Vin=18V, Io=0.6A). Out, the MOS tube is turned off during the Vgs«-0.64V. Therefore, the capacitor C9 in the shutdown phase is discharged to the G pole through the resistor R13; the higher the switching frequency of the product, the smaller the duty ratio, that is, the off time per unit time is long, so the high frequency condition C9 has a long discharge time and its own potential is also low. Figure 7 shows the voltage waveform on the C9 capacitor (high voltage 72V, output load 0.06A). From this figure, it can be seen that after the MOS transistor is turned on, the capacitor voltage begins to drop. After the steady state, the capacitor voltage is only IV, and its corresponding drive. The waveform is shown in Figure 9, where T« 1.07 μs, effective on-time Τοη^Ο.01 μs (drive voltage is greater than the turn-on threshold is active high, ie Vgs > 3.3V), D=0.0093, ie under this condition The MOS tube is in the off state at 99.06%, so the effective discharge time constant T2 R13*C9=3.3ms, and its charging time constant Tl=R10*C9=332ms, so T2 < <T1. When the charging time constant R10 is increased and the value of R13 is increased, the disadvantages are the same as those described under the non-steady state condition.
3 )当外界因素干扰控制信号(例如静电),开关管在较短时间内被彻底关闭, 当干扰消失后, 产品需要经过软启动充电、 MOS 管导通至输出正常 (一般为输 出上升时间 0.5ms) 两段时间后方能进入稳态工作, 此时输出表现为掉电现象。 图 11测试了 PWB4805D-3W产品在高压、轻负载条件下,输入端 4KV静电干扰 后, 输出掉电波形, 该波形显示输入端静电时, 产品输出容易掉电, 严重时干扰 客户系统重启。以下详细介绍掉电原因:电路原理见图 2,启动 12部分, R10=332 K Ω, C9取 1 μ f, R13=3.3 K Ω; 开关管采用 IRFR220,查规格书其门限为 2V-4V, 实测其 导通 门 限 V(th)=3.3V, 因 此 Vin=36V 时最快启 动 时 间 t = -R10*C9*ln(l -^^) =31.9 lms, 由于 R13的限流作用, 实际启动时间会略微 3) When external factors interfere with the control signal (such as static electricity), the switch tube is completely turned off in a short time. When the interference disappears, the product needs to be charged by soft start, and the MOS transistor is turned on until the output is normal (generally the output rise time is 0.5). Ms) After two periods of time, the steady state operation can be entered, and the output appears to be powered down. Figure 11 shows the output power-down waveform of the PWB4805D-3W product under high-voltage and light-load conditions, after 4KV electrostatic interference at the input end. This waveform shows that when the input terminal is static, the output of the product is easy to be powered down, and when it is serious, it interferes with the restart of the customer system. The following is a detailed description of the cause of power failure: the circuit principle is shown in Figure 2, starting part 12, R10=332 K Ω, C9 takes 1 μ f, R13=3.3 K Ω; Switch tube adopts IRFR220, check the specification whose threshold is 2V-4V, and its conduction threshold V(th)=3.3V, so the fastest when Vin=36V Start-up time t = -R 10 *C 9 *ln(l -^^) =31.9 lms, due to the current limit of R13, the actual start-up time will be slightly
Vin  Vin
长于上述计算值, 实测约为 35ms, 因此当静电干扰关断 MOS管后, 产品若经过 软启动电路重新启动, 则关断时间必然变长, 产品输出表现为掉电现象。 该方案 减小 R10取值, 可以增强软启动补偿能力, 减小掉电风险, 然而受输入电压和 负载变化范围大的影响, 减小 R10难以兼容不同输入电压点和负载条件下的补 偿, 补偿强度过大时带来短路功耗增加、 启动电流增大, 补偿过小时又不能解决 上述问题。 发明内容 It is longer than the above calculated value, and the actual measurement is about 35ms. Therefore, when the static electricity is interrupted by the static EMI, the shutdown time will inevitably become longer and the output of the product will be powered down. The scheme reduces the value of R10, which can enhance the soft start compensation capability and reduce the risk of power failure. However, due to the influence of the input voltage and load variation range, it is difficult to reduce the compensation of R10 for different input voltage points and load conditions. When the strength is too large, the short-circuit power consumption increases, the starting current increases, and the compensation is too small to solve the above problem. Summary of the invention
本发明的目的是提供一种自激振荡反激变换器,能够在实现软启动功能的同 时, 为产品的启动和稳态阶段提供合适的驱动补偿, 提高产品抗干扰能力和稳定 性, 同时扩大自激振荡反激变换器输入电压的设计范围和负载的变化范围。  SUMMARY OF THE INVENTION It is an object of the present invention to provide a self-oscillating flyback converter capable of providing a suitable drive compensation for starting and steady-state phases of a product while achieving a soft start function, thereby improving product anti-jamming capability and stability, and at the same time expanding The design range of the input voltage of the self-oscillating flyback converter and the range of variation of the load.
本发明的目的是通过以下技术措施实现的:  The object of the invention is achieved by the following technical measures:
一种自激振荡反激变换器, 直流输入信号依次经过输入滤波电路、主功率电 路和输出滤波电路后输出直流信号, 主功率电路包括主功率管和主变压器; 所述 输出的直流信号依次通过稳压电路、隔离光耦和驱动控制电路对主功率管进行负 反馈控制以实现稳定输出,还包括带恒流源的软启动电路, 该软启动电路连接在 所述输入滤波电路的输出端和所述驱动控制电路之间;所述自激振荡反激变换器 上电后处于非稳态时,通过恒流源向带恒流源的软启动电路充电, 软启动电压上 升到 MOS管门限值后 MOS管导通, 由于恒流源充电速度与软启动电容放电速 度平衡, 软启动电容持续不断向 MOS管提供驱动补偿, 实现启动正常; 所述自 激振荡反激变换器进入稳态以后,恒流源充电速度与软启动电容放电速度维持平 衡, 软启动持续向 MOS管提供驱动补偿, 实现稳态正常工作。  A self-oscillating flyback converter, wherein a DC input signal sequentially passes through an input filter circuit, a main power circuit and an output filter circuit, and outputs a DC signal, the main power circuit includes a main power tube and a main transformer; and the output DC signal passes through The voltage stabilizing circuit, the isolated optocoupler and the drive control circuit perform negative feedback control on the main power tube to achieve a stable output, and further include a soft start circuit with a constant current source connected to the output end of the input filter circuit and Between the drive control circuits; when the self-oscillating flyback converter is in an unsteady state after power-on, the soft start circuit is charged to the soft start circuit with a constant current source through a constant current source, and the soft start voltage rises to the MOS tube threshold After the value is turned on, the MOS tube is turned on. Since the constant current source charging speed is balanced with the soft start capacitor discharging speed, the soft starting capacitor continuously supplies driving compensation to the MOS tube to achieve normal starting; the self-oscillating flyback converter enters the steady state. The constant current source charging speed is balanced with the soft start capacitor discharging speed, and the soft start continues to provide driving compensation to the MOS tube. Steady work properly.
作为本发明的一种实施方式,所述带恒流源的软启动电路包括恒流源、第一 分压电阻、第二分压电阻和启动电容; 所述恒流源的阳极连接到自激振荡反激变 换器的电源输入端,所述恒流源的阴极依次通过所述第一分压电阻和第二分压电 阻连接到自激振荡反激变换器的电源参考端,所述启动电容与所述第一分压电阻 和第二分压电阻的串联支路相并联,所述第一分压电阻和第二分压电阻的连接点 与所述主功率管的栅极相连接。 As an embodiment of the present invention, the soft start circuit with a constant current source includes a constant current source, a first voltage dividing resistor, a second voltage dividing resistor, and a starting capacitor; the anode of the constant current source is connected to the self-excited a power supply input end of the oscillating flyback converter, wherein the cathode of the constant current source is sequentially connected to a power reference terminal of the self-oscillating flyback converter through the first voltage dividing resistor and the second voltage dividing resistor, the starting capacitor And the first voltage dividing resistor And a series branch of the second voltage dividing resistor is connected in parallel, and a connection point of the first voltage dividing resistor and the second voltage dividing resistor is connected to a gate of the main power tube.
更优的,所述带恒流源的软启动电路还包括限流电阻, 所述恒流源的阳极通 过所述限流电阻连接到自激振荡反激变换器的电源输入端。  More preferably, the soft start circuit with a constant current source further includes a current limiting resistor, and the anode of the constant current source is connected to the power input end of the self-oscillating flyback converter through the current limiting resistor.
更优的,所述带恒流源的软启动电路还包括限流电阻, 所述限流电阻连接在 所述恒流源的阴极与所述第一分压电阻和启动电容的连接点之间。  More preferably, the soft start circuit with a constant current source further includes a current limiting resistor connected between a cathode of the constant current source and a connection point between the first voltage dividing resistor and the starting capacitor .
作为本发明的一种实施方式, 所述的恒流源(D1A)为单一恒流源, 或恒流 源并联形式, 或恒流源与三极管、 稳压管、 电阻复合而成的恒流源, 或三极管和 电阻构成的恒流源。  As an embodiment of the present invention, the constant current source (D1A) is a single constant current source, or a constant current source in parallel, or a constant current source formed by a combination of a constant current source and a triode, a Zener, and a resistor. , or a constant current source composed of a triode and a resistor.
与现有技术相比, 本发明具有以下优点:  Compared with the prior art, the present invention has the following advantages:
本发明的优点之一: 提高变换器产品启动能力和带容性负载能力, 使产品在 满负载和带容性负载时能正常启动并稳定工作。 表五列举了型号为 WRF4815P、 功率为 6W 的电源变换器的原方案与采用本发明技术方案后的带容性负载能力 对比。  One of the advantages of the present invention is to improve the starting capability of the inverter product and the capacitive load capacity, so that the product can start normally and work stably under full load and capacitive load. Table 5 lists the original scheme of the power converter of the type WRF4815P and the power of 6W compared with the capacitive load capacity after adopting the technical scheme of the present invention.
表五 Table 5
Figure imgf000010_0001
Figure imgf000010_0001
从表五中可以看出,采用本发明的技术方案后, 变换器产品带容性负载能力 大大提高。  It can be seen from Table 5 that after adopting the technical solution of the present invention, the capacity load capacity of the converter product is greatly improved.
本发明的优点之二: 提高产品轻负载 (10%负载以下) 时输出电压稳定性, 降低反馈绕组在轻负载时能量偏低(轻负载时电流较小, 耦合能量弱)带来的驱 动不足隐患; 同时使得自振荡反激变换器设计更大额定负载电流成为可能。  The second advantage of the invention is: improving the output voltage stability when the product is lightly loaded (below 10% load), reducing the low energy of the feedback winding under light load (small current at light load, weak coupling energy) Hidden dangers; at the same time, it makes it possible to design a larger rated load current for the self-oscillating flyback converter.
本发明优点之三: 降低产品对变压器的工艺要求(反馈与输入耦合系数要求 降低), 降低对开关管门限一致性的要求, 提高了产品的可生产性。 图 10列举了 型号为 IRFR220的 MOS管不同批次之间的门限测试。 测试条件: Vgs=Vds和 Id=250 μ A, T=25°C, 从图中可以看出, MOS 管门限开启电压, 批次之间差异 较大, YG批次 MOS管门限高于 OM批次 MOS管门限, 因此其对反馈绕组能 量的需求也增加, 本发明技术, 为 MOS管提供了合适的补偿, 因此能可靠驱动 MOS管, 降低了对 MOS管导通门限的一致性要求。 The third advantage of the invention is that the process requirements of the transformer are reduced (the feedback and input coupling coefficient requirements are reduced), the requirement for the consistency of the switch tube threshold is reduced, and the product productivity is improved. Figure 10 shows the threshold test between different batches of MOS tubes of the type IRFR220. Test conditions: Vgs=Vds and Id=250 μ A, T=25°C. It can be seen from the figure that the MOS tube threshold turn-on voltage has a large difference between batches, and the YG batch MOS tube threshold is higher than the OM batch. Secondary MOS tube threshold, so it can The demand for the quantity also increases. The technique of the present invention provides suitable compensation for the MOS tube, so that the MOS tube can be reliably driven, and the consistency requirement for the MOS tube conduction threshold is reduced.
本发明的优点之四: 提高产品抗干扰能力, 有效解决了输出掉电现象。 实验 中采用型号为 PWB4805D、功率为 3W的电源变换器产品为单片机供电,输入端 打 4KV静电时, 单片机复位, 该电源变换器输出掉电; 采用本发明的技术方案 改进后, 输入端打 4KV静电, 变换器产品输出正常, 单片机工作正常。  The fourth advantage of the invention is: improving the anti-interference ability of the product, and effectively solving the output power-down phenomenon. In the experiment, the power converter product with the model PWB4805D and power of 3W is used to supply power to the single-chip microcomputer. When the input terminal is 4KV static, the single-chip microcomputer is reset, and the power converter output is powered down. After the technical solution of the present invention is improved, the input terminal is 4KV. Static electricity, the output of the inverter is normal, and the MCU works normally.
本发明的优点之五: 提高了自振荡反激变换器在宽输入电压范围 (4:1 ) 中适 应性。 由于恒流源的启动, 产品在低压、 标称、 高压时启动速度、 补偿强度得到 有效控制, 使得 MOS管驱动在全输入电压和全负载范围以及大容性负载条件下 得到合理的补偿,有效解决了短路功耗、启动电流等问题。表六列举了采用 0.5mA 的恒流源后, PWB4805D、 功率为 3W的电源变换器的短路功耗、 启动性能。 表六  Advantage 5 of the present invention: The adaptability of the self-oscillating flyback converter over a wide input voltage range (4:1) is improved. Due to the start of the constant current source, the starting speed and compensation intensity of the product under low voltage, nominal and high voltage are effectively controlled, so that the MOS tube drive is reasonably compensated under the conditions of full input voltage and full load range and large capacitive load. Solved short-circuit power consumption, starting current and other issues. Table 6 lists the short-circuit power consumption and startup performance of a PWB4805D, 3W power converter with a constant current source of 0.5mA. Table 6
Figure imgf000011_0001
Figure imgf000011_0001
对比表三, 可以看出其短路功耗大大下降, 产品具备长时间短路保护功能; 各输入电压点的启动电流较表三均减小, 标称时减小 67mA, 减小幅度 37% , 降低客户开机启动电源限流风险。 附图说明  Comparing Table 3, it can be seen that the short-circuit power consumption is greatly reduced, and the product has a long-term short-circuit protection function; the starting current of each input voltage point is reduced compared with Table 3, and the nominal reduction is 67 mA, the reduction is 37%, and the reduction is reduced. The customer starts the power supply current limit risk. DRAWINGS
下面结合附图和具体实施例对本发明作进一步的详细说明:  The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments:
图 1为现有技术中自激振荡反激变换器的原理框图;  1 is a schematic block diagram of a prior art self-oscillating flyback converter;
图 2为现有技术中自激振荡反激变换器的电路原理图;  2 is a circuit schematic diagram of a self-oscillating flyback converter in the prior art;
图 3为本发明自激振荡反激变换器的原理框图;  3 is a schematic block diagram of a self-oscillating flyback converter of the present invention;
图 4为本发明实施例一的电路原理图;  4 is a schematic circuit diagram of a first embodiment of the present invention;
图 5为本发明实施例二种带恒流源软启动电路部分的电路原理图; 图 6为本发明实施例三种带恒流源软启动电路部分的电路原理图; 图 7为自激振荡反激变换器的软启动电路中启动电容 C9的电压波形图; 图 8为型号为 PWB4805D、 功率为 3W的电源变换器低压、 满载时的 Vgs 驱动波形图; 5 is a schematic circuit diagram of a portion of a soft start circuit with a constant current source according to an embodiment of the present invention; FIG. 6 is a circuit schematic diagram of three soft start circuit portions with a constant current source according to an embodiment of the present invention; FIG. 7 is a self-oscillation A voltage waveform diagram of the starting capacitor C9 in the soft start circuit of the flyback converter; Figure 8 is a waveform diagram of Vgs driving at low voltage and full load of a power converter of the type PWB4805D and power of 3W;
图 9为型号为 PWB4805D、 功率为 3W的电源变换器轻载、 高压时的 Vgs 驱动波形图;  Figure 9 is a waveform diagram of Vgs driving at a light load and high voltage of a power converter of model PWB4805D and power of 3W;
图 10为型号为 IRFR220的 MOS管的常温门限测试曲线图;  Figure 10 is a graph showing the normal temperature threshold of a MOS tube of the type IRFR220;
图 11为型号 PWB4805D、功率为 3W的电源变换器输入打 4KV静电时, 输 出电压波形图;  Fig. 11 is a waveform diagram of the output voltage when the power converter of the model PWB4805D and the power of 3W is input with 4KV static electricity;
图 12为型号 PWB4805D、功率为 3W的电源变换器不带容性负载和带 1000 μ f容性负载时输出电压上升波形图;  Figure 12 shows the waveform of the output voltage rise of the PWB4805D, 3W power converter without capacitive load and 1000 μf capacitive load;
图 13为恒流源并联形式的电路原理图;  Figure 13 is a circuit schematic diagram of a parallel form of a constant current source;
图 14为恒流源、 电阻、 稳压管、 三极管构成的复合恒流源的电路原理图; 图 15为 PNP三极管、 电阻构成的恒流源的电路原理图;  14 is a circuit schematic diagram of a composite constant current source composed of a constant current source, a resistor, a Zener tube, and a triode; FIG. 15 is a circuit schematic diagram of a constant current source formed by a PNP transistor and a resistor;
图 16为单管驱动控制电路的电路原理图。 具体实施方式  Figure 16 is a circuit schematic of a single tube drive control circuit. detailed description
下面结合附图和具体实施例对本发明作进一步的详细说明。  The invention will be further described in detail below with reference to the drawings and specific embodiments.
如图 3所示,本发明的自激振荡反激变换器, 直流输入信号依次经过输入滤 波电路、主功率电路和输出滤波电路后输出直流信号, 主功率电路包括主功率管 和主变压器; 所述输出的直流信号依次通过稳压电路、隔离光耦和驱动控制电路 对主功率管进行负反馈控制以实现稳定输出,还包括带恒流源的软启动电路, 该 软启动电路连接在所述输入滤波电路的输出端和所述驱动控制电路之间;所述自 激振荡反激变换器上电后处于非稳态时,通过恒流源向带恒流源的软启动电路充 电, 软启动电压上升到 MOS管门限值后 MOS管导通, 由于恒流源充电速度与 软启动电容放电速度平衡, 软启动电容持续不断向 MOS管提供驱动补偿, 实现 启动正常; 所述自激振荡反激变换器进入稳态以后, 恒流源充电速度与软启动电 容放电速度维持平衡,软启动持续向 MOS管提供驱动补偿,实现稳态正常工作。  As shown in FIG. 3, in the self-oscillating flyback converter of the present invention, the DC input signal sequentially passes through the input filter circuit, the main power circuit and the output filter circuit, and then outputs a DC signal, and the main power circuit includes a main power tube and a main transformer; The output DC signal is sequentially subjected to negative feedback control of the main power tube through the voltage stabilization circuit, the isolation optocoupler and the drive control circuit to achieve stable output, and further includes a soft start circuit with a constant current source, the soft start circuit being connected Between the output end of the input filter circuit and the drive control circuit; the self-oscillating flyback converter is charged to a soft start circuit with a constant current source through a constant current source when it is in an unsteady state, soft start After the voltage rises to the MOS tube threshold, the MOS transistor is turned on. Since the constant current source charging speed is balanced with the soft start capacitor discharge speed, the soft start capacitor continuously supplies driving compensation to the MOS tube to achieve normal startup; After the converter enters the steady state, the constant current source charging speed and the soft start capacitor discharge speed are balanced, and the soft start continues to the MOS tube. For drive compensation, achieve steady state normal operation.
参见图 4, 为本发明带恒流源软启动电路的自振荡反激变换器第一实施例。 本实施例一中, 采用脉冲频率调制电路 13作为变换器的驱动控制电路, 本实施 例的实现电路主要包括以下几个组成部分: 输入滤波电路 11、 带恒流源软启动 电路 12、 脉冲频率调制电路 13、 主功率电路 14、 辅助电源 15、 输出滤波电路 16、稳压电路 17,其中输入滤波电路 11、脉冲频率调制电路 13、主功率电路 14、 辅助电源 15、输出滤波电路 16、稳压电路 17的电路结构与背景技术中图 2所示 电路的电路结构相同,本实施例与图 2所示电路的区别在于本实施例与图 2所示 电路的带恒流源软启动电路 12的电路组成结构不同: Referring to FIG. 4 , a first embodiment of a self-oscillating flyback converter with a constant current source soft start circuit of the present invention is shown. In the first embodiment, the pulse frequency modulation circuit 13 is used as the drive control circuit of the converter. The implementation circuit of the embodiment mainly includes the following components: an input filter circuit 11, a soft start circuit with a constant current source, and a pulse frequency. Modulation circuit 13, main power circuit 14, auxiliary power supply 15, output filter circuit 16. The voltage stabilizing circuit 17, wherein the circuit structure of the input filter circuit 11, the pulse frequency modulation circuit 13, the main power circuit 14, the auxiliary power supply 15, the output filter circuit 16, and the voltage stabilization circuit 17 is the same as that of the circuit shown in FIG. The circuit structure is the same. The difference between this embodiment and the circuit shown in FIG. 2 is that the circuit composition of the constant current source soft start circuit 12 of the circuit shown in FIG. 2 is different from that of the circuit shown in FIG. 2:
本实施例中, 主功率电路 14中的主功率管采用 MOS管 TR1 , 主变压器采 用变压器 Tl, 变压器 T1包含有原边绕组 PI、 输出绕组 P2和正反馈绕组 P3。  In this embodiment, the main power tube in the main power circuit 14 uses the MOS tube TR1, and the main transformer uses the transformer T1. The transformer T1 includes the primary winding PI, the output winding P2, and the positive feedback winding P3.
输入滤波电路 11, 包括滤波电容 C0、 滤波电容 C1和滤波电感 L0, 其结构 为公知的 IT型滤波电路原理结构, 在此不详细说明。  The input filter circuit 11 includes a filter capacitor C0, a filter capacitor C1 and a filter inductor L0, and the structure thereof is a well-known IT type filter circuit principle structure, which will not be described in detail herein.
带恒流源软启动电路 12, 包括恒流源 D1A、 第一分压电阻 R13、 第二分压 电阻 R14, 启动电容 C9。 恒流源 D1A的阳极接电源输入端, 恒流源 D1A的阴 极依次通过所述第一分压电阻 R13和第二分压电阻 R14连接到自激振荡反激变 换器的电源参考端,所述启动电容 C9与所述第一分压电阻 R13和第二分压电阻 R14的串联支路相并联, 所述第一分压电阻 R13和第二分压电阻 R14的连接点 与所述主功率管的栅极相连接。 下面详细介绍该电路的工作原理:  The soft start circuit with constant current source 12 includes a constant current source D1A, a first voltage dividing resistor R13, a second voltage dividing resistor R14, and a starting capacitor C9. The anode of the constant current source D1A is connected to the power input end, and the cathode of the constant current source D1A is sequentially connected to the power reference end of the self-oscillating flyback converter through the first voltage dividing resistor R13 and the second voltage dividing resistor R14. The starting capacitor C9 is connected in parallel with the series branch of the first voltage dividing resistor R13 and the second voltage dividing resistor R14, and the connection point of the first voltage dividing resistor R13 and the second voltage dividing resistor R14 is opposite to the main power tube The gates are connected. The working principle of this circuit is described in detail below:
启动阶段: 当接入输入电压时, 电压经恒流源向启动电容 C9充电, 经过时 间 ί = ^ (其中 υώ为 MOS管 TR1的启动门限, C为启动电容 C9的容量, i为恒流 c * i Start-up phase: When the input voltage is connected, the voltage is charged to the starting capacitor C9 via the constant current source, and the elapsed time ί = ^ (where υ ώ is the starting threshold of the MOS transistor TR1, C is the capacity of the starting capacitor C9, and i is the constant current c * i
源 D1A工作区的恒定电流) 后启动电容 C9电压达到 MOS管 TR1的门限电压, 实 现开机软启动功能, 在输出电压正式建立之前, 即非稳态时期, 一方面启动电容 C9通过第一分压电阻 R13向 MOS管 TR1提供能量, 另一方面恒流源 D1A及时为 启动电容 C9提供能量, 选用合适的恒流源 D1A即可以满足启动电容 C9充放电的 能量平衡, 由于启动电容 C9的能量补偿, 使得正反馈绕组 P3只需要极少的能量 就能完成自激振荡过程, 从而避免了间歇式振荡的发生, 使产品启动正常。 After the constant current of the source D1A working area), the voltage of the starting capacitor C9 reaches the threshold voltage of the MOS transistor TR1, and the soft start function is realized. Before the output voltage is officially established, that is, during the non-steady state, the starting capacitor C9 passes the first partial voltage. The resistor R13 supplies energy to the MOS transistor TR1. On the other hand, the constant current source D1A supplies energy to the starting capacitor C9 in time, and the appropriate constant current source D1A can be selected to meet the energy balance of the starting capacitor C9 charging and discharging, due to the energy compensation of the starting capacitor C9. The positive feedback winding P3 requires only a small amount of energy to complete the self-oscillation process, thereby avoiding the occurrence of intermittent oscillations and starting the product normally.
正常工作阶段: 当输入电压变化或者输出负载变化时, 正反馈绕组 P3耦合 能量发生变化, 由于启动电容 C9通过第一分压电阻 R13为 MOS管 TR1的驱动 提供的能量与恒流源 D1A为启动电容 C9提供的能量持平, MOS管 TR1驱动得 到有效补偿,对正反馈绕组 P3耦合能量要求大大降低,确保正反馈绕组 P3自激 振荡正常,提高了产品在不同输入电压和不同负载条件下输出电压的稳定性。表 七为型号为 PW4805D、功率为 3W的电源变换器,恒流源 D1A采用 0.5mA的恒 流源后, 启动电容 C9在不同输入电压和负载条件下的电压值, 对比表二、表四, 采用本发明技术后, 启动电容 C9上的电压基本稳定, 变化范围仅 2.4V, 避免了 原方案中电阻 R10取值不同造成的 MOS管驱动过补偿和欠补偿。 Normal working phase: When the input voltage changes or the output load changes, the coupling energy of the positive feedback winding P3 changes, because the starting capacitor C9 is powered by the first voltage dividing resistor R13 for the driving of the MOS transistor TR1 and the constant current source D1A is activated. Capacitor C9 provides the same level of energy, MOS tube TR1 drive is effectively compensated, the coupling energy requirement for positive feedback winding P3 is greatly reduced, ensuring that the positive feedback winding P3 self-oscillation is normal, improving the output voltage of the product under different input voltages and different load conditions. Stability. Table 7 shows the power converter of model PW4805D with power of 3W. After constant current source D1A adopts constant current source of 0.5mA, the voltage of starting capacitor C9 under different input voltage and load conditions is compared with Table 2 and Table 4. After adopting the technology of the present invention, the voltage on the starting capacitor C9 is basically stable, and the variation range is only 2.4V, which avoids the MOS tube driving over-compensation and under-compensation caused by the difference in the value of the resistor R10 in the original scheme.
表七 Table 7
Figure imgf000014_0001
Figure imgf000014_0001
脉冲频率调制电路 13包括: 电阻 R6、 电阻 R9、 电阻 Rll、 电阻 R111,电容 C5、 电容 C6、 电容 C12, NPN型三极管 TR2、 PNP型三极管 TR3。 电容 C6与 电阻 Rll、 电阻 Rlll、 电阻 R6依次串联, 串联电路的一端接入正反馈绕组 P3 的同名端, 另一端接入三极管 TR2的集电极。 电容 C5与电阻 R9并联, 并联电 流的一端接入三极管 TR2的基极, 另一端接入三极管 TR3的集电极。 电容 C12 与电阻 R111 并联, 并联电路的一端接入三极 TR3 的发射极, 另一端接入三极 TR3的基极。 其工作原理为: 正反馈绕组 P3、 电容 C6、 电阻 R11支路通过与主 功率变压器 T1原副边耦合, 构成自激振荡回路, 控制 MOS管 TR1导通关断; 同时电流环电阻 R5支路、电压环光耦 OC1支路通过由三极管 TR2、三极管 TR3 双管驱动控制电路调节开关管导通占空比, 使产品输出正常。  The pulse frequency modulation circuit 13 includes: a resistor R6, a resistor R9, a resistor R11, a resistor R111, a capacitor C5, a capacitor C6, a capacitor C12, a NPN transistor TR2, and a PNP transistor TR3. The capacitor C6 is connected in series with the resistor R11, the resistor R111, and the resistor R6. One end of the series circuit is connected to the same name end of the positive feedback winding P3, and the other end is connected to the collector of the transistor TR2. The capacitor C5 is connected in parallel with the resistor R9. One end of the parallel current is connected to the base of the transistor TR2, and the other end is connected to the collector of the transistor TR3. Capacitor C12 is connected in parallel with resistor R111. One end of the parallel circuit is connected to the emitter of the three-pole TR3, and the other end is connected to the base of the three-pole TR3. The working principle is as follows: The positive feedback winding P3, the capacitor C6, and the resistor R11 branch are coupled with the primary and secondary sides of the main power transformer T1 to form a self-excited oscillation circuit, and the control MOS transistor TR1 is turned on and off; and the current loop resistor R5 branch The voltage loop optocoupler OC1 branch adjusts the on-duty of the switch tube by the double-tube drive control circuit of the transistor TR2 and the transistor TR3, so that the product output is normal.
主功率电路 14,包括变压器 T1的原边绕组 Pl、输出绕组 P2, MOS管 TR1、 限流电阻 R5, 吸收电容 C14, 输出整流二极管 Dl, 实现电源能量的转换、 传递 以及输入与输出隔离。  The main power circuit 14, including the primary winding P1 of the transformer T1, the output winding P2, the MOS transistor TR1, the current limiting resistor R5, the absorbing capacitor C14, and the output rectifier diode Dl, realize the conversion, transmission and isolation of the input and output of the power source.
辅助电源 15, 包括正反馈绕组 P3、二极管 D3, 其工作原理为反馈绕组同名 端为正时二极管 D3导通, 为光耦提供能量。  Auxiliary power supply 15, including positive feedback winding P3, diode D3, works on the principle that the feedback winding has the same name as the timing diode D3 is conducting, providing energy for the optocoupler.
输出滤波电路 16, 包括电容 C3, 也可采用其它现有滤波电路, 具体可按有 关技术手册选定。  The output filter circuit 16, including the capacitor C3, can also be used with other existing filter circuits, which can be selected according to the relevant technical manual.
稳压电路 17, 主要包括稳压器 ADJ, 其通过光耦 OC1连接到主功率电路及 辅助电源 15及脉冲频率调制电路 13, 在此不再赘述。  The voltage stabilizing circuit 17 mainly includes a voltage regulator ADJ, which is connected to the main power circuit, the auxiliary power source 15 and the pulse frequency modulation circuit 13 through the optocoupler OC1, and will not be described herein.
以下对另外两种实施例简要进行说明, 其中仅示出带恒流源软启动电路部 分, 其他部分的接法与图 4所示电路相同。 图 5为本发明实施例二的带恒流源软启动电路部分,其与实施例一的区别在 于本实施例的带恒流源软启动电路中恒流源 D1A与电源输入端之间串联了用于 限制最大电流的限流电阻 R1A, 该电阻可以限制启动电路最大电流, 同时降低 恒流源两端的分压。 The other two embodiments will be briefly described below, wherein only the portion of the soft start circuit with the constant current source is shown, and the connections of the other portions are the same as those of the circuit shown in FIG. 5 is a portion of a soft start circuit with a constant current source according to a second embodiment of the present invention. The difference from the first embodiment is that the constant current source D1A and the power input end are connected in series with the constant current source soft start circuit of the embodiment. A current limiting resistor R1A for limiting the maximum current, which limits the maximum current of the startup circuit while reducing the voltage division across the constant current source.
图 6为本发明实施例三的带恒流源软启动电路部分,其与实施例二所示电路 的区别仅在于本实施例的带恒流源软启动电路中的用于限制最大电流的限流电 阻 R1A的接入位置发生变化, 限流电阻 R1A连接在所述恒流源 D1A的阴极与 第一分压电阻 R13和启动电容 C9的连接点之间  6 is a portion of a soft start circuit with a constant current source according to a third embodiment of the present invention, which differs from the circuit shown in the second embodiment only in limiting the maximum current in the constant current source soft start circuit of the embodiment. The access position of the flow resistor R1A changes, and the current limiting resistor R1A is connected between the cathode of the constant current source D1A and the connection point of the first voltage dividing resistor R13 and the starting capacitor C9.
需要说明的是上述恒流源可以有多种组成方式,其可以是现有技术中的恒流 源并联形式, 如图 13; 现有技术中的恒流源与三极管、 稳压管、 电阻复合而成 的恒流源, 如图 14其恒流输出为 Io= (V2D— VBE1) / R; 亦可以是现有技术中的三 极管和电阻构成的恒流源, 如图 15, 其恒流输出 Io = Vte/R55 等等。 It should be noted that the above constant current source may have multiple compositions, which may be a parallel form of a constant current source in the prior art, as shown in FIG. 13; a constant current source and a triode, a Zener, and a resistor in the prior art The constant current source, as shown in Fig. 14, has a constant current output of Io = (V 2D - V BE1 ) / R; it can also be a constant current source composed of a triode and a resistor in the prior art, as shown in Fig. 15, which is constant The stream output Io = V te /R 5 . 5 and so on.
除上述说明的几种实施电路外,本行业技术人员通过以上描述与附图举例能 自然联想到的其它等同应用方案, 例如变换器的驱动控制电路采用如图 16中, 由三极管 TR2、 电阻 R6、 电容 C5和电阻 R9组成的单管驱动控制电路取代上述 实施例中的脉冲频率调制电路等,对于本技术领域的普通技术人员来说, 在不脱 离本发明原理的前提下,对本发明进行若干的改进和修饰落入本发明权利要求的 保护范围内。  In addition to the several implementation circuits described above, other equivalent applications that can be naturally associated with the above description and the accompanying drawings by the skilled person, for example, the drive control circuit of the converter is as shown in FIG. 16, by the transistor TR2, the resistor R6. The single-tube drive control circuit composed of the capacitor C5 and the resistor R9 replaces the pulse frequency modulation circuit and the like in the above embodiments. For those skilled in the art, the present invention may be carried out without departing from the principle of the present invention. Modifications and modifications are intended to fall within the scope of the appended claims.

Claims

权利要求 Rights request
1.一种自激振荡反激变换器, 直流输入信号依次经过输入滤波电路、 主功率电 路和输出滤波电路后输出直流信号, 主功率电路包括主功率管和主变压器; 所述输 出的直流信号依次通过稳压电路、 隔离光耦和驱动控制电路对主功率管进行负反馈 控制以实现稳定输出, 其特征在于: 还包括带恒流源的软启动电路, 该软启动电路 连接在所述输入滤波电路的输出端和所述驱动控制电路之间。 1. A self-oscillating flyback converter, wherein a DC input signal sequentially passes through an input filter circuit, a main power circuit, and an output filter circuit, and outputs a DC signal, the main power circuit includes a main power tube and a main transformer; and the output DC signal The main power tube is negatively feedback controlled by the voltage stabilizing circuit, the isolated optocoupler and the driving control circuit to realize a stable output, and is characterized in that: a soft start circuit with a constant current source is further connected, and the soft start circuit is connected to the input Between the output of the filter circuit and the drive control circuit.
2.根据权利要求 1所述的自激振荡反激变换器, 其特征在于: 所述带恒流源的 软启动电路包括恒流源 (D1A) 、 第一分压电阻 (R13 ) 、 第二分压电阻 (R14) 和启动电容 (C9 ) ; 所述恒流源 (D1A) 的阳极连接到自激振荡反激变换器的电 源输入端, 所述恒流源 (D1A) 的阴极依次通过所述第一分压电阻 (R13 ) 和第二 分压电阻 (R14) 连接到自激振荡反激变换器的电源参考端, 所述启动电容 (C9) 与所述第一分压电阻 (R13 ) 和第二分压电阻 (R14) 的串联支路相并联, 所述第 一分压电阻 (R13 ) 和第二分压电阻 (R14) 的连接点与所述主功率管的栅极相连 接。  The self-oscillating flyback converter according to claim 1, wherein: the soft start circuit with a constant current source comprises a constant current source (D1A), a first voltage dividing resistor (R13), and a second a voltage dividing resistor (R14) and a starting capacitor (C9); an anode of the constant current source (D1A) is connected to a power input terminal of the self-oscillating flyback converter, and a cathode of the constant current source (D1A) sequentially passes through The first voltage dividing resistor (R13) and the second voltage dividing resistor (R14) are connected to a power reference terminal of the self-oscillating flyback converter, and the starting capacitor (C9) and the first voltage dividing resistor (R13) Connected in parallel with the series branch of the second voltage dividing resistor (R14), the connection point of the first voltage dividing resistor (R13) and the second voltage dividing resistor (R14) is connected to the gate of the main power tube.
3.根据权利要求 2所述的自激振荡反激变换器, 其特征在于: 所述带恒流源的 软启动电路还包括限流电阻 (R1A) , 所述恒流源 (D1A) 的阳极通过所述限流电 阻 (R1A) 连接到自激振荡反激变换器的电源输入端。  The self-oscillating flyback converter according to claim 2, wherein: the soft start circuit with a constant current source further comprises a current limiting resistor (R1A), and an anode of the constant current source (D1A) The current limiting input (R1A) is connected to the power input terminal of the self-oscillating flyback converter.
4.根据权利要求 2所述的自激振荡反激变换器, 其特征在于: 所述带恒流源的 软启动电路还包括限流电阻 (R1A) , 所述限流电阻 (R1A) 连接在所述恒流源 The self-oscillating flyback converter according to claim 2, wherein: the soft start circuit with a constant current source further comprises a current limiting resistor (R1A), and the current limiting resistor (R1A) is connected to Constant current source
(D1A) 的阴极与所述第一分压电阻 (R13) 和启动电容 (C9) 的连接点之间。 The cathode of (D1A) is connected to the junction of the first voltage dividing resistor (R13) and the starting capacitor (C9).
5.根据权利要求 1至 4任一项所述的自激振荡反激变换器, 其特征在于: 所述 的恒流源 (D1A) 为单一恒流源, 或恒流源并联形式, 或恒流源与三极管、 稳压 管、 电阻复合而成的恒流源, 或三极管和电阻构成的恒流源。  The self-oscillating flyback converter according to any one of claims 1 to 4, characterized in that: the constant current source (D1A) is a single constant current source, or a constant current source is connected in parallel, or constant A constant current source composed of a combination of a current source and a triode, a Zener tube, and a resistor, or a constant current source composed of a triode and a resistor.
PCT/CN2012/074151 2012-02-07 2012-04-16 Self-oscillatory flyback converter WO2013117049A1 (en)

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Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109412136A (en) * 2018-11-21 2019-03-01 扬州大学 A kind of mesolow DC power-supply system stabilising arrangement and its control method
US10338620B2 (en) 2017-11-15 2019-07-02 Infineon Technologies Ag Feedback circuit for regulation loops

Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN104578843B (en) * 2014-12-22 2017-04-19 广州金升阳科技有限公司 Filter circuit of AC/DC (alternating current/direct current) switching converter
CN106026712B (en) * 2016-07-15 2018-05-29 深圳南云微电子有限公司 The secondary side control methods and circuit of Switching Power Supply
CN109039093B (en) * 2018-09-29 2024-01-23 杰华特微电子股份有限公司 Isolation type switching power supply and control method thereof
CN109245513B (en) * 2018-11-09 2024-04-09 深圳南云微电子有限公司 Starting circuit

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH10108462A (en) * 1996-09-27 1998-04-24 Toko Inc Self-excited switching power supply circuit
CN1499704A (en) * 2002-11-01 2004-05-26 ������������ʽ���� Switching mains
CN101272098A (en) * 2008-04-08 2008-09-24 广州金升阳科技有限公司 Twin-triode current control type self-oscillation flyback converter
US20100157629A1 (en) * 2008-12-18 2010-06-24 Panasonic Corporation Semiconductor laser apparatus
CN101997423A (en) * 2010-10-29 2011-03-30 广州金升阳科技有限公司 Self-oscillating flyback converter with function of output short circuit protection

Family Cites Families (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7719249B2 (en) * 2005-01-28 2010-05-18 Semiconductor Components Industries, Llc Soft-start circuit and method therefor
KR101677729B1 (en) * 2010-02-19 2016-11-18 페어차일드코리아반도체 주식회사 Switch control device, power supply device comprising the same, and switch control method

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH10108462A (en) * 1996-09-27 1998-04-24 Toko Inc Self-excited switching power supply circuit
CN1499704A (en) * 2002-11-01 2004-05-26 ������������ʽ���� Switching mains
CN101272098A (en) * 2008-04-08 2008-09-24 广州金升阳科技有限公司 Twin-triode current control type self-oscillation flyback converter
US20100157629A1 (en) * 2008-12-18 2010-06-24 Panasonic Corporation Semiconductor laser apparatus
CN101997423A (en) * 2010-10-29 2011-03-30 广州金升阳科技有限公司 Self-oscillating flyback converter with function of output short circuit protection

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US10338620B2 (en) 2017-11-15 2019-07-02 Infineon Technologies Ag Feedback circuit for regulation loops
CN109412136A (en) * 2018-11-21 2019-03-01 扬州大学 A kind of mesolow DC power-supply system stabilising arrangement and its control method

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