JP5862690B2 - Control device for motor drive device and motor drive system - Google Patents

Description

  The present invention relates to a control device for an electric motor drive device and an electric motor drive system, for example, an electric motor drive device that applies an AC voltage to a synchronous motor.

  Non-Patent Documents 1 and 2 describe a method for controlling a synchronous motor. The synchronous motor has an armature having windings and a field. In Patent Document 1, the primary magnetic flux of the synchronous motor is controlled. More specifically, the γ-axis and δ-axis that are orthogonal to each other are adopted as the control axes, and the γ-axis component of the primary magnetic flux is controlled to zero and the δ-axis component is controlled to the primary magnetic flux command value.

  In Non-Patent Documents 1 and 2, in order to improve control stability, correction is performed with a correction amount based on the γ-axis current, and the rotational speed of the δ-γ rotational coordinate system is calculated. As the correction amount, in Non-Patent Document 1, the product of the γ-axis current and the gain is adopted, and in Non-Patent Document 2, the harmonic component obtained by removing the DC component from the γ-axis current and the gain are used. Product is adopted.

Kaku, Yamamura, Tsunehiro, “Position Sensorless Control Method for DC Brushless Motor”, IEEJ Transactions D, 1991, Vol. 111, No. 8, p.639-644 Hamada, Yamamura, Tsunehiro, "General-purpose inverter for driving synchronous machines", IEEJ Transactions D, 1999, Vol.119, No.5, p.707-712

  However, Non-Patent Documents 1 and 2 neither describe nor suggest how to determine the gain.

  The applicant of the present application has found a relationship between controllability and gain for controlling the rotational speed of the electric motor.

  Accordingly, an object of the present invention is to provide a control technique for an electric motor drive device that can improve the controllability of the rotational speed control.

The first aspect of the control device for the motor drive device according to the present invention applies an AC voltage to the synchronous motor (2) having the field (23) and the armature (21), and generates an AC current (iu, iv). , iw), a control device (3) for controlling the drive device (1), which is a rotary coordinate system having a δc axis and a γc axis that advances 90 degrees in a predetermined advance direction with respect to the δc axis, Based on the rotation angle (θ) with respect to the fixed coordinate system, the AC current is converted into a δc-axis current (iδc) and a γc-axis current (iγc) in the rotary coordinate system, and a compensation gain The rotational speed (ω1) is corrected by correcting the command value (ω1 ^* ) of the rotational speed (ω1) of the rotational coordinate system with a correction amount obtained by the product of (Km) and the compensation value depending on the γc-axis current. A rotation speed calculation unit for calculating the phase, a phase calculation unit (33) for calculating the rotation angle based on the rotation speed, and a flux linkage (Λ0) to the armature by the field A voltage command value in the rotating coordinate system for the AC voltage such that a primary magnetic flux (λ), which is a combination of the armature reaction magnetic flux generated by the AC current, is along the δc axis in the rotating coordinate system. a voltage command calculation unit (35) for generating ([v _δγ ^* ]), and a control signal for generating a control signal for controlling the application of the AC voltage by the driving device based on the voltage command value and the rotation angle A control device for an electric motor drive device comprising a generator (36), wherein the compensation gain takes a first gain value when the rotational speed is smaller than a reference value, and the rotational speed is less than the reference value. When it is larger, a second gain value larger than the first gain value is taken.

  A second aspect of the motor drive device control device according to the present invention is the motor drive device control device according to the first aspect, wherein the compensation value is extracted by a time constant (Tm). The time constant takes a first time constant value when the rotational speed is smaller than a reference value, and the time constant takes the first time constant value when the rotational speed is larger than the reference value. A second time constant value larger than the first time constant value is taken.

  A first aspect of an electric motor drive system according to the present invention includes a control device for an electric motor drive device according to the first or second aspect, and the drive device.

  According to the first aspect of the motor drive control device and the motor drive system according to the present invention, the rotational speed is obtained by performing correction with the compensation gain and the correction amount, and based on this, the primary magnetic flux is δc. The drive device is controlled along the axis. Thereby, the phase delay of the frequency characteristic of the transfer function of the output torque with respect to the load torque of the synchronous motor increases on the low frequency side as the compensation gain increases. Further, by this control, the peak value of the gain (transfer gain) of the frequency characteristic of the transfer function is reduced as the compensation gain is increased.

  According to the first aspect, the compensation gain takes the first gain value when the rotational speed is smaller than the reference value, and the second gain larger than the first gain value when the rotational speed is larger than the reference value. Take the value. Thereby, as will be described in detail in the embodiment, the controllability of the rotation speed can be enhanced in a wide range of the rotation speed.

  According to the second aspect of the control device for the electric motor drive device of the present invention, the phase delay increases on the low frequency side as the time constant increases. The peak value of the transfer gain decreases as the time constant increases. The time constant takes the first time constant value when the rotational speed is smaller than the reference value, and takes the second time constant value larger than the first time constant value when the rotational speed is larger than the reference value. The controllability of the rotation speed can be enhanced in a wide range of the rotation speed.

It is a figure which shows an example of a rotation coordinate and a magnetic flux typically. It is a figure which shows an example of a notional structure of a power converter device. It is a figure which shows an example of a transmission gain and a transmission phase. It is a figure which shows an example of a transmission gain and a transmission phase. It is a figure which shows an example of a transmission gain and a transmission phase.

  Before going into the detailed description of the embodiment, the premise of the present invention will be described.

<1. Premise>
FIG. 1 shows a synchronous motor (hereinafter, simply referred to as “motor”. As a special type of synchronous motor, there is a motor that does not have a field such as a switched reluctance motor. Air gap magnetic flux [λ] (symbol [] represents a vector quantity; the same applies hereinafter) and interlinkage magnetic flux [Λ0] (hereinafter referred to simply as “linkage”) to the armature by the field. It is a vector diagram showing the relationship with "magnetic flux". For example, when the motor has a permanent magnet, the linkage flux [Λ0] is generated by the permanent magnet, and when the motor has a field winding, a current flows through the field winding. Generated by flowing.

  A dq rotating coordinate system is introduced as a rotating coordinate system synchronized with the rotation of the electric motor. Here, the d-axis is set in phase with the flux linkage [Λ0], and the q-axis has a phase toward the direction (hereinafter simply referred to as “rotation direction”) that is desired to rotate with respect to the d-axis. Advance 90 degrees.

  Further, a δ-γ rotating coordinate system and a δc-γc rotating coordinate system are introduced as rotating coordinate systems. The phase advances at a phase angle φ in the direction of rotation of the motor with respect to the δ axis with respect to the d axis and the γ axis with respect to the q axis. The phase advances at a phase angle φc toward the rotational direction of the motor with respect to the δc axis relative to the d axis and the γc axis relative to the q axis, respectively. Hereinafter, for convenience of explanation, the phase angle φ of the δ axis with respect to the d axis is referred to as an actual phase angle φ, and the phase angle φc of the δc axis with respect to the d axis is referred to as an estimated phase angle φc.

  For example, in a motor control method known as “primary magnetic flux control”, the δ axis is set in phase with the gap magnetic flux [λ]. In this case, the actual phase angle φ is grasped as a load angle (phase angle between the interlinkage magnetic flux [Λ0] and the gap magnetic flux [λ]).

As is well known, the gap magnetic flux [λ] is a voltage and current supplied to an electric motor (more specifically, an armature winding included in an armature included in the electric motor) and device constants (for example, inductance, armature) of the electric motor. The resistance component of the winding, the flux linkage) and the rotational speed of the motor. Therefore, the estimated value [λ ^] of the air gap magnetic flux [λ] is obtained from the measured values (or command values, estimated values) of the voltage and current, the device constants, and the rotation speed. In the “primary magnetic flux control” described above, the γ-axis component of the command value [λ ^* ] is zero.

When the δc-γc rotational coordinate system is employed in such control, the estimated phase angle φc matches the actual phase angle φ, so that the rotation of the motor can be appropriately controlled. If the device constant, rotational speed, voltage and current applied to the motor are fully understood, the estimated value [λ ^] obtained based on these is controlled to be equal to the command value [λ ^* ]. This is because the gap magnetic flux [λ] matches the command value [λ ^* ].

<2. Configuration of power conversion device>
FIG. 2 is a block diagram showing the configuration of the motor control device 3 according to the present embodiment and its peripheral devices based on the above premise.

  The electric motor 2 is a three-phase electric motor, and includes an armature 21 and a rotor as a field magnet 23. As technical common sense, the armature 21 has an armature winding 22, and the rotor rotates relative to the armature 21. The case where the field magnet 23 is provided with, for example, a magnet that generates an interlinkage magnetic flux will be described.

The voltage supply source 1 includes, for example, a voltage control type inverter and its control unit, and a three-phase voltage command value [v _x ^* ] = [v _u ^* v _v ^* v _w ^* ] ^t (“ t ″ indicates transposition of the matrix. The same applies hereinafter), and three-phase voltages v _u , v _v , and v _w are applied to the motor 2. As a result, a three-phase current [i _x ] = [i _u i _v i _w ] ^t flows in the electric motor 2. However, components included in the voltage command value [v _x ^* ] and the three-phase current [i _x ] are described, for example, in the order of the U-phase component, the V-phase component, and the W-phase component.

The motor control device 3 is a device that controls the air gap magnetic flux [λ] and the rotation speed (rotation angular velocity in the following example) with respect to the motor 2. The gap magnetic flux [λ] is also referred to as a primary magnetic flux, and the flux of the armature reaction generated by the interlinkage magnetic flux [Λ0] and the armature current (which is also the three-phase current [i _x ]) flowing through the armature 21. Is a synthesis of

  The electric motor control device 3 includes a coordinate conversion unit 34, a control signal generation unit 36, a voltage command calculation unit 35, a rotation speed calculation unit 32, and a phase calculation unit 33.

Coordinate conversion unit 34 converts the three-phase current _{[i x]} detected by the current detection unit 5, the current in .delta.c-[gamma] c rotating coordinate system _[i δγc] = in _{[i δc} i _γc] ^t. For this conversion, the rotation angle θ of the δc-γc rotating coordinate system with respect to the fixed coordinate system (for example, the UVW fixed coordinate system or the α-β fixed coordinate system) of the electric motor 2 is used. Since these conversions are realized by a known technique, the details thereof are omitted here.

The control signal generation unit 36 outputs a control signal to the voltage supply source 1 based on the voltage command value [v _δγ ^* ] and the rotation angle θ in the _δc-γc rotating coordinate system. For example, the control signal generator 36 is a coordinate converter, and converts the voltage command value [v _δγ ^* ] of the _δc-γc rotating coordinate system into a three-phase voltage command value [v _x ^* ] based on the rotation angle θ. . The control unit of the voltage supply source 1 controls the voltage control type inverter based on the control signal.

  The phase calculator 33 integrates the rotational speed ω1 and calculates the rotational angle θ. The rotational speed ω1 is obtained as an output of the rotational speed calculation unit 32. The rotation speed calculation unit 32 will be described in detail later.

The voltage command calculation unit 35 _inputs the current [i _δγc ], the rotation speed ω1 and the primary magnetic flux command value [λ ^* ], and outputs the voltage command value [v _δγ ^* ]. This voltage command value [v _δγ ^* ] is calculated based on, for example, a voltage equation. In general, the following voltage equation (1) is established in the δc-γc rotating coordinate system. However, [I], [J] and the symbol [] surrounding these elements indicate a matrix.

Here, R represents indicates the resistance value of the resistance component of the armature winding 22, s represents a differential _{operator, [λ δγc] = [λ} δc λ γc] t , the primary magnetic flux in .delta.c-[gamma] c rotated coordinates [ λ]. The resistance value R can be stored in the voltage command calculation unit 35 in advance.

  Now, since the calculation result by the differential operator s is 0 in the steady state, the voltage equation in the steady state is derived from the equation (1) as the following equation (2).

The voltage command calculation unit 35 adds the feedforward amount [F] and the feedback amount [B] as shown by the equation (3) to obtain the voltage command value [v _δγ ^* ]. The feedforward amount [F] is obtained by _{placing the} primary magnetic flux [λ _δγc ] as the primary magnetic flux command value [λ ^* ] = [λ _δ ^* 0 (= λ _γ ^* )] ^{t on} the right side of the equation (2). It is done. As the feedback amount [B], for example, an amount based on the deviation between the primary magnetic flux [λ _δγc ] and the primary magnetic flux command value [λ ^* ] can be adopted. Specifically, the feedback gain Gλ is multiplied by the deviation between the primary magnetic flux command value [λ ^* ] and the primary magnetic flux [λ _δγc ]. The feedback gain Gλ can be stored in the voltage command calculation unit 35 in advance.

  In the expression (3), the feedback gain Gλ is shown as a scalar quantity, but it may be a non-zero matrix of 2 rows and 2 columns that acts on the deviation of the primary magnetic flux.

Ideally, when the feedback amount [B] is 0, the δ-axis component λ _δ ^* and the δc-axis component λ _δc are the same, and the γ-axis component λ _γ ^* and the γc-axis component λ _γc are the same. Thus, the steady state represented by the equation (2) is realized by the control in the δc-γc rotating coordinate system.

Further, the feedback amount [B] may be obtained from the current deviation. Specifically, the feedback amount [B] is obtained according to Equation (4). However, the feedback gain Gi (≠ 0) and the command value [i _δγ ^* ] = [i _δ ^* i _γ ^* ] ^t of the current [i _δγc ] were introduced. The feedback gain Gi may be a non-zero matrix of 2 rows and 2 columns that acts on the current deviation.

When the feedback amount [B] is obtained in this way, the current command value [i _δγ ^* ] is input to the voltage command calculation unit 35. The current command value [i _δγ ^* ] may be obtained from the primary magnetic flux command value [λ ^* ] based on the relationship between the current and the magnetic flux ([λ] = [L · i] + [Λ0]).

The voltage command value [v _δγ ^* ] may be obtained only from the feedback amount [B].

The rotational speed calculation unit 32 performs a correction using a correction amount based on the current i _γc on the rotational speed command value ω1 ^* to calculate the rotational speed ω1. Such correction is also described in Non-Patent Documents 1 and 2. First, correction of the rotational speed ω1 shown in Non-Patent Document 1 will be described. More specifically, the rotational speed ω1 is calculated using the following equation.

Here, ωrm ^* indicates a rotational speed command for the mechanical rotational speed of the electric motor 2, n indicates the number of pole pairs of the field 23, ωcorr indicates a compensation amount, and Km is a gain (hereinafter referred to as a compensation gain). ). The compensation amount ωcorr is, for example, the product of the current i _γc and the compensation gain Km in a steady state. For simplicity, the rotation direction when the mechanical load of the electric motor 2 is rotating forward, the mechanical rotation speed command value ωrm ^* , the rotation speed ω1 of the rotating coordinate system, and the command value ω1 ^* are rotating forward. In this case, the rotation direction is the same. In other words, for the sake of simplicity, the forward rotation direction with respect to the rotational speed command value ωrm ^* , the rotational speed ω1 and the command value ω1 ^* is set to the same direction as the forward rotation direction of the mechanical load. For example, each value assumes a positive value during forward rotation.

According to the equation (5), correction is performed so that the rotational speed command value ω1 ^* is reduced as the current i _γc increases. Thereby, as explained in Non-Patent Document 1, the control can be further stabilized.

In the illustration of FIG. 2, the rotation speed calculation unit 32 includes a speed command correction unit 321, a speed compensation unit 322, and a subtracter 323. The speed compensation unit 322 receives the current i _γc and outputs a value obtained by multiplying the current i _γc by the compensation gain Km as a correction amount (Km · i _γc ). In the example of FIG. 2, the rotational speed ω <b> 1 is also input to the speed compensation unit 322. This is because in the present embodiment, as will be described in detail later, the compensation gain Km is determined according to the rotational speed ω1.

Speed command correcting unit 321 inputs the correction amount and the rotation speed command value _{^{ωrm * (Km · i γc)}} . However, in FIG. 2, the input of the correction amount to the speed command correction unit 321 is omitted. The speed command correction unit 321 calculates the average value of the correction amount (Km · i _γc ) to calculate the compensation amount ωcorr, and further calculates the rotational speed command value ω1 ^* based on the equation (6). The subtractor 323 receives the rotational speed command value ω1 ^* and the correction amount (Km · iγc), and calculates the rotational speed ω1 by subtracting the correction amount from the rotational speed command value ω1 ^* (formula (5)).

  Next, calculation of the rotational speed ω1 of Non-Patent Document 2 will be described. More specifically, the rotational speed ω1 is calculated using the following equation.

Here, Tm represents a time constant of the high-pass filter. An element indicated by “Tm · s / (1 + Tm · s)” indicates high-pass filtering using a time constant Tm. Therefore, according to Equation (7), the rotational speed ω1 is corrected using the product of the harmonic component of the current i _γc extracted with the time constant Tm and the compensation gain Km. The compensation gain Km can also be grasped as a gain of high-pass filter processing.

Since the current i _γc is considered to be substantially constant in the steady state, “Tm · s / (1 + Tm · s) · i _γc ” is substantially zero, and the rotational speed ω1 is equal to the rotational speed command value ω1 ^* . On the other hand, when the current i _γc fluctuates at a relatively high frequency, the rotational speed ω1 is corrected based on the harmonic component of the current i _γc . Thereby, as explained in Non-Patent Document 2, stable operation of the electric motor 2 with suppressed turbulence is realized.

In this case, the speed command correction unit 321 inputs the rotational speed command value Omegarm ^*, calculates the rotational speed command value .omega.1 ^* using the equation (8). The speed compensation unit 322 receives the current i _γc , performs high-pass filter processing on the current i _γc to remove a direct current component, and outputs a value obtained by multiplying the current i _γc by the compensation gain Km as a correction amount. Even in this case, as will be described in detail later, the compensation gain Km is determined according to the rotational speed ω1. The subtracter 323 calculates the rotational speed ω1 by subtracting the correction amount from the rotational speed command value ω1 ^* .

Also in the present embodiment, similarly to Non-Patent Documents 1 and 2, the rotational speed ω1 is calculated by performing correction with the compensation amount obtained by the product of the value depending on the current i _γc and the compensation gain Km. Furthermore, this embodiment intends to improve the controllability of the rotational speed control by the method of setting the compensation gain Km.

<3. Setting method of compensation gain Km>
<3-1. Compensation of Rotational Speed Based on Equation (7) and Equation (8)>
In order to consider the controllability of the rotational speed control, the transfer function of the output torque τe of the electric motor 2 with respect to the load torque τL of the electric motor 2 is considered. The equations required to calculate this transfer function are listed below.

Expression (9) is an equation of motion of the electric motor 2, and J indicates inertia. Expression (10) is an expression showing the relationship between the rotational speed ω1 of the δc-γc rotational coordinate system and the rotational speed (n · ωrm) of the dq rotational coordinate system (see also FIG. 1), and the rotational speed ω1. , (N · ωrm) is equal to the differential value of the phase angle φc. Expression (11) is an expression indicating the output torque τe. Equation (12) is an expression showing the [gamma] c-axis component lambda _{[gamma] c} of the primary magnetic flux [lambda]. In equation (12), L represents the inductance of the armature winding 22, and Λ0 represents the magnitude of the flux linkage [Λ0] (scalar amount). As is well known, the equation (12) is derived from the fact that the primary magnetic flux [λ] is a combination of the magnetic flux [L · i] and the interlinkage magnetic flux [Λ0] due to the electron reaction.

Assuming that ideal primary magnetic flux control is performed, an approximate expression of λ _{γc ≈0} is established. Further, as described in Non-Patent Documents 1 and 2, the primary magnetic flux control is usually performed in a range where the phase angle φc is small. Therefore, an approximate expression of sin φc≈φc is established. Further, in a steady state, the time change (s · ω1 ^* ) of the rotational speed command value ω1 ^* is approximated to zero.

  Using the calculation formula of the rotational speed ω1 shown in the equations (7) and (8), the equations (9) to (12), and the above approximation, the transfer function of the output torque τe with respect to the load torque τL is derived. The transfer function is expressed by the following equation.

  FIG. 3 shows an example of a gain (hereinafter referred to as a transfer gain) and a phase (hereinafter referred to as a transfer phase) in the frequency characteristic of this transfer function. FIG. 3 shows the transfer gain and the transfer phase when the compensation gain Km takes values Km1 to Km6, respectively. The values Km1 to Km6 increase as the subscript number increases. Therefore, the value Km1 is the smallest and the value Km6 is the largest.

  Since the transfer gain is the ratio of the magnitude (amplitude) of the output torque τe to the magnitude (amplitude) of the load torque τL, the difference between the load torque τL and the output torque τe when the transfer gain moves away from 1 (zero dB). Will increase. As can be understood from the equation (9), the difference between the load torque τL and the output torque τe causes a change in the rotational speed ωrm. Therefore, from the viewpoint of reducing fluctuations in the rotational speed ωrm, it is desirable that the transfer gain is close to 1 (zero dB).

  The transmission phase is a phase difference between the load torque τL and the output torque τe, and this phase difference also causes a difference between the load torque τL and the output torque τe. Therefore, it is desirable that the transmission phase is close to zero. A reduction in the transmission phase in the negative range means that the phase delay of the output torque τe with respect to the load torque τL increases.

  In the present embodiment, when the compensation gain Km is determined based on the rotational speed ω1, first, the influence of the compensation gain Km on the frequency component included in the load torque τL is considered. This is because when the load torque τL changes according to the rotational position, the main frequency component of the load torque τL has a positive correlation with the rotational speed ωrm. For example, when the electric motor 2 drives a one-cylinder compressor, the same frequency component as the rotational speed ωrm is mainly included in the load torque τL. Therefore, when the rotational speed ωrm is small, the frequency of the main frequency component of the load torque τL is small.

  As can be understood from the graph of the transfer gain in FIG. 3, the smaller the compensation gain Km, the higher the Q value of the transfer gain and the peak value, and the frequency at which the peak value is taken (hereinafter also referred to as the resonance frequency). Called) increases as the compensation gain Km decreases. In the illustration of FIG. 3, when the compensation gain Km takes the value Km1, the resonance frequency is the highest and is 20 strong rad / s, and the peak value is the highest and is slightly less than 30 dB.

  As shown in FIG. 3, in the low frequency region (region where the frequency is lower than about 10 rad / s), the transfer gain approaches 1 (zero dB) in a wider range as the compensation gain becomes smaller. Therefore, when the load torque τL includes many frequency components in the low frequency region, a small value is adopted as the compensation gain Km so that the transfer gain approaches 1 (zero dB).

  As can be understood from the graph of the transmission phase in FIG. 3, in the low frequency region, the smaller the compensation gain Km, the closer the transmission phase approaches zero. Therefore, also from the viewpoint of bringing the transmission phase close to zero, when the load torque τL includes many frequency components in the low frequency region, it is desirable to employ a small value as the compensation gain Km.

  By adopting a small value as the compensation gain Km as described above, it is possible to control the fluctuation of the rotational speed ωrm while suppressing the fluctuation in the low frequency region. In other words, the rotational speed can be controlled with high controllability.

  On the other hand, unlike the low frequency region, the high frequency region (region where the frequency is higher than about 10 rad / s) includes the peak value of the transfer gain when the compensation gain Km is small. In the illustration of FIG. 3, the peak values when the compensation gain Km takes the values Km1 and Km2 are both greater than 20 dB. Therefore, when the load torque τL includes a lot of frequency components in the high frequency region, the output torque τe is greatly increased, particularly in the frequency components near the resonance frequency, causing fluctuations in the rotational speed ωrm. Therefore, when the load torque τL includes many frequency components in the high frequency region, it is desirable to employ a larger value for the compensation gain Km in order to reduce the peak value of the transfer gain.

  On the other hand, even in the high frequency region, the phase delay starts to increase from near zero on the lower frequency side as the compensation gain Km increases. However, in the high frequency region, it is desirable to make the transmission gain close to zero (dB) in preference to the transmission phase. The reason is discussed below.

  In the high frequency region, the peak value of the transfer gain when the compensation gain Km is small is included, and this peak value exceeds, for example, 20 dB. Therefore, the amplitude of the output torque τe at the resonance frequency reaches several tens of times the amplitude of the load torque τL at the resonance frequency. In such a case, even if the transmission phase is brought close to zero, the difference between the load torque τL and the output torque τe cannot be reduced efficiently (a few tens of percent of the amplitude of the output torque τe is reduced) I can only do it). For example, if the amplitudes of the load torque τL and the output torque τe are 1 and 50, respectively, the maximum value of the difference between {50 · sin (θ−Δθ)} and {sinθ} is when the transmission phase Δθ is 0 degree. Although the value is minimum, the value is 49 (the reduction amount is 1%). That is, the difference cannot be reduced efficiently. Therefore, in this case, it is desired to reduce the transfer gain itself.

  That is, since the transfer gain is small in the low frequency region, the difference between the output torque τe and the load torque τL is efficiently reduced by reducing the transfer phase. On the other hand, since the transfer gain is large in the high frequency region, the difference is efficiently reduced by preferentially reducing the transfer gain.

  In view of the above contents, in the present embodiment, the compensation gain Km is determined based on the rotational speed ω1. Since the main frequency component of the load torque τL has a positive correlation with the rotational speed ω1, when the rotational speed ω1 is smaller than the reference value ωref, the first gain value K1 is adopted as the compensation gain Km, and the rotational speed ω1. Is greater than the reference value ωref, the second gain value K2 (> K1) is adopted as the compensation gain Km. The gain values K1 and K2 and the reference value ωref can be determined in advance by experiment or simulation, and are stored in the speed compensation unit 322, for example. The speed compensation unit 322 determines the magnitude relationship between the rotational speed ω1 and the reference value ωref, and determines the compensation gain Km as described above based on the determination result.

  The reference value ωref can be determined as follows, for example. That is, the motor 2 is operated using the compensation gain Km (for example, the value Km1) employed when the rotational speed ω1 is low, and the rotational speed ω1 is gradually increased. Then, the rotational speed ω1 when the fluctuation range of the rotational speed ωrm generated when the frequency component (particularly higher-order frequency component) of the load torque τL approaches the resonance frequency may be adopted as the reference value ωref. .

  As described above, according to the present embodiment, the rotation speed of electric motor 2 can be controlled with good controllability in a wider rotation speed range.

  Although the value of the compensation gain Km is set based on the magnitude of the rotational speed ω1, the value of the compensation gain Km may be set more finely. For example, when the rotational speed ω1 is smaller than the first reference value, the first value is adopted, and when the rotational speed ω1 is larger than the first reference value and smaller than the second reference value, the second value (> first Value) and the third value (> second value) may be adopted when the rotational speed ω1 is larger than the second reference value. In short, the compensation gain Km only needs to have a positive correlation with the rotational speed ω1. In other words, the compensation gain Km may be increased in a non-monotonic manner with respect to the rotational speed ω1. In any case, the compensation gain Km takes the first value when the rotation speed ω1 is smaller than a certain reference value, and the second value larger than the first value when the rotation speed ω1 is larger than the reference value. It is because it takes.

In addition, the case where the rotation direction when the mechanical load of the electric motor 2 is rotating forward and the rotation direction when the rotation speed ω1 of the rotating coordinate system is rotating forward is also considered. That is, the case where the direction of forward rotation of the mechanical load and the direction of forward rotation of the rotational speed command value ωrm ^* , rotational speed ω1 and command value ω1 ^* are set opposite to each other is also considered. In this case, the compensation gain Km may be set based on the magnitude of the rotational speed (−ω1) multiplied by (−1) so as to be the same as the rotational direction of the mechanical load.

  Alternatively, the compensation gain Km may be set based on the absolute value of the rotational speed ω1. Thereby, even when the rotational speed ω1 takes both positive and negative (four-quadrant operation), it is possible to appropriately control.

<3-2. Compensation of Rotational Speed Based on Equation (5) and Equation (6)>
In Equation (7), the limit value of “Tm · s / (1 + Tm · s)” is 1 when the time constant Tm approaches infinity. If this limit value is used, Expression (7) matches Expression (5). Therefore, if the compensation amount ωcorr is ignored for the sake of simplicity, in the transfer function of Expression (13), Tm is brought close to infinity, so that the transfer function when the speed compensation of Expression (5) and Expression (6) is used. Can be derived. The transfer function using the speed compensation of the equations (5) and (6) is derived as follows.

  FIG. 4 shows an example of transfer gain and transfer phase in the frequency characteristic of this transfer function. FIG. 4 shows the transmission gain and the transmission phase when the compensation gain Km takes Km1 to Km6, respectively. As can be understood from FIG. 4, the peak value of the transfer gain decreases as the compensation gain Km increases. Further, as the compensation gain Km is increased, the phase delay is increased on the low frequency region side in the region where the frequency is lower than 20 strong rad / s.

  As can be understood from the transmission phase graph of FIG. 4, in the low frequency region, the smaller the compensation gain Km, the smaller the phase delay in the wider frequency region. Further, as can be understood from the graph of the transfer gain, the transfer gain maintains a value close to 1 (zero dB) in a relatively wide range in the low frequency region. Therefore, when the rotational speed ω1 is smaller than the reference value ωref, a small value (first gain value) is adopted as the compensation gain Km.

  On the other hand, the peak value of the transfer gain is included in the high frequency region, and the transfer gain resonates and greatly increases as the compensation gain Km is the smallest value. Therefore, when the rotational speed ω1 is larger than the reference value ωref, a second gain value larger than the first gain value is adopted as the compensation gain Km in order to preferentially reduce the transfer gain.

  Thereby, the rotational speed of the electric motor 2 can be controlled with a favorable rotational speed characteristic in a wider rotational speed range.

<4. Setting method of time constant Tm>
FIG. 5 shows an example of the transfer gain and the transfer phase in the frequency characteristic of the transfer function of Expression (13). FIG. 5 shows the transfer gain and the transfer phase when the time constant Tm takes values Tm1 to Tm5, respectively. The values Tm1 to Tm5 are smaller as the subscript number is smaller. As can be understood from FIG. 5, the peak value of the transfer gain decreases as the time constant Tm increases. Further, as the compensation gain Km is larger, the phase delay is larger on the lower frequency region side in the region where the frequency is lower than 20 strong rad / s.

  As can be understood from the transmission phase graph of FIG. 5, in the low frequency region, the smaller the time constant Tm, the smaller the phase delay in the wider frequency region. Further, as can be understood from the graph of the transfer gain, the transfer gain maintains a value close to 1 (zero dB) in a relatively wide range in the low frequency region. Therefore, when the rotational speed ω1 is smaller than the reference value ωref, the first temporary constant value is adopted as the time constant Tm.

  On the other hand, the peak value of the transfer gain is included in the high frequency region, and when the smallest value Km1 is adopted as the time constant Tm, the transfer gain resonates and greatly increases. Therefore, when the rotational speed ω1 is larger than the reference value ωref, a second time constant value larger than the first temporary constant value is adopted as the time constant Tm in order to preferentially reduce the transfer gain. The first temporary constant value and the second time constant value are determined in advance and stored in the speed compensation unit 322, for example.

  Thereby, the rotational speed of the electric motor 2 can be controlled with a favorable rotational speed characteristic in a wider rotational speed range.

  Similar to the compensation gain Km, the time constant Tm only needs to have a positive correlation with the rotational speed ω1. In other words, the time constant Tm may be increased non-monotonically with respect to the rotational speed ω1.

  The various embodiments described above can be appropriately combined as long as the functions of each other are not impaired.

  The above block diagram is schematic, and each unit may be configured by hardware, or may be configured by a microcomputer (including a storage device) whose function is realized by software. Various procedures executed by each unit or various means or various functions implemented may be realized by hardware.

  The microcomputer executes each processing step (in other words, a procedure) described in the program. The storage device is composed of one or more of various storage devices such as a ROM (Read Only Memory), a RAM (Random Access Memory), a rewritable nonvolatile memory (EPROM (Erasable Programmable ROM), etc.), and a hard disk device, for example. Is possible. The storage device stores various information, data, and the like, stores a program executed by the microcomputer, and provides a work area for executing the program. It can be understood that the microcomputer functions as various means corresponding to each processing step described in the program, or can realize that various functions corresponding to each processing step are realized.

DESCRIPTION OF SYMBOLS 1 Drive device 2 Synchronous motor 21 Armature 22 Armature winding 23 Field vδ ^* , vγ ^* Voltage command value iδ ^* Current command value

Claims (3)

Based on the control signal, an AC voltage is applied to a synchronous motor (2) having a field (23) and an armature (21), and an AC current (iu, iv, iw) is output (1) Control device (3) for controlling
Based on a rotation angle (θ) with respect to a fixed coordinate system of a rotating coordinate system having a δc axis and a γc axis that advances 90 degrees in a predetermined advance direction with respect to the δc axis, the alternating current is expressed in the rotating coordinate system. a coordinate conversion unit (34) for converting into δc-axis current (i _δc ) and γc-axis current (i _γc );
By correcting the command value (ω1 ^* ) of the rotational speed (ω1) of the rotational coordinate system with a correction amount obtained by the product of the compensation gain (Km) and the compensation value depending on the γc-axis current, the rotational speed ( Rotation speed calculation unit for obtaining ω1),
A phase calculator (33) for calculating the rotation angle based on the rotation speed;
A primary magnetic flux (λ), which is a combination of the flux linkage (Λ0) to the armature by the field and the armature reaction magnetic flux generated by the alternating current, is along the δc axis in the rotating coordinate system. A voltage command calculation unit (35) for generating a voltage command value ([v _δγ ^* ]) in the rotating coordinate system for the AC voltage;
A control device for an electric motor drive device comprising a control signal generator (36) that generates the control signal based on the voltage command value and the rotation angle,
The compensation gain takes a first gain value when the rotational speed is smaller than a reference value, and takes a second gain value larger than the first gain value when the rotational speed is larger than the reference value. The control device for the motor drive device.   The compensation value is a harmonic component of the γc-axis current (iγc) extracted with a time constant (Tm), and the time constant is a first time constant value when the rotational speed is smaller than a reference value. 2. The control device for an electric motor drive device according to claim 1, wherein when the rotational speed is larger than the reference value, a second time constant value larger than the first time constant value is adopted. A control device for an electric motor drive device according to claim 1 or 2,
An electric motor drive system comprising the drive device.

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