JP4300494B2 - High frequency power transformer and power conversion device using the same - Google Patents

Description

【００３２】
なお、本実施例において、高周波パワートランス２０の偏磁の原因となる主スイッチ２、３、４、および５のオン期間のバラツキを抑制するため、図３の主スイッチ２、３、４および５にはパワーＭＯＳ−ＦＥＴを用い、これらＭＯＳ−ＦＥＴのターンオフタイムのバラツキを抑制するためターンオフ時のゲート電流波高値を大きく取ることのできるゲート駆動回路を採用した。
【００３３】
また、図３の回路において、帰還ダイオード６、７、８および９は高周波パワートランス２０の励磁エネルギーを入力直流電源１に回生することによりコンバータの高効率化を図るとともに同高周波パワートランス２０の偏磁を抑制する機能を有する。
【００３４】
高周波パワートランス２０には表２に示す磁心を用いた。表２において、磁心イから磁心チはＦｅを主成分とし結晶粒径５０ｎｍ以下の微細な結晶粒がその組織の体積全体の５０％以上を占めるナノ結晶軟磁性合金薄帯巻磁心である。このうち磁心チはカットした上で非磁性の絶縁体ギャップを挿入したもの、それ以外はノーカットである。
また、磁心リはノーカットのＦｅ基非晶質軟磁性合金薄帯巻磁心、磁心ヌはノーカットの部分的に結晶質を含むＦｅ基非晶質軟磁性合金薄帯巻磁心、磁心ルはノーカットのＭｎ−Ｚｎフェライト磁心、磁心ヲはカットした上で非磁性の絶縁体のギャップを挿入したＭｎ−Ｚｎフェライト磁心である。
【００３５】
【表２】

【００３６】
表２の磁心のうち磁心イから磁心ヌは、いずれもその寸法が外径４４ｍｍ、内径２４ｍｍ、高さ２０ｍｍのトロイダル形状であり、その寸法が外径４６.５ｍｍ、内径２１.５ｍｍ、高さ２３ｍｍのプラスチック製絶縁ケースに挿入した。また、磁心ルおよび磁心ヲは、表２に示すよう１２０℃の飽和磁束密度Ｂｓは０.３２Ｔしかないため偏磁に対する余裕度をとるとその動作時の磁束密度の変化量ΔＢは０.３５Ｔ程度しか取れない。そこで、磁心ルおよび磁心ヲはいずれも前記磁心イから磁心ヌと同一寸法の磁心を２ヶ重ね、その寸法が外径４６.５ｍｍ、内径２１.５ｍｍ、高さ４４ｍｍのプラスチック製絶縁ケースに挿入した。
【００３７】
高周波パワートランス２０の巻線仕様を表３に示す。
表３において、本発明Ａから本発明Ｇ、および比較例ｍから比較例ｑに適用したサンドイッチ巻と記載されている巻線構造を図１に示す。図１は、本発明Ａから本発明Ｇ、および比較例ｍから比較例ｑの高周波パワートランス２０の径方向断面の１／４周部分を示したものである。同図において、５０はケースを含む磁心、黒で塗りつぶされている円５１および白抜きの円５２は図３の１次巻線２１、網掛けの円５３および斜線の円５４は各々図３の２次巻線２２と２３に相当するものである。
【００３８】
２次巻線５３および５４は各々０.２３φのポリウレタン絶縁被覆電線を６９本用いて構成したリッツ線を２本パラ合計４本を交互に並列にケースで覆われた磁心５０に略均等にバイファイラ巻した。図１ではその１ターン分の巻線の断面図が示されている。
【００３９】
一方、１次巻線は、０.９φの３層絶縁被覆電線を４本パラでその巻数の１／２にあたる８ターンをケースを含む磁心５０に対し略均等に巻いた巻線５１と０.９φの３層絶縁被覆電線を４本パラで残り１／２の巻数にあたる８ターンをケースを含む磁心５０に対し略均等に巻いた巻線５２で、前記２次巻線５３および５４をサンドイッチ状に挟み込むと同時に、前記巻線５１と巻線５２を直列接続することにより１６ターンの１次巻線２１を構成している。なお、図１では、上記１次巻線の約４ターン分の巻線の断面図が示されている。
【００４０】
【表３】

【００４１】
表３において、比較例ａから比較例ｌに適用した均等巻と記載されている巻線構造を図２に示す。図２は、比較例ａから比較例ｌの高周波パワートランス２０の径方向断面の１／４周部分を示したものである。
同図において、５０はケースを含む磁心、黒で塗りつぶされている円６１は図３の１次巻線２１、網掛けの円６２および斜線の円６３は各々図３の２次巻線２２と２３に相当するものである。
【００４２】
１次巻線６１は、１.４φのテフロンコート絶縁被覆電線を２本パラで磁心５０に対し略均等に１６ターン巻いて構成している。なお、図２では、上記１次巻線６１の約４ターン分の巻線の断面図が示されている。
一方、２次巻線６２および６３は、各々０.２３φのポリウレタン絶縁被覆電線を６９本用いて構成したリッツ線を２本パラ合計４本を交互にならべた形でケースで覆われた磁心５０に略均等にバイファイラ巻されており、図２ではその１ターン分の巻線の断面図が示されている。
【００４３】
表４に、表３に示す２４種類の高周波パワートランス２０について、２次巻線端全てを短絡して１次巻線で測定した漏れインダクタンス、同高周波パワートランス２０を前記回路構成が図３、仕様が表１のフルブリッジ型ＤＣ−ＤＣコンバータに実装したときの同高周波パワートランス２０の偏磁の程度、動作時の磁束密度の変化量ΔＢおよび温度上昇ΔＴを示す。
【００４４】
偏磁については、表１の仕様の範囲において、入力電圧と負荷が急変をしても偏磁による異常がない場合を○、入力電圧と負荷が急変したときのみ偏磁による異常がでた場合を△、定常動作でも偏磁による異常を生じた場合を×とした。
また、動作時の磁束密度の変化量ΔＢおよび温度上昇ΔＴは、周囲温度２５℃において入力電圧２６０Ｖ、出力電圧４０Ｖ、負荷電流３０Ａの入出力条件のもとで連続通電してその値が飽和した時点で測定した結果である。
【００４５】
【表４】

【００４６】
表４からわかるように、サンドイッチ巻構造の低漏れインダクタンスのノーカットのナノ結晶軟磁性合金薄帯巻磁心を用いた本発明ＡからＧの高周波パワートランスによれば、偏磁を実用上問題のないレベルに押さえることができるとともに、その温度上昇ΔＴも実用上支障のない許容値である５０℃以下に押さえることができる。
なお、ここで温度上昇ΔＴの許容値は、表１の動作時の周囲温度の上限である４０℃と動作時のＤＣ−ＤＣコンバータケース内部の温度上昇想定上限値を３０℃を足した７０℃をＥ種絶縁の許容温度である１２０℃から差し引いて５０℃以下とした。
【００４７】
一方、漏れインダクタンスが大きなノーカットのナノ結晶軟磁性合金薄帯巻磁心を用いた比較例ａからｇ、ノーカットのＦｅ基非晶質軟磁性合金薄帯巻磁心を用いた比較例ｉあるいはノーカットのＭｎ−Ｚｎフェライト磁心を用いた比較例ｋの高周波パワートランス２０では、偏磁の影響により最大出力を安定に取り出せず、温度上昇ΔＴの測定が不可能であった。
【００４８】
また、ギャップを設けたナノ結晶軟磁性合金薄帯巻磁心チ、ノーカットの部分的に結晶質を含むＦｅ基非晶質軟磁性合金薄帯巻磁心ヌおよびギャップを設けたＭｎ−Ｚｎフェライト磁心ヲを用いた漏れインダクタンスが大きい比較例ｈ、ｊおよびｌの高周波パワートランス２０は偏磁の影響を押さえることができるが、温度上昇ΔＴが実用上要求される５０℃を超えるため問題となることがわかる。
【００４９】
ギャップ付のナノ結晶軟磁性合金薄帯巻磁心チ、ノーカットのＦｅ基非晶質軟磁性合金薄帯巻磁心リ、ノーカットの部分的に結晶質を含むＦｅ基非晶質軟磁性合金薄帯巻磁心ヌ、ノーカットのＭｎ−Ｚｎフェライト磁心ルおよびギャップを用いたＭｎ−Ｚｎフェライト磁心ヲを用い、サンドイッチ巻構造として低漏れインダクタンスとした比較例ｍからｑの高周波パワートランス２０においては偏磁の影響を押さえることができるが、比較例ｐを除きその温度上昇ΔＴが５０℃を超えるため問題となることがわかる。
【００５０】
一方、比較例ｐでは偏磁の影響を押さえられると同時に温度上昇ΔＴも５０℃以内に押さえることができるが、Ｍｎ−Ｚｎフェライト磁心ヲを用いているためその飽和磁束密度の制約からΔＢをナノ結晶軟磁性合金薄帯巻磁心イからチおよびＦｅ基非晶質軟磁性合金薄帯巻磁心リとヌを用いた場合に比べ約１／２に選定しなくてはならず前記のように磁心体積を２倍としているため高周波パワートランス２０の体積も約２倍になるという問題があることがわかる。
【００５１】
なお、本実施例で得られた結果に基づき詳細に検討した結果、ノーカットのナノ結晶軟磁性合金薄帯巻磁心にセンタータップを持たない１次巻線とセンタータップ付き２次巻線を設けた高周波パワートランス２０において、２次巻線端の全てを短絡して１次巻線端で測定した５０ｋＨｚにおける漏れインダクタンスの値が０.３μＨ以下であれば、実用上、偏磁の影響で図３の回路構成と表１の仕様のフルブリッジ型ＤＣ−ＤＣコンバータの動作に支障をきたすことはないことがわかった。
【００５２】
また、本発明ＡからＧについて比較すると５０ｋＨｚにおける比透磁率μr(50kHz)が２０,０００を超えるナノ結晶軟磁性合金薄帯巻磁心を用いた本発明ＡからＤの場合には、その温度上昇ΔＴが実用上の上限である５０℃に対し１０％以上のマージンを持つ４５℃未満とすることができるため、信頼性が高く高性能な高周波パワートランスと電力変換装置を実現できることがわかる。
【００５３】
（実施例２）
前記実施例１と全く同様の回路構成と仕様において、スイッチング周波数ｆを５０ｋＨｚから２０ｋＨｚに変更したフルブリッジ型ＤＣ−ＤＣコンバータの高周波パワートランス２０、および同フルブリッジ型ＤＣ−ＤＣコンバータの性能について検討した。
【００５４】
高周波パワートランス２０の巻線仕様を表５に示す。なお、比較例ｒ、ｓ、ｔおよびｕを除き、前記実施例１の表２に記載される磁心を用いた表３の巻線仕様のものを使用した。
また、表５において、比較例ｒから比較例ｕについて記載されるサンドイッチ巻と均一巻の構成は、前記実施例１における図１および図２と同様の構成のものである。
【００５５】
【表５】

【００５６】
比較例ｒ、ｓ、ｔおよびｕで用いた磁心の直流磁気特性と５０ｋＨｚにおける比透磁率μr(50kHz)を表６に示す。表６の磁心ワはノーカットのＭｎ−Ｚｎフェライト磁心、磁心カはカットした上で非磁性の絶縁体ギャップを挿入したＭｎ−Ｚｎフェライト磁心である。
また、磁心ワおよび磁心カは、表６に示すように１２０℃の飽和磁束密度Ｂｓが前記表２に示される磁心イから磁心ヌの１／４程度しかないため、それぞれ前記磁心イから磁心ヌと同一寸法の磁心を４ヶ重ねて、その寸法が外径４６.５ｍｍ、内径２１.５ｍｍ、高さ８４ｍｍのプラスチック製絶縁ケースに挿入した。
【００５７】
【表６】

【００５８】
表７に、表５に示す２４種類の高周波パワートランス２０について、２次巻線端全てを短絡して１次巻線で測定した漏れインダクタンス、同高周波パワートランス２０を前記回路構成が図３、仕様が表１のフルブリッジ型ＤＣ−ＤＣコンバータの高周波パワートランス２０に実装したときの偏磁の程度、動作時の磁束密度の変化量ΔＢおよび同高周波パワートランス２０の温度上昇ΔＴを示す。
【００５９】
偏磁については、表１の仕様の範囲において、入力電圧と負荷が急変をしても偏磁による異常がない場合を○、入力電圧と負荷が急変したときのみ偏磁による異常がでた場合を△、定常動作でも偏磁による異常を生じた場合を×とした。
また、動作時の磁束密度の変化量ΔＢおよび温度上昇ΔＴは、周囲温度２５℃において入力電圧２６０Ｖ、出力電圧４０Ｖ、負荷電流３０Ａの入出力条件のもとで連続通電してその値が飽和した時点で測定した結果である。
【００６０】
【表７】

【００６１】
表７からわかるように、サンドイッチ巻構造の低漏れインダクタンスのノーカットのナノ結晶軟磁性合金薄帯巻磁心を用いた本発明ＡからＧの高周波パワートランス２０によれば、偏磁を実用上問題のないレベルに押さえることができるとともに、その温度上昇ΔＴも実用上支障のない５０℃以下に押さえることができる。
【００６２】
これに対し、漏れインダクタンスが大きなノーカットのナノ結晶軟磁性合金薄帯巻磁心を用いた比較例ａからｇ、ノーカットのＦｅ基非晶質軟磁性合金薄帯巻磁心を用いた比較例ｉおよびノーカットのＭｎ−Ｚｎフェライト磁心を用いた比較例ｒの高周波パワートランスでは、偏磁の影響により最大出力を安定に取り出せず、温度上昇ΔＴの測定が不可能であった。
【００６３】
また、ギャップを設けたナノ結晶軟磁性合金薄帯巻磁心チ、ノーカットの部分的に結晶質を含むＦｅ基非晶質軟磁性合金薄帯巻磁心ヌおよびギャップを用いたＭｎ−Ｚｎフェライト磁心カを用いた漏れインダクタンスが大きい比較例ｈ、ｊおよびｓの高周波パワートランス２０は偏磁の影響を押さえることができるが、温度上昇ΔＴが実用上要求される５０℃を超えるため問題となることがわかる。
【００６４】
さらに、ギャップ付のナノ結晶軟磁性合金薄帯巻磁心チ、ノーカットのＦｅ基非晶質軟磁性合金薄帯巻磁心リ、ノーカットの部分的に結晶質を含むＦｅ基非晶質軟磁性合金薄帯巻磁心ヌ、ノーカットのＭｎ−Ｚｎフェライト磁心ワおよびギャップを用いたＭｎ−Ｚｎフェライト磁心カを用い、サンドイッチ巻構造として低漏れインダクタンスとした比較例ｍ、ｎ、ｏ、ｔ、ｕの高周波パワートランスにおいては、偏磁の影響を押さえることができるが、比較例ｔとｕを除き、その温度上昇ΔＴが５０℃を超えるため問題となることがわかる。
【００６５】
一方、比較例ｔおよびｕでは偏磁の影響を実用上十分なレベルに押さえられると同時に温度上昇ΔＴも５０℃以内に押さえることができる。しかし、Ｍｎ−Ｚｎフェライト磁心ヲを用いているため１２０℃の飽和磁束密度の制約からΔＢを０.３８Ｔと、ナノ結晶軟磁性合金薄帯巻磁心イからチおよびＦｅ基非晶質軟磁性合金薄帯巻磁心リとヌを用いた場合の１／４以下に選定しなくてはならないため、前記のように磁心体積を４倍にしているため高周波パワートランス２０の体積も約４倍以上になるという問題があることがわかる。
【００６６】
なお、本実施例で得られた結果に基づき詳細に検討した結果、ノーカットのナノ結晶軟磁性合金薄帯巻磁心にセンタータップを持たない１次巻線とセンタータップ付き２次巻線を設けた高周波パワートランス２０において、２次巻線端の全てを短絡して１次巻線端で測定した５０ｋＨｚにおける漏れインダクタンスの値が０.３μＨ以下であれば、実用上、偏磁による影響で図３の回路構成と表１の仕様の駆動周波数２０ｋＨｚのフルブリッジ型ＤＣ−ＤＣコンバータの動作に支障が生じることはないことがわかった。
【００６７】
また、本発明ＡからＧについて比較すると５０ｋＨｚにおける比透磁率μr(50kHz)が２０,０００を超えるナノ結晶軟磁性合金薄帯巻磁心を用いた本発明ＡからＤの場合には、その温度上昇ΔＴが実用上の上限である５０℃に対し２０％以上のマージンを持つ４０℃未満とすることができるため、信頼性が高く高性能な高周波パワートランス２０およびこれを用いた電力変換装置を実現できることがわかる。
【００６８】
さらに、本発明Ｃ、ＤおよびＥのように直流磁気特性における残留磁束密度Ｂｒと飽和磁束密度Ｂｓの比である角型比Ｂｒ／Ｂｓが０.２以下の場合には、動作時の磁束密度の変化量ΔＢを表７に示す１.６９Ｔの約１.１倍の１.８６Ｔまで大きくしても偏磁の影響による本ＤＣ−ＤＣコンバータの安定動作に支障がないことが確認された。
【００６９】
この結果、高周波パワートランス２０の１次巻線２１の巻数を１ターン減少させることが可能となるが、本実施例では、入力電圧と出力電圧の関系から１次巻線と２次巻線の巻数比を４：１とする必要があり、巻数を変更することは得策でない。
そこで、巻線仕様を同一としたままで、磁心ハ、ニおよびホの断面積を１０％減じることによりΔＢを１.８６Ｔに増加し、磁心および高周波パワートランスの体積を小型化して動作を確認した結果、偏磁の影響による問題もなく、その温度上昇ΔＴも５０℃未満のより小型軽量の高周波パワートランス２０が得られた。
【００７０】
（実施例３）
同一磁心寸法の高周波パワートランスを前記図３の回路構成のフルブリッジ型ＤＣ−ＤＣコンバータに実装し、入力電圧ＤＣ２６０Ｖ、出力電圧４０Ｖ、周囲温度２５℃での同高周波パワートランスの温度上昇が５０℃のときに、出力できる最大電力を、駆動周波数を２ｋＨｚから２００ｋＨｚまで変えて測定した。
【００７１】
本実施例で使用した高周波パワートランス２０の巻線仕様を表８から表１４に示す。表８は駆動周波数２ｋＨｚ、表９は５ｋＨｚ、表１０は１０ｋＨｚ、表１１は２０ｋＨｚ、表１２は５０ｋＨｚ、表１３は１００ｋＨｚおよび表１４は２００ｋＨｚでの検討に用いたものである。上記表８から表１４の高周波パワートランス２０の磁心には、前記表２の磁心イから磁心ヌ、および表１５に示す磁気特性の磁心ヨおよび磁心タを使用した。表１５の磁心ヨはノーカットのＭｎ−Ｚｎフェライト磁心、磁心タはカットした上で非磁性の絶縁体ギャップを挿入したＭｎ−Ｚｎフェライト磁心である。
【００７２】
なお、表８から表１４に記載される巻線構造におけるサンドイッチ巻と均等巻は前記実施例１および実施例２の場合と同様の巻線構造を意味する。また、表８から表１４に示すように高周波パワートランス２０の巻線の巻数が駆動周波数により異なるのは、それぞれの駆動周波数で最も大きな出力電力を取り出せるように決定したためである。
【００７３】
前記表８から表１４の高周波パワートランス２０を前記図３の回路構成のフルブリッジ型ＤＣ−ＤＣコンバータに実装し、入力電圧ＤＣ２６０Ｖ、出力電圧４０Ｖ、周囲温度２５℃の同高周波パワートランスの温度上昇が５０℃のとき、出力できる最大電力を駆動周波数を２ｋＨｚから２００ｋＨｚまで変えて測定した結果を表１６に示す。
表１６において、−は偏磁などの影響で測定ができなかったことを示す。また、()内に示すのはその最大出力電力が得られた高周波パワートランス２０の巻線構造を示すもので、表８から表１４に対応している。
【００７４】
【表８】

【００７５】
【表９】

【００７６】
【表１０】

【００７７】
【表１１】

【００７８】
【表１２】

【００７９】
【表１３】

【００８０】
【表１４】

【００８１】
【表１５】

【００８２】
表１６からわかるように、本発明によれば５ｋＨｚ以上１００ｋＨｚ以下駆動周波数範囲において、偏磁の影響による不安定な動作をすることなしに、同一磁心体積で比較例よりも大きな出力電力を安定に供給しうる高周波パワートランス２０が得られる。
【００８３】
また、表１６に示す本発明の高周波パワートランス２０について、２次巻線端の全てを短絡して１次巻線端で測定した５０ｋＨｚにおける漏れインダクタンスは、いずれも０.３μＨ以下の値を示した。
一方、同表に示す比較例のうちノーカットのナノ結晶軟磁性合金薄帯巻磁心である磁心イから磁心トに均等巻の巻線構造とした高周波パワートランス２０において、同様にして測定した漏れインダクタンスはいずれも０.３μＨを超えており、このためこれらの高周波パワートランス２０では偏磁の影響により不安定な動作を引き起こし、安定して出力を取り出せなかったこともわかった。
【００８４】
以上から、ノーカットのナノ結晶軟磁性合金薄帯巻磁心にセンタータップを持たない１次巻線とセンタータップ付きの２次巻線を設けた高周波パワートランス２０においては、その２次巻線端の全てを短絡して１次巻線端で測定した５０ｋＨｚにおける漏れインダクタンスの値が０.３μＨ以下であれば、実用上偏磁の影響による不安定な動作を示すことなしに安定に出力電力を取り出すことができ、信頼性の高い高周波パワートランス２０およびこれを用いた高効率で高信頼性のフルブリッジ型ＤＣ−ＤＣコンバータが得られることがわかった。
【００８５】
【表１６】

【００８６】
また、表１６において、５０ｋＨｚにおける比透磁率μr(50kHz)が２０,０００以上の値を示すノーカットのナノ結晶軟磁性合金薄帯巻磁心である磁心イから磁心ニにセンタータップを持たない１次巻線とセンタータップ付きの２次巻線を設けた本発明の高周波パワートランス２０は高周波領域での磁心損失が小さいため、２０ｋHz以上を超える高周波において、より大きな出力電力を取り出すことができ、特に高周波での仕様に適することがわかる。
【００８７】
さらに、表１６において、直流磁気特性における残留磁束密度Ｂｒと飽和磁束密度Ｂｓの比である角型比Ｂｒ／Ｂｓが０.２以下のノーカットのナノ結晶軟磁性合金薄帯巻磁心である磁心ハから磁心トにセンタータップを持たない１次巻線とセンタータップ付きの２次巻線を設けた本発明の高周波パワートランス２０は、１０ｋHz以下の周波数で動作時の磁束密度の変化量を大きく取ることができ表８、表９および表１０のように巻数を少なくできるため、より大きな出力電力を取り出すことができることがわかる。
【００８８】
【発明の効果】
以上説明したように、本発明によれば、特に複雑な偏磁抑制回路を設けることなしに実用上障害となる偏磁の発生を抑制し得るとともに低磁心損失で小型のノーカットのナノ結晶軟磁性合金薄帯巻磁心を用いた小型で温度上昇が小さく高効率の高周波パワートランス、およびこれを用いた高効率で信頼性の高い電力変換装置が得られる。
なお、前記実施例では、高周波パワートランスを用いた代表的な電力変換装置としてフルブリッジ型ＤＣ−ＤＣコンバータへの応用例について詳細に説明したが、本発明はハーフブリッジ型コンバータを始めとするセンタータップを持たない１次巻線と、少なくとも1組以上のセンタータップ付き２次巻線を設けた高周波パワートランス全般、および同高周波パワートランスを用いた電力変換装置全般に適用され、同様に有効な効果を発揮し、その効果は極めて大きい。
【図面の簡単な説明】
【図１】本発明による、高周波パワートランス２０の１実施例の巻線構造断面の概念図。
【図２】比較例である高周波パワートランスの巻線構造断面の概念図。
【図３】フルブリッジ型ＤＣ−ＤＣコンバータの回路構成ブロック図。
【図４】図３のフルブリッジ型ＤＣ−ＤＣコンバータの高周波パワートランス１０の１次巻線２１の端子電圧概念図。
【図５】偏磁がない場合の図３のフルブリッジ型ＤＣ−ＤＣコンバータの高周波パワートランス２０の動作Ｂ−Ｈループ概念図。
【図６】偏磁により磁心が飽和したときの図３のフルブリッジ型ＤＣ−ＤＣコンバータの高周波パワートランス２０の動作Ｂ−Ｈループ概念図。
【符号の説明】
１：入力直流電源
２、３、４、５：主スイッチ素子
６、７、８、９：帰還ダイオード
２０：高周波パワートランス
２１：高周波パワートランス２０の１次巻線
２３：高周波パワートランス２０の２次巻線
３１、３２：出力整流ダイオード
３３：出力平滑チョークコイル
３４：出力平滑コンデンサ
３５、３６：出力端子
３７：負荷[0001]
BACKGROUND OF THE INVENTION
The present invention does not have a center tap in a non-cut nanocrystalline soft magnetic alloy ribbon core in which fine crystal grains having a crystal grain size of 50 nm or less occupy 50% or more of the entire volume of the structure. The present invention relates to a high-frequency power transformer provided with a secondary winding and at least one set of secondary windings with a center tap, and a power converter using the same.
[0002]
[Prior art]
A full bridge type DC-DC converter shown in FIG. 3 is used as one of insulated power converters using a high-frequency power transformer. In FIG. 3, 1 is an input DC power source, 2, 3, 4 and 5 are main switches, 6, 7, 8 and 9 are feedback diodes, 20 is a primary winding without a center tap and a secondary winding with a center tap. A high-frequency power transformer provided with wires, 21 is a primary winding of the high-frequency power transformer 20, 22 and 23 are secondary windings of the high-frequency power transformer 20, 31 and 32 are output rectifier diodes, and 33 is an output smoothing choke coil , 34 are output smoothing capacitors, 35 and 36 are output terminals, and 37 is a load.
[0003]
In the full-bridge type DC-DC converter of FIG. 3, the main switches 2 and 3 and 4 and 5 are each set as one set of switches, and these two sets of switches are alternately switched. A high-frequency voltage such as v21 in FIG. 4 is applied to the primary winding 21. From the secondary windings 22 and 23 of the high-frequency power transformer 20, output rectifier diodes 31 and 32, a smoothing choke coil 33, a capacitor smoothing 34 Electric power is supplied to the load 37 via. In FIG. 4, the period in which the main switches 2 and 3 are on is Ton23, the period in which the main switches 4 and 5 are on is Ton45, and Tp is the period. (Ton23 + Ton45) / Tp is the on-duty ratio Don, and in order to keep the output voltage Vo constant with respect to the fluctuation of the voltage E of the input DC power supply 1 and the fluctuation of the load 37, the control is performed by changing Don with constant Tp. PWM control (pulse width modulation control) is generally used. The driving frequency f of the high frequency power transformer 20 is given by 1 / Tp.
[0004]
An operation BH loop conceptual diagram of the high-frequency power transformer 20 in this converter is shown in FIG. The direction of the magnetic field generated in the high-frequency power transformer 20 when current flows from the black circle side of the primary winding 21 of the high-frequency power transformer 20 shown in FIG. 3 is on the positive side of the H axis in FIG. Therefore, during the period Ton23 in which the main switches 2 and 3 are on, the magnetic flux density of the high-frequency power transformer 20 changes by 2 Bm from the point a to the point b in FIG. 5, and in the period Ton45 in which the main switches 4 and 5 are on. The magnetic flux density of the high-frequency power transformer 20 changes by -2 Bm from the point b to the point a in FIG. That is, the high frequency power transformer 20 in this converter operates to draw a symmetric minor loop with respect to the origin of the BH loop.
[0005]
In the high-frequency power transformer 20 in this converter, downsizing and low loss are important issues. As a general technique for reducing the size of the high-frequency power transformer 20, the drive frequency is increased. However, an extremely high frequency that does not consider the high frequency characteristics of elements such as the magnetic core used in the high frequency power transformer 20, the main switches 2, 3, 4 and 5, the feedback diodes 6, 7, 8 and 9, or the output rectifier diodes 31 and 32. The increase not only increases the loss of these elements, but also increases the loss of the high-frequency power transformer 20, causing a decrease in efficiency of the converter and a decrease in reliability due to an excessive temperature rise.
[0006]
For the high-frequency power transformer 20 in this converter, it is generally necessary to select a magnetic core that can be miniaturized and has a low loss at a driving frequency selected in consideration of the high-frequency characteristics of the main switches 2, 3, 4, and 5. .
[0007]
For example, when the output power is relatively small up to about several kW, a power MOS-FET is normally selected for the main switches 4 and 5, and the drive frequency is selected to be about 50 kHz or more. In this case, the magnetic core of the high-frequency power transformer 20 is conventionally an Mn—Zn ferrite core having a small saturation magnetic flux density Bs of about 0.5 T at room temperature but a small core loss at several hundred kHz or more. It was.
[0008]
On the other hand, in the region where the output power exceeds several kW, the IGBT is generally selected for the main switches 2, 3, 4 and 5, and the drive frequency is selected from several kHz to about 20 kHz.
In this case, Sakakibara, Saito, Kamo, Toyota, Yamauchi, Yoshizawa: “Inverter Transformer Using a Microcrystalline Alloy for Iron Core”, IEEJ Technical Report, MAG-90-194, December 6, 1990 ( Hereinafter, the magnetic core of the high-frequency power transformer 20 has a magnetic core loss at 20 kHz when the saturation magnetic flux density at room temperature exceeds 1 T, which is more than twice that of the Mn-Zn ferrite core. Fe-based nanocrystalline soft magnets, which are mainly composed of Fe as described in JP-A-63-302504, and fine crystal grains having a crystal grain size of 50 nm or less occupy 50% or more of the entire volume of the structure. It is known that an alloy ribbon magnetic core is excellent.
[0009]
In the Fe-based nanocrystalline soft magnetic alloy ribbon core, the above-mentioned document 1, or Fukunaga, Koga, Eguchi, Ota, Kakehashi: “Magnetic properties of gap-cut cores using iron-based magnetic ribbons; The wound magnetic core is formed by resin impregnation treatment or surface fixing treatment as described in the meeting material, MAG-89-203, December 1, 1989 (hereinafter referred to as Reference 2). It is known that the core loss increases remarkably when resin or varnish penetrates between layers of the nanocrystalline soft magnetic alloy ribbon and stress is applied to the nanocrystalline soft magnetic alloy ribbon.
[0010]
By the way, in the magnetic core for the high frequency power transformer 20, a cut magnetic core is widely used in order to facilitate winding work. In order to cut the nanocrystalline soft magnetic alloy ribbon core, the core is impregnated with an impregnation material such as an epoxy adhesive, and the layers of the nanocrystalline soft magnetic alloy ribbon constituting the wound core are separated. After fixing with the impregnating material, it is necessary to cut with a rotating grindstone or the like. In addition, after the cut, the end face is also mirror-polished. However, as described in the literature 1 and literature 2, the core loss of the magnetic core obtained by cutting the nanocrystalline soft magnetic alloy ribbon core using such a technique is remarkably increased.
[0011]
For this reason, Yoshizawa, Mori, Arakawa, Yamauchi: "High-frequency magnetic properties of Fe-Cu-Nb-Si-B-based nanocrystalline alloys to reduce the core loss of cores cut using nanocrystalline soft magnetic alloy ribbons. As described in the Institute of Electrical Engineers of Japan, MAG-94-202, November 22, 1994 (hereinafter referred to as Reference 3), the saturation magnetostriction constant λs is 10 ^-6 Although it is effective to use a small nanocrystalline soft magnetic alloy ribbon as below and to perform interlayer insulation treatment with the ribbon surface covered with ceramics, even in that case, normal nanocrystalline soft magnetism that does not impregnate or cut This is about 1.4 times the core loss of the alloy ribbon magnetic core.
[0012]
For this reason, in order to effectively utilize the low core loss that is characteristic of nanocrystalline soft magnetic alloy ribbon cores, the nanocrystalline soft magnetic alloy ribbon that constitutes the core is configured so that stress is not applied as much as possible. There is a need to. As a nanocrystalline soft magnetic alloy ribbon core with such a configuration, the uncut core is housed in an insulating case such as plastic or ceramic using silicon grease or gel-like silicon rubber as a cushioning material. In general, those in which it is difficult for the stress to be directly applied to the wound magnetic core are used.
[0013]
[Problems to be solved by the invention]
As in the full-bridge type DC-DC converter of FIG. 3, the power conversion is performed such that the magnetic flux density of the high-frequency power transformer 20 draws a BH minor loop that is symmetrical with respect to the origin of the BH loop as shown in FIG. In the high frequency power transformer 20 of the apparatus, the high frequency power transformer 20 is magnetically saturated and excited by the demagnetization in which the BH minor loop at the time of operation operates asymmetrically with respect to the origin of the BH loop as shown in FIG. It is extremely important to prevent the current from increasing significantly and to ensure safe operation of the main switches 2, 3, 4 and 5.
[0014]
As is well known, the magnetic bias of the high-frequency power transformer 20 is mainly caused by variations in the electrical characteristics of the main switches 2, 3, 4 and 5, and the excitation current depends on the circuit impedance. Equilibrate by value. However, when the high-frequency power transformer 20 has a large demagnetization, the BH minor loop during operation reaches one saturation region as shown in FIG. An excessive current flows between the main electrodes 5 and 5, and the main switches 2, 3, 4 and 5 sometimes break down. In particular, when the voltage of the input DC power supply 1 is suddenly changed or the load 27 is suddenly changed, the amount of change ΔB in the magnetic flux density during the operation of the high-frequency power transformer 20 becomes transiently large. The main switches 2, 3, 4 and 5 have a high risk of destruction.
[0015]
When an overcurrent protection circuit with a high response speed that suppresses overcurrent flowing between the main electrodes of the main switches 2, 3, 4, and 5 is provided, the main switches 2, 3, It is possible to suppress an excessive current from flowing through 4 and 5, and to prevent these main switches from being destroyed. However, when the overcurrent protection circuit operates, there is a problem that it is impossible to ensure the constant voltage accuracy of the output voltage because sufficient power cannot be supplied to the output side.
[0016]
As the most general method for preventing the magnetism of the high-frequency power transformer 20, a magnetic core having a small magnetic permeability is adopted for the high-frequency power transformer 20, and the operating magnetic flux density peak value Bm of the same magnetic core is used. Selection is made so as to be a sufficiently small value with respect to the saturation magnetic flux density Bs. As a method for obtaining a magnetic core having a low magnetic permeability, the simplest method is to provide a gap in the cut magnetic core to reduce its effective relative magnetic permeability. According to this method, there is also an advantage that the effective relative magnetic permeability of the magnetic core can be arbitrarily selected by adjusting the gap width.
[0017]
However, in the case of a nanocrystalline soft magnetic alloy ribbon core, a method for reducing the effective relative permeability by providing a gap as described above is also described in the literature 1 to literature 3, By providing a cut magnetic core and providing a gap, the core loss greatly increases, so that the characteristic of low core loss, which is the greatest advantage when used in the high-frequency power transformer 20, is impaired, and leakage that occurs in the gap portion There was a problem that copper loss increased due to the influence of magnetic flux.
[0018]
In a nanocrystalline soft magnetic alloy ribbon core, a method for reducing the relative permeability without providing a gap is to heat-treat while applying a magnetic field in the ribbon width direction (winding core height direction) of the winding core. There are a technique and a technique of applying stress to the nanocrystalline soft magnetic alloy ribbon forming the wound magnetic core. However, in a composition system with a relatively small magnetic core loss, the level of the relative magnetic permeability that can be achieved by using the former method reaches several tens of thousands at 50 kHz, which is a value that is one digit larger than several thousand of Mn—Zn ferrite magnetic cores. However, the level did not reach a level sufficient to prevent demagnetization. On the other hand, the latter method has a problem in that the core loss is significantly increased as described in the above-mentioned documents 1 and 2.
[0019]
Therefore, like the full-bridge type DC-DC converter of FIG. 3, the operation of drawing a BH minor loop in which the magnetic flux density of the high-frequency power transformer 20 is symmetrical with respect to the origin of the BH loop as shown in FIG. In order to use the nanocrystalline soft magnetic alloy ribbon core for the high-frequency power transformer 20 of the power converter to perform the characteristic of low magnetic core loss, the high-frequency transformer 20 of the high-frequency transformer 20 is controlled in order to suppress extreme demagnetization. In addition to detecting the amount of bias, a bias suppression circuit that can independently control the on-period of each of the two switches formed by the main switches 2, 3, 4, and 5 is added to correct this. There is a problem that the number of parts increases.
[0020]
In the above description, the full-bridge type DC-DC converter is taken as an example to describe the problems of the high-frequency power transformer 20 using the nanocrystalline soft magnetic alloy ribbon core and the power conversion device using the same. High-frequency power of other power converters such as a half-bridge converter that has a primary winding without a center tap and at least one set of secondary windings with a center tap in the crystalline soft magnetic alloy ribbon core The transformer 20 and the power conversion device using the same have the same problem.
[0021]
The object of the present invention is to prevent the occurrence of practically obstructed levels of magnetic demagnetization, which has been difficult to realize with the above-described prior art, and has a low core loss and a small uncut nanocrystalline soft magnetic alloy ribbon wound core. To provide a high-frequency power transformer having a primary winding without a center tap and at least one set of secondary windings with a center tap, and a highly efficient and reliable power converter using the same. is there.
[0022]
[Means for Solving the Problems]
In the present invention, fine crystal grains containing Fe as a main component and having a crystal grain size of 50 nm or less occupy 50% or more of the entire volume of the structure. , The relative permeability at 50 kHz is 20,000 or more, and the squareness ratio Br / Bs which is the ratio of the residual magnetic flux density Br and the saturation magnetic flux density Bs in the DC magnetic characteristics is 0.2 or less, In a high frequency power transformer provided with a primary winding having no center tap on an uncut nanocrystalline soft magnetic alloy ribbon core and at least one set of secondary windings with a center tap, At least one set of the secondary windings with the center tap is formed by bifilar winding the windings constituting the windings substantially uniformly on the nanocrystalline soft magnetic alloy ribbon magnetic core. Sandwiched by a primary winding wound substantially uniformly around a nanocrystalline soft magnetic alloy ribbon core, and The high frequency power transformer is characterized in that the leakage inductance value at 50 kHz measured at the primary winding end with all the secondary winding ends short-circuited is 0.3 μH or less.
[0023]
By adopting such a configuration, a primary winding having no center tap and at least one set of secondary windings with a center tap are provided on the uncut nanocrystalline soft magnetic alloy ribbon core having a low magnetic core loss. Magnetic saturation due to magnetic bias, which has been a problem when using high-frequency power transformers in power converters such as full-bridge converters and half-bridge converters, can be prevented without adding a complicated magnetic bias suppression circuit. The loss of the high frequency power transformer can be further reduced, and further miniaturization and higher efficiency can be achieved. preferable.
[0027]
In the high-frequency power transformer, when the driving frequency is selected in the range of 5 kHz or more and 100 kHz or less, it is preferable that the selected frequency can be reduced in size and efficiency as compared with the conventional high-frequency power transformer.
[0028]
The power conversion device using the high-frequency power transformer according to the present invention can be reduced in size and efficiency as compared with the conventional power conversion device, and the excitation current can be increased by biasing the high-frequency power transformer with a simple circuit configuration. Therefore, it is preferable that safe operation of the main switch can be achieved and reliability is improved.
[0029]
【Example】
Examples of the present invention will be described in detail below.
Example 1
The performance of the high-frequency power transformer 20 of the full-bridge type DC-DC converter whose circuit configuration is shown in FIG. 3 and whose specification is given in Table 1 and whose switching frequency f is 50 kHz was examined.
[0030]
In FIG. 3, 1 is an input DC power source, 2, 3, 4 and 5 are main switches, 6, 7, 8 and 9 are feedback diodes, 20 is a primary winding without a center tap and a secondary winding with a center tap. A high-frequency power transformer provided with wires, 21 is a primary winding of the high-frequency power transformer 20, 22 and 23 are secondary windings of the high-frequency power transformer 20, 31 and 32 are output rectifier diodes, and 33 is an output smoothing choke coil , 34 are output smoothing capacitors, 35 and 36 are output terminals, and 37 is a load.
[0031]
[Table 1]

[0032]
In this embodiment, the main switches 2, 3, 4, and 5 shown in FIG. 3 are suppressed in order to suppress variations in the ON period of the main switches 2, 3, 4, and 5 that cause the bias of the high-frequency power transformer 20. A power MOS-FET is used, and a gate driving circuit that can take a large gate current peak value at the time of turn-off is employed in order to suppress variations in turn-off time of these MOS-FETs.
[0033]
In the circuit of FIG. 3, the feedback diodes 6, 7, 8 and 9 improve the efficiency of the converter by regenerating the excitation energy of the high frequency power transformer 20 to the input DC power source 1, and also offset the high frequency power transformer 20. It has a function of suppressing magnetism.
[0034]
A magnetic core shown in Table 2 was used for the high-frequency power transformer 20. In Table 2, the magnetic core i to magnetic core H are nanocrystalline soft magnetic alloy ribbon cores in which fine crystal grains having Fe as a main component and a crystal grain size of 50 nm or less occupy 50% or more of the entire volume of the structure. Of these, the magnetic core is cut and inserted with a non-magnetic insulator gap, and the others are uncut.
In addition, the magnetic core is an uncut Fe-based amorphous soft magnetic alloy ribbon core, the magnetic core is an uncut partially crystalline Fe-based amorphous soft magnetic alloy ribbon core, and the magnetic core is an uncut magnetic core. The Mn—Zn ferrite magnetic core is a Mn—Zn ferrite magnetic core that has been cut and inserted with a nonmagnetic insulating gap.
[0035]
[Table 2]

[0036]
Of the magnetic cores in Table 2, the magnetic cores a to n have a toroidal shape with an outer diameter of 44 mm, an inner diameter of 24 mm, and a height of 20 mm. The dimensions are an outer diameter of 46.5 mm, an inner diameter of 21.5 mm, and a height. It was inserted into a 23 mm plastic insulating case. In addition, as shown in Table 2, the magnetic core and the magnetic core have a saturation magnetic flux density Bs at 120 ° C. of only 0.32 T. Therefore, if a margin for demagnetization is taken, the amount of change ΔB in the magnetic flux density during the operation is 0.35 T. Only take a degree. Therefore, both the magnetic core L and the magnetic core W are stacked with two magnetic cores having the same dimensions as those of the magnetic core A and inserted into a plastic insulating case having an outer diameter of 46.5 mm, an inner diameter of 21.5 mm, and a height of 44 mm. did.
[0037]
Table 3 shows the winding specifications of the high-frequency power transformer 20.
In Table 3, FIG. 1 shows a winding structure described as a sandwich winding applied to Invention A to Invention G and Comparative Example m to Comparative Example q. FIG. 1 shows a ¼ circumferential portion of the radial cross section of the high-frequency power transformer 20 of the present invention A to the present invention G and the comparative example m to the comparative example q. In FIG. 3, 50 is a magnetic core including a case, black circle 51 and white circle 52 are primary winding 21, shaded circle 53 and shaded circle 54 in FIG. This corresponds to the secondary windings 22 and 23.
[0038]
The secondary windings 53 and 54 are bifilar substantially evenly in a magnetic core 50 covered with a case of two Litz wires each composed of 69 polyurethane insulated wires of 0.23φ and a total of four litz wires in parallel. I wound it. FIG. 1 shows a cross-sectional view of the winding for one turn.
[0039]
On the other hand, the primary winding is composed of four windings of 0.9φ three-layer insulation-covered electric wires, and eight windings, which are half of the number of turns, wound about the magnetic core 50 including the case 51 and 0.00. The secondary windings 53 and 54 are sandwiched by a winding 52 in which four 9-dia. Three-layer insulation-coated wires are wound approximately evenly on the magnetic core 50 including the case, with eight remaining turns corresponding to ½ windings. At the same time, the winding 51 and the winding 52 are connected in series to form a 16-turn primary winding 21. Note that FIG. 1 shows a cross-sectional view of about four turns of the primary winding.
[0040]
[Table 3]

[0041]
In Table 3, FIG. 2 shows a winding structure described as equal winding applied to Comparative Example a to Comparative Example l. FIG. 2 shows a ¼ circumferential portion of the radial cross section of the high-frequency power transformer 20 of Comparative Example a to Comparative Example l.
In the figure, 50 is a magnetic core including a case, black circle 61 is primary winding 21 in FIG. 3, shaded circle 62 and shaded circle 63 are secondary winding 22 in FIG. 23.
[0042]
The primary winding 61 is formed by winding a 1.4φ Teflon-coated insulating coated electric wire approximately equally 16 turns around the magnetic core 50 with two paras. Note that FIG. 2 shows a cross-sectional view of about four turns of the primary winding 61.
On the other hand, each of the secondary windings 62 and 63 has a magnetic core 50 covered with a case in which four litz wires, each consisting of 69 0.23φ polyurethane insulation-coated electric wires, are alternately arranged. 2 is a bifilar winding, and FIG. 2 shows a sectional view of the winding for one turn.
[0043]
Table 24 shows the leakage inductance of the 24 types of high-frequency power transformers 20 shown in Table 3 that are short-circuited at the ends of the secondary windings and measured by the primary winding. The circuit configuration of the high-frequency power transformer 20 is shown in FIG. The degree of demagnetization of the high-frequency power transformer 20 when the specification is mounted on the full-bridge type DC-DC converter shown in Table 1, the amount of change ΔB in magnetic flux density during operation, and the temperature rise ΔT are shown.
[0044]
As for the demagnetization, within the specification range of Table 1, the case where there is no abnormality due to the demagnetization even if the input voltage and the load change suddenly, the case where the abnormality due to the demagnetization occurs only when the input voltage and the load change suddenly Δ, and the case where abnormality due to the bias magnetism occurs even in the steady operation is marked with ×.
In addition, the amount of change ΔB in magnetic flux density during operation and the temperature rise ΔT were saturated by continuous energization under an input / output condition of an input voltage of 260 V, an output voltage of 40 V, and a load current of 30 A at an ambient temperature of 25 ° C. It is the result measured at the time.
[0045]
[Table 4]

[0046]
As can be seen from Table 4, according to the high-frequency power transformers of the present invention A to G using the non-cut nanocrystalline soft magnetic alloy ribbon core having a sandwich winding structure and a low leakage inductance, there is no practical problem with the demagnetization. In addition to being able to be held down to a level, the temperature rise ΔT can also be kept below 50 ° C., which is a practically acceptable tolerance.
Here, the allowable value of the temperature rise ΔT is 70 ° C. obtained by adding 40 ° C., which is the upper limit of the ambient temperature during operation shown in Table 1, and 30 ° C. to the upper limit of temperature rise inside the DC-DC converter case during operation. Was subtracted from 120 ° C., which is the allowable temperature for E-type insulation, to 50 ° C. or less.
[0047]
On the other hand, Comparative Examples a to g using an uncut nanocrystalline soft magnetic alloy ribbon core having a large leakage inductance, Comparative Example i using an uncut Fe-based amorphous soft magnetic alloy ribbon core, or Uncut Mn In the high-frequency power transformer 20 of Comparative Example k using a —Zn ferrite magnetic core, the maximum output could not be stably taken out due to the influence of the demagnetization, and the temperature rise ΔT could not be measured.
[0048]
In addition, a nanocrystalline soft magnetic alloy ribbon core having a gap, a Fe-based amorphous soft magnetic alloy ribbon core having a partially cut crystalline structure, and a Mn-Zn ferrite core having a gap are also provided. Although the high frequency power transformers 20 of comparative examples h, j, and l having a large leakage inductance using S can suppress the influence of bias, the temperature rise ΔT exceeds 50 ° C. that is practically required, which may cause a problem. Recognize.
[0049]
Nanocrystalline soft magnetic alloy ribbon core with gap, uncut Fe-based amorphous soft magnetic alloy ribbon core, uncut partially crystalline Fe-based amorphous soft magnetic alloy ribbon In the high-frequency power transformer 20 of Comparative Examples m to q, which uses a magnetic core, an uncut Mn-Zn ferrite magnetic core, and a Mn-Zn ferrite magnetic core using a gap and has a low-leakage inductance as a sandwich winding structure, the influence of demagnetization However, it can be understood that the temperature rise ΔT exceeds 50 ° C. except for the comparative example p.
[0050]
On the other hand, in Comparative Example p, it is possible to suppress the influence of the demagnetization, and at the same time, the temperature rise ΔT can be suppressed to within 50 ° C. However, since the Mn—Zn ferrite core is used, ΔB is nanoscaled due to the saturation magnetic flux density limitation The crystal core should be selected to be approximately ½ compared to the case of using the magnetic soft magnetic alloy ribbon core I to H and Fe-based amorphous soft magnetic alloy ribbon core. Since the volume is doubled, it can be seen that there is a problem that the volume of the high-frequency power transformer 20 is also doubled.
[0051]
As a result of detailed examination based on the results obtained in this example, a primary winding without a center tap and a secondary winding with a center tap were provided in an uncut nanocrystalline soft magnetic alloy ribbon core. In the high-frequency power transformer 20, if the value of the leakage inductance at 50 kHz measured at the primary winding end with all of the secondary winding ends short-circuited is 0.3 μH or less, it is practically affected by the demagnetization. It has been found that there is no problem in the operation of the full-bridge type DC-DC converter having the circuit configuration and the specifications shown in Table 1.
[0052]
In comparison with the present inventions A to G, in the case of the present inventions A to D using a nanocrystalline soft magnetic alloy ribbon core having a relative magnetic permeability μr (50 kHz) exceeding 20,000 at 50 kHz, the temperature rise Since ΔT can be less than 45 ° C. with a margin of 10% or more with respect to 50 ° C., which is the practical upper limit, it can be seen that a high-reliability and high-performance high-frequency power transformer and power converter can be realized.
[0053]
(Example 2)
Examine the performance of the high-frequency power transformer 20 of the full-bridge type DC-DC converter in which the switching frequency f is changed from 50 kHz to 20 kHz, and the same full-bridge type DC-DC converter with the same circuit configuration and specifications as in the first embodiment. did.
[0054]
Table 5 shows the winding specifications of the high-frequency power transformer 20. Except for Comparative Examples r, s, t, and u, the winding specifications shown in Table 3 using the magnetic core described in Table 2 of Example 1 were used.
Further, in Table 5, the configurations of the sandwich winding and the uniform winding described for Comparative Example r to Comparative Example u are the same as those in FIGS. 1 and 2 in Example 1.
[0055]
[Table 5]

[0056]
Table 6 shows the DC magnetic characteristics of the magnetic cores used in Comparative Examples r, s, t, and u and the relative permeability μr (50 kHz) at 50 kHz. The magnetic cores in Table 6 are uncut Mn-Zn ferrite magnetic cores, and the magnetic cores are Mn-Zn ferrite magnetic cores which are cut and inserted with a nonmagnetic insulator gap.
Further, as shown in Table 6, the magnetic core core and the magnetic core core have a saturation magnetic flux density Bs at 120 ° C. which is only about ¼ of the magnetic core I shown in Table 2 above. Were stacked in a plastic insulating case having an outer diameter of 46.5 mm, an inner diameter of 21.5 mm, and a height of 84 mm.
[0057]
[Table 6]

[0058]
Table 24 shows the leakage inductance measured by the primary winding with all secondary winding ends short-circuited for the 24 types of high-frequency power transformers 20 shown in Table 5. The circuit configuration of the high-frequency power transformer 20 is shown in FIG. The degree of magnetization when mounted on the high-frequency power transformer 20 of the full bridge type DC-DC converter shown in Table 1, the amount of change ΔB in magnetic flux density during operation, and the temperature rise ΔT of the high-frequency power transformer 20 are shown.
[0059]
As for the demagnetization, within the specification range of Table 1, the case where there is no abnormality due to the demagnetization even if the input voltage and the load change suddenly, the case where the abnormality due to the demagnetization occurs only when the input voltage and the load change suddenly Δ, and the case where abnormality due to the bias magnetism occurs even in the steady operation is marked with ×.
In addition, the amount of change ΔB in magnetic flux density during operation and the temperature rise ΔT were saturated by continuous energization under an input / output condition of an input voltage of 260 V, an output voltage of 40 V, and a load current of 30 A at an ambient temperature of 25 ° C. It is the result measured at the time.
[0060]
[Table 7]

[0061]
As can be seen from Table 7, according to the high-frequency power transformer 20 of the present invention A to G using the non-cut nanocrystalline soft magnetic alloy ribbon core having a low leakage inductance of the sandwich winding structure, the magnetic demagnetization is a practical problem. The temperature rise ΔT can be suppressed to 50 ° C. or less, which is practically satisfactory.
[0062]
On the other hand, Comparative Examples a to g using an uncut nanocrystalline soft magnetic alloy ribbon core having a large leakage inductance, Comparative Example i using an uncut Fe-based amorphous soft magnetic alloy ribbon core and an uncut In the high-frequency power transformer of Comparative Example r using the Mn—Zn ferrite magnetic core, the maximum output could not be stably taken out due to the influence of the demagnetization, and the temperature rise ΔT could not be measured.
[0063]
Further, a nanocrystalline soft magnetic alloy ribbon core having a gap, a Fe-based amorphous soft magnetic alloy ribbon core having a partially uncut crystalline structure, and a Mn-Zn ferrite magnetic core using the gap. The high frequency power transformers 20 of comparative examples h, j, and s having a large leakage inductance using can suppress the influence of magnetic bias, but may cause a problem because the temperature rise ΔT exceeds 50 ° C. that is practically required. Recognize.
[0064]
In addition, nanocrystalline soft magnetic alloy ribbon cores with gaps, uncut Fe-based amorphous soft magnetic alloy ribbon cores, and uncut partially Fe-based amorphous soft magnetic alloy thin films High frequency power of comparative examples m, n, o, t, and u using a wound magnetic core, an uncut Mn-Zn ferrite core and a Mn-Zn ferrite core using a gap, and having a low leakage inductance as a sandwich winding structure In the transformer, it is possible to suppress the influence of the demagnetization, but it can be seen that the temperature rise ΔT exceeds 50 ° C. except for the comparative examples t and u.
[0065]
On the other hand, in Comparative Examples t and u, the influence of the demagnetization can be suppressed to a practically sufficient level, and at the same time, the temperature rise ΔT can be suppressed within 50 ° C. However, since Mn-Zn ferrite magnetic core is used, ΔB is 0.38T due to the saturation magnetic flux density restriction at 120 ° C, and the nanocrystalline soft magnetic alloy ribbon magnetic core A to H and Fe based amorphous soft magnetic alloy Since it is necessary to select 1/4 or less of the case where the ribbon magnetic core is used, the volume of the high frequency power transformer 20 is increased to about 4 times or more since the volume of the magnetic core is quadrupled as described above. It turns out that there is a problem of becoming.
[0066]
As a result of detailed examination based on the results obtained in this example, a primary winding without a center tap and a secondary winding with a center tap were provided in an uncut nanocrystalline soft magnetic alloy ribbon core. In the high-frequency power transformer 20, if the value of the leakage inductance at 50 kHz measured at the primary winding end with all the secondary winding ends short-circuited is 0.3 μH or less, it is practically affected by the bias magnetism. It has been found that there is no problem in the operation of the full-bridge type DC-DC converter with the circuit configuration of Table 1 and the drive frequency of 20 kHz specified in Table 1.
[0067]
In comparison with the present inventions A to G, in the case of the present inventions A to D using a nanocrystalline soft magnetic alloy ribbon core having a relative magnetic permeability μr (50 kHz) exceeding 20,000 at 50 kHz, the temperature rise Since ΔT can be set to less than 40 ° C. with a margin of 20% or more with respect to 50 ° C., which is a practical upper limit, a highly reliable high-performance high-frequency power transformer 20 and a power conversion device using the same are realized. I understand that I can do it.
[0068]
Further, when the squareness ratio Br / Bs, which is the ratio of the residual magnetic flux density Br and the saturation magnetic flux density Bs in the DC magnetic characteristics as in the present invention C, D and E, is 0.2 or less, the magnetic flux density during operation is It was confirmed that there is no problem in the stable operation of the DC-DC converter due to the influence of the magnetic bias even if the change amount ΔB is increased to 1.86 T, which is about 1.1 times 1.69 T shown in Table 7.
[0069]
As a result, the number of turns of the primary winding 21 of the high-frequency power transformer 20 can be reduced by one turn. In this embodiment, however, the primary winding and the secondary winding are related to the relationship between the input voltage and the output voltage. It is necessary to set the turns ratio of 4: 1, and changing the number of turns is not a good idea.
Therefore, while keeping the winding specifications the same, ΔB is increased to 1.86T by reducing the cross-sectional area of the magnetic cores C, D, and E by 10%, and the operation is confirmed by reducing the volume of the magnetic core and the high-frequency power transformer. As a result, a smaller and lighter high-frequency power transformer 20 having a temperature rise ΔT of less than 50 ° C. was obtained without any problem due to the influence of the demagnetization.
[0070]
(Example 3)
A high-frequency power transformer having the same magnetic core dimensions is mounted on the full-bridge type DC-DC converter having the circuit configuration shown in FIG. 3, and the temperature rise of the high-frequency power transformer is 50 ° C. when the input voltage is DC 260 V, the output voltage is 40 V, and the ambient temperature is 25 ° C. The maximum power that can be output was measured while changing the drive frequency from 2 kHz to 200 kHz.
[0071]
Tables 8 to 14 show winding specifications of the high-frequency power transformer 20 used in this example. Table 8 shows the driving frequency 2 kHz, Table 9 5 kHz, Table 10 10 kHz, Table 11 20 kHz, Table 12 50 kHz, Table 13 100 kHz, and Table 14 200 kHz. As the magnetic cores of the high-frequency power transformer 20 in Tables 8 to 14, the magnetic cores I to N in Table 2 and the magnetic cores Y and T having the magnetic characteristics shown in Table 15 were used. The magnetic cores in Table 15 are uncut Mn-Zn ferrite magnetic cores, and the magnetic cores are Mn-Zn ferrite magnetic cores that are cut and inserted with a nonmagnetic insulator gap.
[0072]
Note that the sandwich winding and the uniform winding in the winding structures described in Tables 8 to 14 mean the same winding structure as in the first and second embodiments. Further, as shown in Tables 8 to 14, the number of turns of the winding of the high-frequency power transformer 20 differs depending on the driving frequency because it is determined so that the largest output power can be extracted at each driving frequency.
[0073]
The high-frequency power transformer 20 shown in Tables 8 to 14 is mounted on the full-bridge DC-DC converter having the circuit configuration shown in FIG. 3, and the temperature rise of the high-frequency power transformer with the input voltage DC260V, the output voltage 40V, and the ambient temperature 25 ° C Table 16 shows the results of measuring the maximum power that can be output when the driving frequency is changed from 2 kHz to 200 kHz when the temperature is 50 ° C.
In Table 16, “-” indicates that measurement could not be performed due to the influence of a bias magnetism. In addition, () shows the winding structure of the high-frequency power transformer 20 from which the maximum output power was obtained, and corresponds to Tables 8 to 14.
[0074]
[Table 8]

[0075]
[Table 9]

[0076]
[Table 10]

[0077]
[Table 11]

[0078]
[Table 12]

[0079]
[Table 13]

[0080]
[Table 14]

[0081]
[Table 15]

[0082]
As can be seen from Table 16, according to the present invention, in the driving frequency range of 5 kHz or more and 100 kHz or less, an output power larger than that of the comparative example can be stably achieved with the same magnetic core volume without performing an unstable operation due to the influence of the demagnetization. A high-frequency power transformer 20 that can be supplied is obtained.
[0083]
In addition, in the high frequency power transformer 20 of the present invention shown in Table 16, the leakage inductance at 50 kHz measured at the primary winding end with all the secondary winding ends short-circuited shows a value of 0.3 μH or less. It was.
On the other hand, in the comparative example shown in the table, the leakage inductance measured in the same manner in the high-frequency power transformer 20 having a winding structure with a uniform winding from the magnetic core i which is an uncut nanocrystalline soft magnetic alloy ribbon core to the magnetic core. Therefore, it was also found that these high-frequency power transformers 20 caused unstable operation due to the influence of the demagnetization and could not output stably.
[0084]
From the above, in the high-frequency power transformer 20 in which the primary winding without the center tap and the secondary winding with the center tap are provided on the uncut nanocrystalline soft magnetic alloy ribbon core, If the value of leakage inductance at 50 kHz measured at the primary winding end with all short-circuited is 0.3 μH or less, the output power can be stably taken out without showing unstable operation due to the influence of magnetic bias. It was found that a high-frequency power transformer 20 with high reliability and a high-efficiency and high-reliability full-bridge DC-DC converter using the same can be obtained.
[0085]
[Table 16]

[0086]
Further, in Table 16, the primary core having no center tap from the magnetic core i which is an uncut nanocrystalline soft magnetic alloy ribbon core having a relative permeability μr (50 kHz) at 50 kHz of 20,000 or more. Since the high frequency power transformer 20 of the present invention having a winding and a secondary winding with a center tap has a small magnetic core loss in a high frequency region, it can take out a larger output power at a high frequency exceeding 20 kHz, It can be seen that it is suitable for high frequency specifications.
[0087]
Further, in Table 16, a magnetic core C which is an uncut nanocrystalline soft magnetic alloy ribbon core having a squareness ratio Br / Bs which is a ratio of the residual magnetic flux density Br and the saturation magnetic flux density Bs in DC magnetic characteristics is 0.2 or less. The high frequency power transformer 20 of the present invention having a primary winding without a center tap and a secondary winding with a center tap in the magnetic core takes a large amount of change in magnetic flux density during operation at a frequency of 10 kHz or less. As shown in Table 8, Table 9, and Table 10, the number of turns can be reduced, and it can be seen that a larger output power can be taken out.
[0088]
【The invention's effect】
As described above, according to the present invention, it is possible to suppress the occurrence of magnetic demagnetization that impedes practical use without providing a particularly complicated magnetic demagnetization circuit, and at the same time, a small uncut nanocrystal soft magnetism with low magnetic core loss. A high-efficiency, high-efficiency high-frequency power transformer using an alloy ribbon magnetic core and a small temperature rise, and a high-efficiency and highly reliable power converter using the same can be obtained.
In the above-described embodiments, application examples to a full-bridge type DC-DC converter as a typical power conversion device using a high-frequency power transformer have been described in detail. However, the present invention is centered on a half-bridge type converter. Applicable to all high-frequency power transformers with a primary winding without a tap and at least one set of secondary windings with a center tap, and power converters using the same high-frequency power transformer. The effect is demonstrated, and the effect is extremely large.
[Brief description of the drawings]
FIG. 1 is a conceptual diagram of a cross section of a winding structure of one embodiment of a high-frequency power transformer 20 according to the present invention.
FIG. 2 is a conceptual diagram of a cross section of a winding structure of a high-frequency power transformer as a comparative example.
FIG. 3 is a block diagram of a circuit configuration of a full bridge type DC-DC converter.
4 is a conceptual diagram of a terminal voltage of a primary winding 21 of a high frequency power transformer 10 of the full bridge type DC-DC converter of FIG. 3;
5 is a conceptual diagram of an operation BH loop of the high frequency power transformer 20 of the full bridge type DC-DC converter of FIG. 3 when there is no bias.
6 is a conceptual diagram of an operation BH loop of the high-frequency power transformer 20 of the full-bridge DC-DC converter of FIG.
[Explanation of symbols]
1: Input DC power supply
2, 3, 4, 5: Main switch element
6, 7, 8, 9: Feedback diode
20: High frequency power transformer
21: Primary winding of the high-frequency power transformer 20
23: Secondary winding of the high-frequency power transformer 20
31, 32: Output rectifier diode
33: Output smoothing choke coil
34: Output smoothing capacitor
35, 36: Output terminals
37: Load

Claims (3)

Fine crystal grains mainly composed of Fe and having a crystal grain size of 50 nm or less occupy 50% or more of the entire volume of the structure , the relative permeability at 50 kHz is 20,000 or more, and the residual magnetic flux density Br in the DC magnetic characteristics is The squareness ratio Br / Bs, which is the ratio of the saturation magnetic flux density Bs, is 0.2 or less, a primary winding having no center tap in an uncut nanocrystalline soft magnetic alloy ribbon core, and at least one or more sets of centers In the high-frequency power transformer provided with the tapped secondary winding, at least one of the center tapped secondary windings is configured so that each winding constituting the center winding is substantially equal to the nanocrystalline soft magnetic alloy ribbon magnetic core. to are bifilar wound, the secondary winding of the nanocrystalline soft magnetic alloy ribbon wound magnetic core is sandwiched winding by the primary winding which is wound in a substantially uniform, and the secondary winding end Frequency power transformer values of leakage inductance in 50kHz measured at the primary winding end hands shorted to equal to or less than 0.3MyuH. 2. The high frequency power transformer according to claim 1, wherein the driving frequency is in a range of 5 kHz to 100 kHz. A power converter using the high-frequency power transformer according to claim 1.

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