JP3237322B2 - CN ratio measuring means - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a demodulator for demodulating a received signal in a satellite communication or a mobile satellite communication system.

[0002]

2. Description of the Related Art A conventional CN ratio measurement method includes a receiving wave.
The after direct交検wave, unmodulated portion of the detection output converted A / D, statistically than the A / D converted data sequence C
/ N has been proposed. FIG. 15 shows a conventional C
It is an N ratio measuring instrument, for example, IEICE Technical Report, CS82-73 (198
2), "CN ratio measurement method for TDMA satellite communication" (Mizuno et al.)
It is described in. In FIG. 15, reference numeral 51 denotes an input terminal, and 52 denotes a BP for determining the bandwidth of the CN ratio measurement circuit.
F, 53, 54 are phase detectors for quadrature detection composed of limiters and multipliers, 55 and 56 are LPFs for removing harmonic components, and 57 is an arithmetic unit 1 for squaring the average value of the in-phase detection output. 58 is a computing unit 2 for calculating the mean square value of the quadrature detection output, 59 is a computing unit 3 for squaring the mean value of the quadrature detection output,
Reference numeral 60 denotes an adder for adding the output of the operation unit 2 and the output of the operation unit 3; 61, a tank used for synchronous detection; 62, a delay element used for delay detection; 63, a phase of the output of the tank 61 or the delay element 62; Shifting π / 2 phase shifter, 6
4 is a multiplier for doubling the output of the adder 60, and 65 is an operation unit 1.
A divider 66 divides the output by the output of the adder 60 or the output of the multiplier 64, and 66 is an output terminal for outputting the result of the division.

In general, when calculating C / N from a received signal, the carrier power is obtained by squaring the data at the Nyquist point of the demodulated signal or the average value of the in-phase or quadrature components of the unmodulated portion of the received signal. The noise power can be determined by determining the variance. However, since the noise power of the received signal is the sum of the noise powers of the in-phase and quadrature components, it is necessary to double the variance obtained using the in-phase or quadrature components. The C / N of the received signal is obtained based on the above method.

Next, the operation will be described with reference to FIG. In the following C / N measurement, it is assumed that only the unmodulated portion of the received signal is used, and the case of differential detection is also shown in the figure, but for simplicity, only the case of synchronous detection will be described. Also, let L be the total number of data points used in the following processing, and let in-phase and quadrature components of the quadrature-detected and LPF-passed signals be ei and eq, respectively. Further, let the average of x be E [x]. Received wave y
(t) is passed through the BPF 52, and the C / N of the measurement circuit is changed to the reception C / N.
Setting it higher than N ensures the accuracy of the measurement. TA
The NK 61 reproduces a carrier from the output of the BPF 52 and outputs a reproduced carrier (hereinafter, a reproduced carrier). From the phase detector 1 (53) composed of a limiter and a multiplier,
The in-phase component of the received signal subjected to quadrature detection using the received wave passed through the BPF 52 and the recovered carrier is output, and the phase detector 2
From (54), orthogonal components are similarly output. The arithmetic unit 1 (57) obtains the mean square value (E [ei]) ² of the amplitude of the in-phase component ei of the signal passed through the LPF 1 (55) over L points. From the nature of Gaussian noise,
Since the noise component becomes 0 by calculating the average value of the amplitude,
The carrier power of the received signal can be determined. Arithmetic unit 2
At (58), the mean square value E [e of the amplitude over L points of the quadrature component eq of the signal passed through the LPF 2 (56).
q ² ]. In the operation unit 3 (59), the operation unit 1 (5
Similarly to 7), the mean square value (E [eq]) ² of the amplitude over the L points of the orthogonal component of the signal is obtained. Adder 60
Then, from E [eq ² ] output from the operation unit 2 (58), (E [eq]) ² output from the operation unit 3 (59)
The variance of the orthogonal component is obtained by subtracting. Multiplier 6
In step 4, the variance output from the adder 60 is doubled to obtain the noise power of the received signal. The divider 65 obtains C / N by dividing the carrier power output from the operation unit 1 (57) by the noise power output from the multiplier 64, and outputs the value from the output terminal 66.

[0005]

SUMMARY OF THE INVENTION As described above, the conventional C
In the N-ratio measuring device, since the C / N measurement is performed using the unmodulated part of the received wave, there is a disadvantage that the transmitting side must add the unmodulated part to the transmission data and transmit the data.
Since the carrier power and the noise power are calculated using the in-phase or quadrature components of the received wave, the C / N measurement cannot be performed unless delay detection or synchronous detection is performed as shown in FIG. There was a disadvantage.
Further, in the calculation of the variance, since an average is obtained after accumulating data for one burst and the variance is obtained using the value, there is a disadvantage that the amount of calculation is large and real-time processing cannot be performed.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned drawbacks, and it is possible to remove a modulation component by accumulating and adding a sampled data sequence. By performing N measurement and using the square value of the amplitude of the received wave, the object is to perform C / N measurement even on a signal that has been subjected to quasi-synchronous detection, and to perform C / N measurement before demodulation. I do. Another object is to reduce the amount of calculation.

[0007]

[0008]

[0009]

[0010]

In order to achieve the above object, a CN ratio measuring means according to the first invention converts a phase-modulated received signal and generates an output corresponding to the value. An arithmetic unit, a selector for periodically selecting the output of the arithmetic unit for each different sampling point corresponding to a predetermined sampling period, a cumulative adder for cumulatively adding the signal selected by the selector, and the cumulative adder. A C / N measurement unit is provided for obtaining a power-to-noise power ratio of a received signal using a plurality of cumulative addition values that are respectively cumulatively added for different sample points.

In order to achieve the above object, a CN ratio measuring means according to a second aspect of the present invention converts a phase-modulated received signal to generate an output corresponding to the value, and corresponds to a predetermined sampling period. A selector for periodically selecting the output of the arithmetic unit, a cumulative adder for cumulatively adding the signal selected by the selector, and a power-to-noise power ratio of the received signal using a ratio of the plurality of cumulative added values. A C / N measurement unit to be obtained is provided.
In order to achieve the above object, a CN ratio measuring means according to a third aspect of the present invention converts a phase-modulated received signal to generate an output according to the value, and performs the arithmetic operation in accordance with a predetermined sampling period. A selector for periodically selecting the output of the selector, a cumulative adder for cumulatively adding the signals selected by the selector, and a C / C for obtaining the power-to-noise power ratio of the received signal using the shapes of the plurality of cumulative added values. An N measuring unit is provided. CN ratio measuring means according to the fourth invention to achieve the above object, in the first invention to third invention, the arithmetic unit,
A quadrature detector for extracting the in-phase and quadrature components of the signal from the phase-modulated received signal, an A / D converter for A / D-converting the output of the quadrature detector, and converting the output of the A / D converter according to the value And a conversion output device for generating an output. In order to achieve the above object, the CN ratio measuring means according to the fifth invention provides:
In the first invention to the third invention, the arithmetic unit is an A / D converter for A / D-converting a phase-modulated reception signal, and a quadrature for extracting in-phase and quadrature components of the signal from the output of the A / D converter. A detection unit for converting the output of the quadrature detection unit and generating an output corresponding to the output;

[0012]

[0013]

[0014]

[0015]

According to the first aspect of the present invention, the CN ratio measuring means obtains a plurality of cumulative addition values to obtain noise power and signal power, and measures the CN ratio from the noise power and signal power.

The CN ratio measuring means of the second invention estimates and outputs C / N from a plurality of accumulated values by calculation. No.
CN ratio measuring means of the present invention of 3 estimates the shape using several accumulated value double. Further, C / N is estimated and output by comparing the estimated shape with the reference. The CN ratio measuring means of the fourth invention firstly separates the received wave into an in-phase component (X) and a quadrature component (Y) by quadrature detection, A / D-converts this signal, calculates, and calculates the calculated value. A plurality of cumulative addition values are obtained, and the noise power of the received signal is obtained by using the plurality of cumulative addition values, and the C / N is estimated and output from the cumulative addition value and the noise power by calculation. The CN ratio measuring means of the fifth invention first performs A / D conversion of the received wave, extracts an in-phase component (X) and a quadrature component (Y) from the A / D-converted signal, and then calculates and calculates the calculated value. A plurality of cumulative addition values are obtained, further, noise power of the received signal is obtained using the plurality of cumulative addition values, and C / N is estimated by calculation from the cumulative addition value and the noise power and output.

[0017]

[Embodiment 1] FIG. 1 is a C showing an embodiment of the present invention.
FIG. 2 is a block diagram of the N ratio measuring means, and FIG. 2 is a flowchart showing the operation of FIG. Conventionally, in quadrature detection, a reproduced carrier reproduced on the receiving side is used, but in consideration of the case of quasi-synchronous detection, here, for simplicity, a fixed oscillator of the receiver that is not necessarily synchronized with the carrier on the transmitting side is used. Is also referred to as quadrature detection. In FIG. 1, 1 is an input terminal of a received waveform, 2 is a fixed oscillator for quadrature detection, 3 is a quadrature detector, 4 is an in-phase output of a quadrature detector 3,
A / D converter that samples and quantizes orthogonal components, 5 is a filter that shapes the waveform of the A / D converted signal, 6 is a buffer memory that stores the waveform-shaped signal for one burst in memory, and 7 is a buffer. An XY-R ² converter for calculating the square of the signal amplitude using the output of the memory 6;
Is a selector for selecting the adder to which the output of the XY-R ² converter 7 is to be input, 9 is a cumulative adder composed of N cumulative adders, and 10 is a plurality of output from the cumulative adder 9 , A noise power calculator 11 for calculating the noise power of the received signal using the cumulative addition value of C / N by calculating from the cumulative addition value output from the cumulative addition unit 9 and the noise power output from the noise power calculation unit 10. , A reference numeral 12 denotes an output terminal for outputting the C / N estimated by the C / N calculator 11, a reference numeral 100 denotes a selector 8, a cumulative adder 9, a noise power calculator 10, and C / N C / N composed of N calculation unit 11
It is a measuring unit.

The operation of the CN measuring means will be described below with reference to FIG. Here, for the sake of simplicity, an embodiment will be described for the case of the burst mode. As shown in FIG. 3, the burst length stored in the memory is L symbols, and N samples are performed per symbol. Also,
The symbol period on the transmitting side and the sample period on the receiving side are synchronized.
Further, it is assumed that the initial phase difference from the transmission-side sample point at the time of sampling is 0 (the first sample point is the Nyquist point). Among the A / D-converted quadrature detector outputs, let X (i, j) and n1 (i, j) denote the data and noise of the in-phase component at the j-th sample point of the i-th symbol, respectively. Data and noise are represented by Y (i, j) and n, respectively.
2 (i, j) (1 ≦ i ≦ L, 1 ≦ j ≦ N). These are XY
-R ² of the amplitude of Motoma' signal by passing it through a transducer 2
Assuming that the power value is R (i, j), R (i, j) can be expressed as in Equation 1. R (i, j) = {X (i, j) + n1 (i, j)} ² + {Y (i, j) + n2 (i, j)} ² (Equation 1) The j-th symbol in each symbol If the square value of the signal amplitude is extracted by the selector and the accumulated value accumulated over one burst in the accumulator j is Pj, Pj is expressed by the following equation (2).
It is represented by

[0019]

(Equation 1)

In equation (2), X (i, j) and n1 (i, j), Y
Since (i, j) and n2 (i, j) are independent, the third term is L
Is assumed to be large enough to be zero. Also, n1 (i, j)
, N2 (i, j) are independent for all i, then n1 (i, j) and n2 (i, j) are the in-phase noise at the j-th sample point of the i-th symbol, respectively. Since it is an orthogonal component, the second term is the total noise power at the j-th sample point, but its power value is the same regardless of j. From the above results, Equation 2 can be easily expressed as Equation 3.

[0021]

(Equation 2)

Consider the first term in equation (3). Hereinafter, the case of four-phase shift keying (hereinafter referred to as QPSK) will be described. However, for the sake of simplicity, in quadrature detection, transmission and reception carriers are assumed to be synchronized, there is no frequency difference and no initial phase difference, and intersymbol interference up to k symbols before and after each of the sampled signals is assumed. Assume there is an effect.

FIG. 4 is a diagram showing a state in which the in-phase or quadrature component of the quadrature detection output is set to an initial phase difference = 0 and N samples are performed per symbol. In FIG. 4, the solid line shows the waveform of the received signal having the amplitude a and the symbol period T. Here, an example of the received waveform when a series of data consisting of 1 and -1 is transmitted is shown. Assume that the sample point is shifted from the original Nyquist point by τ (0 ≦ τ ≦ T). The sampled signal y is represented by the sum of the signal component at that time and the intersymbol interference component. If the impulse response of the combined filter for transmission and reception is represented by Expression 4, generally, i
The sampled signal yi represents the transmitted data as d (-
1 or 1) is expressed as in Equation 5. Here, α is the roll-off rate of the filter, and FIG. 5 shows an example of the transfer function of the combined filter for transmission and reception in order to explain α.

[0024]

(Equation 3)

Thus, X (i, j) and Y (i, j) can be expressed as shown in equations (6) and (7).

[0026]

(Equation 4)

From Equations 6 and 7, when the first term in Equation 3 is obtained,

[0028]

(Equation 5)

In the equation 8, di, di (i ≦ 0, L + 1 ≧
Although i) does not exist, here, considering the approximation of the following equation, if di and di (i ≦ 0, L + 1 ≧ i) are PN patterns,

[0030]

(Equation 6)

Here, assuming that the absolute value of the amplitude of the signal at the Nyquist point is 1 (2a ² = 1), that L is sufficiently large, and that the data is random since it is a PN pattern. Since the second term is negligible in comparison with the first term, it can be simply approximated as in Expression 10.

[0032]

(Equation 7)

Therefore, Pj is as follows.

[0034]

(Equation 8)

Consider Cj in equation (11). So far, Cj and Pj have been considered as discrete values,
Since the initial phase difference is not always 0, Cj and Pj are considered as continuous functions C (τ) and P (τ) (−T / 2 ≦ τ ≦ T / 2).
When the burst length L is fixed, C (τ) is uniquely determined by the roll-off rate. For example, C (τ) (− T
/ 2 ≦ τ ≦ T / 2) normalized by C (0) can be expressed as in Expression 12.

[0036]

(Equation 9)

Now, the cumulative addition values at τ = 0 and t1 can be expressed as follows: P (0) = C (0) + N (Equation 13) P (t1) = C (t1) + N (Equation 14) Therefore, it can be seen from Expressions 12 to 14 that C / N can be obtained by estimating the amount corresponding to N.
An example of a method for estimating N will be described below.

In the noise calculation unit 10, as shown in FIG.
(0), subtract a value n0 sufficiently smaller than N from P (t1),
P (t1) / P (0) is compared with a known k (t1), and this operation is repeated until the difference between the two becomes minimum. Assuming that the difference between the two becomes smallest after repeating p times, the noise power N can be expressed as N = p · n0 (Equation 15). In the C / N calculator, the noise calculator 1
First, the signal power C is determined using the noise power N output from 0 and the accumulated value P (0) at the Nyquist point. C (0) = P (0) −p · n0 (Equation 16) Further, C / N is obtained as follows using C and N. C / N = (P (0) −p · n0) / (p · n0) (Equation 17) The C / N obtained by the C / N calculator 11 is output to the output terminal 12.
Output from

In the above examination, it is assumed that there is a point of τ = 0 among the sample points, and C (0) and C (0)
/ N is obtained, but there is not always a point where τ = 0 in an actual operation. In this case, P (τ) is estimated by interpolation or the like from a plurality of Pj values, the point at which P (τ) is maximized is assumed to be a point of τ = 0, and the estimated P
Similar processing may be performed by obtaining a value at the point of τ = t1 in (τ).

Embodiment 2 FIG. FIG. 7 is a block diagram of a CN ratio measuring means showing another embodiment of the present invention, and FIG. 8 is a flowchart showing the operation of FIG. In FIG. 7, an input terminal 1, an A / D converter 4, a filter 5, a memory 6, an XY-R ² converter 7 , C /
The N measuring unit 100 and the output terminal 12 are the same as in the first embodiment. Reference numeral 13 denotes a quadrature converter for extracting an in-phase component (X) and a quadrature component (Y) from a signal A / D converted by an IF (Intermediate Frequency) and stored in a memory; XY-R to find the square of R)
^{It includes a two-} conversion circuit and the C / N measurement unit according to the second invention.

In the first embodiment, A / D conversion is performed on a signal that has been subjected to quasi-synchronous or synchronous detection by quadrature detection before performing C / N. D conversion (hereinafter referred to as IF sampling)
After the / D conversion, quadrature conversion is performed by signal processing such as digital multiplication, and the in-phase component (X) and the quadrature component (Y)
And the same processing as in the first embodiment is performed.

Embodiment 3 FIG. FIG. 9 is a block diagram of a C / N measuring section showing another embodiment of the present invention, and FIG. 10 is a flowchart showing the operation of FIG. In FIG. 9, the selector 8 and the accumulator 9 are the same as those in the first or second embodiment. Reference numeral 14 denotes a C / N for calculating the C / N of the received signal using a plurality of cumulative values output from the cumulative adding section 9.
It is a calculation unit.

In the same manner as in the first or second embodiment, the accumulative adder 9 obtains N accumulative added values. P (0) and P
_Assuming that the ratio of (t1) is P _RATE (t1), P _RATE (t1) becomes as shown in Expression 18 from Expressions 13 and 14.

[0044]

(Equation 10)

From equation 18, X = C / N =-(P _RATE (t1) -1) / (P _RATE (t1) -k (t1)) (Equation 19) Here, as described above, k (t1) Is uniquely determined by the roll-off rate α, so that C / N is uniquely determined by P _RATE (t1). Thus, the C / N calculator 1
In step 4, C / N is estimated.

In the third embodiment, similarly to the first or second embodiment, when there is no point of τ = 0 among the sample points, P (τ) is obtained from a plurality of Pj values by interpolation or the like. )
Is estimated, and the values of P (0) and P (t1) may be obtained.

Embodiment 4 FIG. FIG. 11 is a block diagram of a C / N measuring section showing an embodiment of the present invention, and FIG. 12 is a flowchart showing the operation of FIG. In FIG. 11, the selector 8 and the accumulator 9 are the same as those in the first or second embodiment. Reference numeral 15 denotes a shape estimating circuit for estimating the shape of P using a plurality of cumulative addition values output from the accumulating unit 9, and 16 denotes C / C by comparing the shape estimated by the shape estimating circuit 15 with a reference. N is a comparator for estimating N.

When the roll-off rate is fixed as determined in the third embodiment, P _RATE (t1) is uniquely determined by C / N. P when C / N is changed
FIG. 13 shows _RATE (τ) (−T / 2 ≦ τ ≦ T / 2). Here, the C / N is estimated by comparing several types of P _RATE (τ) obtained in advance and the shape of P composed of P1 to PN obtained from a plurality of cumulative addition values. The method is described below. First, in the same manner as in the first embodiment, N cumulative addition values are obtained by the cumulative addition unit 9. The shape estimating circuit 15 takes out a plurality of cumulative addition values from the N cumulative addition values, and uses them to estimate the shape of P composed of P1 to PN by an interpolation formula or the like as shown in FIG. I do. In the comparator 16, the P estimated by the shape estimation circuit 15 is obtained.
Is compared with the shapes of several types of P _RATE (τ) obtained in advance, and when the shapes match or the two are closest, the C / N of the P _RATE (τ) is calculated as the C / N of the received signal. Judge as N and output.

Other Embodiments 1 In the first to fourth embodiments, the C / N is obtained after the received signal is temporarily stored in the buffer memory for each burst, but the larger the number of symbols to be cumulatively added, the better. Since the noise is averaged and the C / N estimation accuracy is improved, in a system where delay is allowed, the number of bursts to be processed may be increased, for example, by measuring the C / N every two bursts. This processing may be performed in real time without storing the received signal in the buffer memory.

Other Embodiment 2 In the embodiment 1, in the quadrature detection, the carrier wave for transmission and reception is assumed to be synchronized, and there is no frequency difference and initial phase difference. Since the calculation is performed using the square value of the amplitude, a similar result can be obtained even in a quasi-synchronous state.

Third Embodiment In the first to fourth embodiments, the calculation is performed using the square value of the amplitude of the received signal. However, for example, the absolute value of the amplitude of the received signal may be used.

Other Embodiment 4 In the first embodiment, a value n0 sufficiently smaller than N is repeatedly subtracted from P (0) and P (t1). | P _RATE (t1) -k (t1) | In the case of >> 0 (Equation 20), the value of n0 is increased, and in the case of | P _RATE (t1) -k (t1) | ≒ 0 (Equation 21), the value of n0 is decreased. By using the adaptive processing, the estimation can be made faster and more accurately.

Fifth Embodiment In the first and second embodiments, the waveform shaping filter is provided after the A / D converter.
It may be replaced with any place where the operation can be performed (for example, IF band). However, in this case, needless to say, a filter for removing harmonics generated by the quadrature detector is required. Also, note that when processing in real time
No ri is required.

[0054]

[0055]

[0056]

[0057]

As described above, according to the first aspect of the invention, since the method of accumulating the received signals is employed, the C / N measurement can be performed even if there is no unmodulated part, as compared with the conventional CN ratio measuring device. C / N measurement can be performed in real time before demodulation. Furthermore, there is no need to perform operations such as dispersion, the amount of operations can be reduced, and the circuit configuration can be simplified as compared with the related art.

[0058] As described above, according to the second invention, the first
The same effect as that of the invention can be obtained. As described above, according to the third invention, the same effect as that of the first invention can be obtained, and furthermore, the C / N can be measured with higher accuracy by comparing with the reference after interpolation. . As described above, according to the fourth aspect , the same effects as those of the first to third aspects can be obtained. As mentioned above,
According to the fifth aspect , since the measurement can be performed from the IF reception signal, it can also be applied to a demodulator that demodulates from the IF sampling data. In this case, there is no need to create a baseband signal. Therefore, the circuit configuration can be simplified as compared with the related art, and the same effects as those of the first to third aspects can be obtained.

[Brief description of the drawings]

FIG. 1 is a block diagram of a CN ratio measuring device showing a first embodiment of the present invention.

FIG. 2 is a flowchart showing the operation of FIG.

FIG. 3 is a diagram illustrating a state of a burst and a sample.

FIG. 4 is a diagram for explaining an operation of performing N samples of a received signal per symbol.

FIG. 5 is a diagram illustrating a roll-off rate α of a filter.

FIG. 6 is a diagram showing a shape of P (τ).

FIG. 7 is a block diagram of a CN ratio measuring device showing a second embodiment of the present invention.

FIG. 8 is a flowchart showing the operation of FIG. 7;

FIG. 9 is a block diagram of a C / N measuring unit according to a third embodiment of the present invention.

FIG. 10 is a flowchart showing the operation of FIG. 9;

FIG. 11 is a block diagram of a C / N measuring unit according to a fourth embodiment of the present invention.

FIG. 12 is a flowchart showing the operation of FIG. 11;

FIG. 13 is a diagram showing an example of P _RATE (τ).

FIG. 14 is a view for explaining a method for estimating the shape of P;

FIG. 15 is a block diagram of a conventional CN ratio measuring device.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 Input terminal 2 Fixed oscillator 3 Quadrature detector 4 A / D converter 5 Filter 6 Buffer memory 7 XY-R ² converter 8 Selector 9 Cumulative addition unit 10 Noise power calculation unit 11 C / N calculation unit 12 Output terminal 13 Quadrature Converter 14 C / N calculation unit 15 Shape estimation circuit 16 Comparator 51 Input terminal 52 BPF 53 Phase detection unit 1 54 Phase detection unit 2 55 LPF1 56 LPF2 57 Operation unit 1 58 Operation unit 2 59 Operation unit 3 60 Adder 61 TANK 62 delay element 63 π / 2 phase shifter 64 multiplier 65 divider 66 output terminal 100 C / N measurement unit

Claims (5)

(57) [Claims] 1. An arithmetic unit for converting a phase-modulated received signal to generate an output corresponding to the value, and a selection unit for periodically selecting an output of the arithmetic unit for each of different sample points corresponding to a predetermined sampling period. , A cumulative adder for cumulatively adding the signal selected by the selector, and a power-to-noise power ratio of the received signal using a plurality of cumulatively added values each of which is cumulatively added for each different sample point by the cumulative adder. Seeking C
/ N ratio measuring means, comprising a / N measuring unit. 2. An arithmetic unit for converting a phase-modulated received signal to generate an output according to the value, a selector for periodically selecting an output of the arithmetic unit in response to a predetermined sampling period, and the selector Ratio measuring means, comprising: a cumulative adder for cumulatively adding a signal selected according to (1) and a C / N measuring unit for determining a power-to-noise power ratio of a received signal using a ratio of a plurality of cumulatively added values. . 3. An arithmetic unit for converting a phase-modulated received signal to generate an output according to the value, a selector for periodically selecting an output of the arithmetic unit in response to a predetermined sampling period, and the selector Ratio measuring means characterized by comprising: a cumulative adder for cumulatively adding a signal selected according to (1), and a C / N measuring unit for determining a power-to-noise power ratio of a received signal using a shape of a plurality of cumulative added values. . 4. The quadrature detector for extracting in-phase and quadrature components of a signal from a phase-modulated received signal, an A / D converter for A / D converting an output of the quadrature detector,
/ Converts D converter output claims 1 to characterized in that it has a converter output circuit causing an output corresponding to the value
3. The CN ratio measuring means according to any one of 3 . 5. The arithmetic unit comprises: an A / D converter for A / D converting a phase-modulated received signal; a quadrature detection means for extracting in-phase and quadrature components of a signal from an output of the A / D converter; CN ratio measuring means according to any one of claims 1 to 3 characterized by having a conversion output unit causing an output corresponding to the value to convert the detection means output.