JP2016092721A - Signal processor and signal processing method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To remove noise from a DSB signal, accurately.SOLUTION: A signal processor, for receiving a double sideband wave signal and removing noise superimposed only on one sideband wave on an RF bandwidth, includes demodulation means for orthogonal demodulation of the double sideband signal into a baseband signal having a positive frequency bandwidth and a negative frequency bandwidth, and noise elimination means for eliminating a noise component included in an in-phase component outputted from the demodulation means, based on a quadrature component outputted from the demodulation means. The signal processor also includes quadrature component bandwidth extraction means for extracting a noise component of one frequency bandwidth within the positive frequency bandwidth and the negative frequency bandwidth on a quadrature component outputted by the demodulation means. The noise elimination means includes phase rotation means for rotating a noise component, extracted by the quadrature component bandwidth extraction means, by -π/2 or π/2, and synthesis means for removing noise, superimposed on a signal component, based on a phase rotation noise component outputted by the phase rotation means, from a baseband signal component which is an in-phase component of output from the demodulation means.SELECTED DRAWING: Figure 1

Description

  The present invention relates to signal processing in which noise is removed during demodulation in an environment where noise is superimposed only on one of the upper band band and the lower band band in broadcasting and communication using both side bands. The present invention relates to an apparatus and a signal processing method.

  As a method for demodulating a desired broadcast wave by removing the noise from the desired broadcast wave on which the noise is superimposed, techniques described in the following patent documents are known. The following Patent Document 1 discloses a first antenna that receives a broadcast wave stronger than the other antenna and obtains a first signal, and a second antenna that receives noise stronger than the other antenna and obtains a second signal. Is used to adjust the amplitude and phase of the second signal so that the signal level after synthesis is reduced, and synthesizes the first signal. In this technique, the amplitude and phase of the second signal are adjusted when the reception level of the desired broadcast wave is smaller than a certain threshold value. That is, when the noise power is larger than the power of the broadcast wave, the amplitude and phase of the second signal are adjusted so that the level of the combined signal becomes small. Since both antennas receive noise from the same noise source, the amplitude and phase of the noise received by both antennas differ according to the difference in distance from each antenna to the noise source. In order to compensate for this, the amplitude of the second signal is made to coincide with the amplitude of the first signal, the phase of the second signal is changed by π with respect to the phase of the first signal, and The second signal is synthesized in reverse phase. By adjusting the amplification factor and phase of the second signal in this way, a signal with noise canceled from the received broadcast wave can be obtained even when a desired broadcast wave can be received.

  Further, in the technique of Patent Document 2 below, in receiving an AM radio broadcast wave by a radio receiver mounted on a vehicle, pulsed noise emitted from an electronic device of the vehicle is mixed in the AM radio broadcast wave. This is a technology for removing sexual noise. In this technique, first, pulse noise in a band other than the AM radio broadcast wave band is detected, and the noise source is specified by obtaining the magnitude of the period of the pulse noise and the fluctuation range of the level. Depending on the noise source, in the broadcast wave on which pulse noise is superimposed, the harmonic band of the pulse noise near the broadcast wave frequency and the band of the noise period are attenuated for the time corresponding to the time width of the pulse noise. By doing so, the person who listens to AM radio broadcasting is prevented from giving discomfort due to pulse noise.

JP2012-257155A Patent No. 5012246

  The technique of Patent Document 1 is a technique that uses two antennas so that only noise can be received and is adjusted in advance so that the level of the combined signal of the received signals of the two antennas becomes small. . For this reason, the technique of Patent Document 1 requires two antennas to cancel noise, and the setting for removing noise does not affect the adjustment of noise removal by the reception level of a desired broadcast wave. Thus, it is necessary to perform in an environment where the reception level is small. Further, since the amplitude and phase of the second signal are adjusted uniformly regardless of the frequency characteristics of the noise, the noise is not completely removed. Further, in the case of a DC-DC converter using the PWM method, the frequency characteristic of noise changes due to a change in pulse width even if the basic period does not change. For this reason, the method of Patent Document 1 cannot completely remove noise.

  The technique of Patent Document 2 detects pulse noise in a band other than the broadcast wave band, and attenuates an appropriate frequency band according to the type of noise for the time corresponding to the pulse width at the detection timing. Technology. Therefore, the broadcast wave is essentially attenuated only during the period of pulse noise. This causes discomfort to those who listen to AM radio broadcasts.

  Therefore, an object of the present invention is to accurately remove this noise during demodulation in an environment where noise is superimposed only on one of the upper band band or the lower band band in broadcasting and communication using both side band waves. Is to do.

According to a first aspect of the present invention for solving the above problems, a double sideband signal is orthogonally demodulated in a signal processing apparatus that receives a double sideband signal and removes noise superimposed on only one sideband in the RF band. Then, based on the quadrature component output from the demodulating unit that demodulates the baseband signal having the positive frequency band and the negative frequency band, the noise component included in the in-phase component output from the demodulating unit is removed. A signal processing apparatus having a noise removing unit.
The gist of the present invention is to remove the noise component superimposed on the in-phase component from the quadrature component, paying attention to the fact that the quadrature component does not include the signal component and only the noise component appears. The noise component can be removed on the time axis, the frequency axis, or a combination of both.

  Further, in the present invention, there is an orthogonal component band extracting means for extracting one band noise component of one of the positive frequency band and the negative frequency band among the orthogonal components output from the demodulating means, and the noise removing means The phase rotation unit that rotates the phase of the one-band noise component extracted by the quadrature component band extraction unit by −π / 2 or π / 2, and the phase rotation that the phase rotation unit outputs to the in-phase component output by the demodulation unit It is possible to provide a signal processing apparatus having a synthesis unit that synthesizes noise components and removes noise components included in the in-phase components. Note that the sign of the phase is defined as positive in the sign of the phase in the complex coordinate system, that is, the left rotation direction regardless of the positive frequency band and the negative frequency band.

  The present invention removes noise from a demodulated signal using the following principle. When quadrature demodulation is performed on both sideband waves (for example, AM broadcast waves) that are not orthogonally multiplexed, a signal component and a noise component appear in the in-phase component of the baseband, no signal component appears in the orthogonal component, and only the noise component Appears.

1. When noise is superimposed only on the upper sideband and the noise component in the positive frequency band of the orthogonal component is extracted.
Regarding the noise component, since the in-phase component and the quadrature component after quadrature demodulation are real functions, the spectrum of the noise component in the baseband is in a complex conjugate relationship between the positive frequency band and the negative frequency band. That is, for the noise component of the in-phase component, the amplitude A (ω) and the phase φ _r (ω) at the frequency ω of the spectrum are the amplitude A (−ω) and the phase {−φ _r (− ω)}. Further, regarding the spectrum of the noise component of the quadrature component, the amplitude is equal to the amplitude of the noise component of the in-phase component, and the amplitude A (ω) and the phase φ _i (ω) at the frequency ω are the amplitude A at the frequency (−ω). Equal to (−ω) and phase {−φ _i (−ω)}. When noise is superimposed only on the upper side band in the RF band, the noise component of the quadrature component is the same as the noise component of the in-phase component in both the positive frequency band and the negative frequency band on the time axis. The phase is delayed by π / 2. That is, the following equation is established.

φ _i (ω) = φ _r (ω) −π / 2,
−φ _i (−ω) = − φ _r (−ω) −π / 2,
φ _i (−ω) = φ _r (−ω) + π / 2 = −φ _r (ω) + π / 2
However, the phase φ is positive in the left rotation direction in the complex coordinate system. It is defined that the wave vector rotates in the positive direction in the positive frequency region, and the wave vector rotates in the negative direction in the negative frequency band.

This relationship is established for all positive frequency band frequencies. Therefore, if the time waveform (complex function) f _i ⁺ (t) of the noise component A (ω) exp (j φ _i (ω)) of the orthogonal component in the positive frequency band is known, the phase of each frequency component of the noise component is obtained. Is subtracted from the in-phase component of the positive frequency band (signal component + noise component), the time of the noise component A (ω) exp (j φ _r (ω)) of the in-phase component The waveform (complex function) f _r ⁺ (t) can be removed. Specifically, the signal component F of the in-phase component can be obtained by subtracting jf _i ⁺ (t) from the in-phase component of the positive frequency band (that is, the signal component F ⁺ (t) + the noise component f _r ⁺ (t)). ⁺ (T) is obtained. In the double sideband wave, the signal can be demodulated with only one band component, and therefore the signal can be demodulated by the above processing by the real part of F ⁺ (t).

2. When noise is superimposed only on the upper sideband and noise components in the negative frequency band of the orthogonal component are extracted.
Extraction of the noise component of the orthogonal component may be performed for the negative frequency band. That is, if the time waveform (complex function) f _i ⁻ (t) of the noise component A (−ω) exp (j φ _i (−ω)) of the orthogonal component in the negative frequency band is known, each frequency component of the noise component Is subtracted from the in-phase component (signal component + noise component) of the negative frequency band by subtracting the time waveform obtained by advancing the phase of π / 2 from the in-phase component noise component A (−ω) exp (j φ _r (−ω )) of the time waveform (complex function) f _r ^- a (t) can be removed. Specifically, by subtracting −jf _i ⁻ (t) from the in-phase component in the negative frequency band (that is, signal component F ⁻ (t) + noise component f ⁻ (t)), the signal component F of the in-phase component is obtained. ^- (T) is obtained. As described above, the extraction of the noise component of the orthogonal component can be performed in either the positive frequency band or the negative frequency band.

3. When noise is superimposed on the upper band band, when correcting the in-phase component in a band other than the extracted quadrature component band Extract the negative frequency band of the quadrature noise component and extract the positive frequency of the in-phase component You may make it correct | amend the signal of a zone | band. Since there is a relationship of φ _r (ω) = − {φ _i (−ω) −π / 2}, the noise component A (−ω) exp (j φ _i (−ω)) of the orthogonal component in the negative frequency band time waveform f _i ^- the complex conjugate of the phase of [pi / 2 by advancing the time waveform of each frequency component of the (t), from the in-phase component of the positive frequency bands, is subtracted, the time waveform of the noise component of the in-phase component (complex The function f _r ⁺ (t) can be removed. Specifically, if {−jf _i ⁻ (t)} ^* is subtracted from the in-phase component in the positive frequency band (that is, signal component F ⁺ (t) + noise component f ⁺ (t)), the in-phase component A signal component F ⁺ (t) is obtained. However, * represents a complex conjugate operation.

Conversely, the positive frequency band of the noise component of the quadrature component may be extracted to correct the signal of the negative frequency band of the in-phase component. Since there is a relationship φ _r (−ω) = φ _i (−ω) −π / 2 = − {φ _i (ω) + π / 2}, the noise component A (ω) exp ( j φ _i (ω)) of the time waveform f _i ⁺ (t), the complex conjugate of the time waveform obtained by advancing the phase of each frequency component by π / 2 from the in-phase component (signal component + noise component) of the negative frequency band By subtracting, the time waveform (complex function) f _r ⁻ (t) of the noise component A (−ω) exp (j φ _r (−ω)) of the in-phase component can be removed. Specifically, by subtracting {jf _i ⁺ (t)} ^* from the in-phase component in the negative frequency band, the signal component F ⁻ (t) of the in-phase component is obtained.
As described above, the extraction of the noise component of the quadrature component is performed in either the positive frequency band or the negative frequency band, and the quadrature component in which the phase of π / 2 is advanced to the in-phase component of the same code band, Alternatively, the noise superimposed on the in-phase component can be removed by synthesizing the complex conjugate of the quadrature component with the π / 2 phase advanced to the in-phase component of the band of the different sign.

4). When noise is superimposed only on the lower sideband and the noise component in the positive frequency band of the orthogonal component is extracted.
Compared with the case where noise is superimposed only on the upper side band, the relationship between the positive frequency band and the negative frequency band is only reversed. Therefore, the following equation is established by replacing ω in the above equation with -ω.

−φ _i (−ω) = − φ _r (−ω) + π / 2,
φ _i (ω) = φ _r (ω) + π / 2,
φ _i (ω) = φ _r (ω) + π / 2 = −φ _r (−ω) + π / 2
When noise is superimposed only on the lower sideband in the RF band, the noise component of the quadrature component is the same as the noise component of the in-phase component. The phase is advanced by π / 2. Therefore, compared with the case where noise is superimposed on the upper sideband, the factors (j) and (−j) to be applied to the extracted orthogonal component are reversed, and similarly, the noise superimposed on the in-phase component can be removed. .

If the time waveform f _i ⁺ (t) of the noise component of the orthogonal component in the positive frequency band is known, the time waveform obtained by delaying the phase of each frequency component of the noise component by π / 2 is calculated from the in-phase component of the positive frequency band. By subtracting, the time waveform (complex function) f _r ⁺ (t) of the noise component of the in-phase component can be removed. Specifically, by subtracting −jf _i ⁺ (t) from the in-phase component in the positive frequency band, the signal component F ⁺ (t) of the in-phase component is obtained.

5. When noise is superimposed only in the lower sideband, and the noise component in the negative frequency band of the orthogonal component is extracted.
Extraction of the noise component of the orthogonal component may be performed for the negative frequency band. Also in this case, since the noise component of the quadrature component is advanced by π / 2 on the time axis with respect to the noise component of the in-phase component, compared to the case where noise is superimposed on the upper sideband, Similarly, the noise superimposed on the in-phase component can be removed only by reversing the factors (j) and (−j) to be applied to the extracted orthogonal component. If the time waveform (complex function) f _i ⁻ (t) of the noise component of the orthogonal component in the negative frequency band is known, the time waveform obtained by delaying the phase of each frequency component of the noise component by π / 2 is obtained. in-phase component, is subtracted, the time waveform (complex function) of the noise component of the phase component f _r ^- a (t) can be removed. Specifically, by subtracting jf _i ⁻ (t) from the in-phase component in the negative frequency band, the signal component F ⁻ (t) of the in-phase component is obtained. As described above, the extraction of the noise component of the orthogonal component can be performed in either the positive frequency band or the negative frequency band.

6). When noise is superimposed on the lower sideband, when correcting the in-phase component in a band other than the extracted quadrature component, the noise component of the quadrature component is corrected in the same way as when noise is superimposed on the upper sideband. The negative frequency band may be extracted to correct the positive frequency band signal of the in-phase component. Since there is a relationship φ _r (ω) = − {φ _i (−ω) + π / 2}, each frequency component of the time waveform (complex function) f _i ⁻ (t) of the noise component of the orthogonal component in the negative frequency band By subtracting the complex conjugate of the time waveform delayed by π / 2 from the in-phase component of the positive frequency band, the time waveform (complex function) f _r ⁺ (t) of the noise component of the in-phase component can be removed. Specifically, by subtracting {jf _i ⁻ (t)} ^* from the in-phase component in the positive frequency band, the signal component F ⁺ (t) of the in-phase component is obtained.

Conversely, the positive frequency band of the noise component of the quadrature component may be extracted to correct the signal of the negative frequency band of the in-phase component. Since there is a relationship of φ _r (−ω) = − φ _i (ω) + π / 2, the phase of each frequency component of the time waveform f _i ⁺ (t) of the noise component of the orthogonal component in the positive frequency band is represented by π / 2. the complex conjugate of only delayed the time waveform, the phase component of the negative frequency band, is subtracted, the time waveform f _r of the noise component of the in-phase component ^- a (t) can be removed. Specifically, by subtracting {−jf _i ⁺ (t)} ^* from the in-phase component in the negative frequency band, the signal component F ⁻ (t) of the in-phase component is obtained.
As described above, the extraction of the noise component of the quadrature component is performed in either the positive frequency band or the negative frequency band, and the quadrature component delayed by the phase of π / 2 is changed to the in-phase component of the band of the same code, or By combining the complex conjugate of the quadrature component with the π / 2 phase delayed with the in-phase component of the band of the different sign, the noise superimposed on the in-phase component can be removed.

7). When performing phase rotation processing and synthesis processing in frequency space.
As described below, the phase rotation process and the synthesis process may be performed in a frequency space, and the result may be inversely Fourier transformed to obtain a demodulated signal on the time axis from which noise has been removed.
The spectrum of the orthogonal component can be obtained by Fourier-transforming the noise component in the positive frequency band of the orthogonal component to obtain the spectrum. By multiplying this spectrum by j or -j, the spectrum of the in-phase component of the noise component is obtained. The spectrum of the in-phase component can be obtained by Fourier-transforming the in-phase component (signal + noise) in the positive frequency band. If the spectrum of the noise component of the in-phase component is subtracted from the spectrum of the in-phase component and inverse Fourier transform is performed, a signal waveform in the positive frequency band from which noise is removed can be obtained. The same applies to the negative frequency band.
The present invention is based on the above principle.

  In the present invention, it has in-phase component band extraction means for extracting one of the positive frequency band and negative frequency band from the in-phase component output from the demodulation means, and the phase rotation means is extracted by the quadrature component band extraction means. A cross-correlation calculating means for calculating a cross-correlation value between the generated single-band noise component and the single-band signal component extracted by the in-phase component band extracting means, and based on a sign of a value output from the cross-correlation calculating means It is desirable to determine the rotation direction of the phase. This feature is that the result of cross-correlation between the noise component of the in-phase component and the noise component of the quadrature component is j or -j. In removing the noise component of the in-phase component from the detected noise component of the quadrature component, the sign of the phase of π / 2 is determined. As described above, the phase of the noise component of the in-phase component in the positive frequency band is uniquely related to the phase of the quadrature component, the phase of the quadrature component in the negative frequency band, and the phase of the noise component of the in-phase component in the negative frequency band. is there. Therefore, the calculation of the cross correlation value is not necessarily performed between the frequency bands of the same code.

  Also, in the present invention, the cross-correlation calculating means is a product of one and the other complex conjugate of the one-band signal component output from the in-phase component band extracting means and the one-band noise component extracted by the quadrature component band extracting means. The cross-correlation value may be calculated based on the moving time average. It is a specific example of cross-correlation calculation by complex calculation.

  In the present invention, the synthesizing unit may synthesize the phase rotation noise component and the one-band signal component extracted by the in-phase component band extracting unit. That is, noise is removed from either the positive frequency band or the negative frequency band. In this case, an SSB from which noise is removed is obtained.

  Further, in the present invention, the in-phase component band extracting means includes a first in-phase component band extracting means for extracting the one-band signal component as the first one-band signal component, and the other frequency band for the first one-band signal component. And a second in-phase component band extracting means for extracting the second one-band signal component of the first half-band signal component, and the phase rotation means outputs the one-band noise component extracted by the quadrature component band extracting means by −π / 2 or π / 2. A first phase rotation unit for rotating the phase; and a phase inversion unit for calculating a complex conjugate of the signal output from the first phase rotation unit. The combining unit converts the first one-band signal component into the first phase rotation. A first synthesizing unit for removing noise superimposed on the first one-band signal component based on the first phase rotation noise component output from the unit; and a phase inversion for outputting the second one-band signal component from the phase inversion unit. Based on the noise component, the second one-band signal generation And a second combining unit that removes noise superimposed on the minutes.

  The single-band noise component, the in-phase component noise component in the positive frequency band and the noise component in the negative frequency band are uniquely determined by π / 2 phase rotation or complex conjugate (ie, phase inversion) after phase rotation. It is determined. Therefore, in this aspect, in the quadrature component, noise is removed from the signal of one or both bands of the in-phase component based on the one-band noise component extracted from one of the positive frequency band and the negative frequency band. be able to.

  In the present invention, the in-phase component band extracting means extracts the first in-phase component band extracting means for extracting the one-band signal component as the first one-band signal component, and the first one-band signal component in the other frequency band. A second in-phase component band extracting unit that extracts a two-single band signal component, and the quadrature component band extracting unit uses a noise component in a frequency band having the same sign as the first one-band signal component as a first one-band noise component. A first orthogonal component band extracting means for extracting; a second orthogonal component band extracting means for extracting a second single band noise component of the other frequency band with respect to the first single band noise component; Are extracted by the first phase rotation means for rotating the phase of the first one-band noise component extracted by the first orthogonal component band extraction means by −π / 2 or π / 2, and by the second orthogonal component band extraction means. The second one-band noise component is π / 2. Includes a second phase rotation unit that rotates the phase by −π / 2, and the combining unit converts the first one-band signal component based on the first phase rotation noise component output from the first phase rotation unit, First synthesis means for removing noise superimposed on the first one-band signal component, and second second-band signal component based on the second phase rotation noise component output from the second phase rotation means. You may have the 2nd synthetic | combination means which removes the noise superimposed on the signal component. In this case, both noises can be removed from the in-phase components of the positive frequency band and the negative frequency band. Therefore, since it can demodulate as a double sideband wave from which noise was removed, reception sensitivity improves.

  Further, the sign of the rotation phase by the first phase rotation means and the sign of the cross-correlation value determined by one phase rotation means also determines the sign of the phase rotation by the other phase rotation means. May be. That is, the phase rotation code determined by the noise component of the orthogonal component in one frequency band in the baseband is a different sign from the phase rotation code determined by the noise component of the orthogonal component in the other frequency band. Therefore, the phase rotation code can be determined by signal processing in one frequency band.

  The first phase rotation means includes first cross-correlation calculation means as a cross-correlation calculation means for calculating a cross-correlation value from the first one-band noise component and the first one-band signal component. The phase rotation direction is determined based on the first value output from the calculating means, and the second phase rotating means calculates the cross-correlation value between the second one-band noise component and the second one-band signal component. There may be provided a second cross-correlation calculating means as the calculating means, and the phase rotation direction may be determined based on the second value output from the second cross-correlation calculating means. In this case, the rotation code of the phase of the noise component of the orthogonal component is determined from each frequency band.

  In the present invention, the demodulating means includes a phase-locked loop unit for controlling the frequency and phase of the demodulated carrier so that the beat signal of the error frequency of the demodulated carrier contained in the quadrature component after quadrature demodulation becomes zero. It is desirable to have. Synchronous detection can be performed, and noise can be reliably removed.

  In addition, the demodulation means obtains a beat signal of the error frequency of the demodulated carrier wave with respect to the modulated carrier wave from the moving average of the baseband signal, and based on the beat signal, a signal obtained by correcting the fluctuation due to the beat signal of the baseband signal It is desirable to have synchronization means for making a band signal. Since the spectrum in the baseband is shifted in frequency by the beat frequency for both the in-phase component and the quadrature component, the signal component can be demodulated and noise can be reliably removed by correcting this frequency shift.

  Further, the present invention is a signal processing method for receiving a double sideband signal and removing noise superimposed only on one sideband wave in the RF band. The signal processing method is characterized in that a baseband signal having a negative frequency band is demodulated and a noise component included in the demodulated in-phase component is removed based on the demodulated quadrature component. In the method invention, a noise component in one of the positive frequency band and the negative frequency band among the demodulated orthogonal components is further extracted as a one-band noise component, and the extracted noise component is −π / 2 or By rotating the phase by π / 2 and demodulating the baseband signal component, which is the demodulated in-phase component, the noise superimposed on the signal component can be removed based on the phase-rotated noise component. In the present invention, the signal component of one of the positive frequency band and the negative frequency band in the in-phase component is extracted as a single band signal component, and the sign of phase rotation is the single band signal component and the single band noise component. It may be determined based on the cross-correlation value. In short, the present invention is characterized in that noise contained in the in-phase component is eliminated from the quadrature component obtained by quadrature demodulation, although it is not necessary to perform quadrature demodulation for signal demodulation.

  According to the present invention, in an environment where noise is superimposed only on one of the upper sideband band or the lower sideband band, this noise can be accurately removed during demodulation. Can be improved.

The block diagram of the signal processing apparatus which concerns on the specific Example 1 of this invention. FIG. 3 is a frequency characteristic diagram of an input signal and a demodulated signal of the signal processing apparatus according to the first embodiment. FIG. 3 is a frequency characteristic diagram of an in-phase component and a quadrature component in a baseband after demodulation of the signal processing apparatus according to the first embodiment. FIG. 3 is a frequency characteristic diagram of an output signal from which an in-phase component, a quadrature component, and noise are removed in a baseband positive frequency band after demodulation of the signal processing device according to the first embodiment. Explanatory drawing for demonstrating the effect | action of the signal processing apparatus of Example 1. FIG. The block diagram of the signal processing apparatus which concerns on the specific Example 2 of this invention. The block diagram of the signal processing apparatus which concerns on the specific Example 3 of this invention. The block diagram of the signal processing apparatus which concerns on the specific Example 4 of this invention. The block diagram of the signal processing apparatus which concerns on the modification of the specific Example 4 of this invention. The block diagram of the signal processing apparatus which concerns on the specific Example 5 of this invention. The block diagram of the signal processing apparatus which concerns on the specific Example 6 of this invention.

  Hereinafter, the present invention will be described based on specific examples. The present invention is not limited to the following examples.

  A configuration of a signal processing apparatus 1 according to a specific embodiment of the present invention is shown in FIG. A present Example is a signal processing apparatus which suppresses the noise mixed in the AM radio receiver in HV (hybrid vehicle). It is assumed that the HV is equipped with a DC-DC converter that is controlled at a carrier frequency of 100 kHz. The AM radio broadcast wave is assigned a frequency band of 531 kHz to 1602 kHz. The switching noise generated from the DC-DC converter basically becomes a line spectrum string that is an integral multiple of the fundamental frequency of 100 kHz in the frequency space. This noise enters the AM radio broadcast band and gives noise to the AM radio broadcast wave. In the case of such noise, since the noise spectrum has an interval of 100 kHz, there is no noise in both the upper band band and the lower band band of AM radio broadcasting. This embodiment is a signal processing device that cancels this type of noise entering the AM radio broadcast band. However, the present invention is not limited to such noise. In double-sided band transmission that is not orthogonally multiplexed, all of the noise mixed only in one band of the upper sideband and the lower sideband. Can be used in any environment. In addition, as long as the double-sideband transmission has a complex conjugate relationship with each other, the signals in each band may be AM modulation, FM modulation, phase modulation, or other modulation schemes.

  In the signal processing apparatus 1 of the present embodiment, the AM radio broadcast signal received by the antenna 11 is amplified by the amplifier 12, sampled by the A / D converter 13 at a constant period Δt, and converted into a digital value. , A device processed by the CPU. The configuration of FIG. 1 is expressed in blocks for each functional unit of digital processing. The signal output from the A / D converter 13 is a real number, but the orthogonal demodulator 20 and subsequent data processing are all performed in complex numbers. The quadrature demodulating unit 20 serving as a demodulating unit includes a mixer 21, a demodulated carrier wave generating unit 22, an in-phase component extracting unit 23, and a quadrature component extracting unit 24. A baseband signal is obtained by the orthogonal demodulator 20. In relation to handling with complex signals, this baseband has a positive frequency band corresponding to the upper sideband band and a negative frequency band corresponding to the lower sideband band. The filter unit 30 includes a filter 31 that receives an in-phase component baseband signal and extracts a positive frequency band, and a filter 32 that receives a quadrature component baseband signal and extracts a positive frequency band. . The filter 31 constitutes in-phase component band extracting means for extracting one frequency band of the in-phase component, and the filter 32 constitutes quadrature component band extracting means for extracting one frequency band of the quadrature component.

  Further, the phase rotation unit 40 constituting the phase rotation unit includes a multiplier 41 that multiplies the baseband signal in the positive frequency band of the orthogonal component output from the filter 32 by −j or j according to the setting of the changeover switch 42. Have. That is, this multiplication corresponds to rotating the phase by all frequencies, uniformly -π / 2 or π / 2. Then, the synthesizing unit 60 constituting the synthesizing unit adds and synthesizes the baseband signal in the positive frequency band of the in-phase component output from the filter 31 and the output signal (phase rotation noise component) of the phase rotating unit 40. The filters 31 and 32 are filters that extract a positive frequency band, and are parallel circuits of a Hilbert filter and an all-pass delay circuit. That is, this filter obtains I (t) + jQ (t) = {(δ (t) + jh (t)} * I (t) from the real function I (t), where h (t ) Denotes the impulse response of the Hilbert filter, * denotes the convolution integral, and the transfer function of δ (t) + jh (t) is a positive bandpass filter in which 0 is negative when ω is negative and 2 when ω is positive. The noise removing means includes a filter 32, a phase rotation unit 40, and a synthesis unit 60.

In this embodiment, the received signal is directly sampled. However, the demodulator that shifts the band is configured by an analog circuit and temporarily drops to the intermediate frequency band to sample the IF signal. Also good. In addition, orthogonal demodulation is performed by numerical calculation and is performed by extracting a real part and an imaginary part. However, orthogonal demodulation is demodulation using cos (ω _c t) and sin (ω _c t). May be.

この受信信号ｒ（ｔ）のフーリエ変換であるスペクトルは図２（ａ）に示すようになり、上側帯波帯域と下側帯波帯域とを有している。Ｓ^- は下側帯波のスペクトル、Ｓ⁺ は上側帯波のスペクトルであり、Ａは搬送波の振幅、ρは、放送局から受信装置までにおいて、ＲＦ帯域やＩＦ帯域の復調に至る前の伝送路で重畳された雑音のスペクトルである。Ａは実数、Ｓ^- 、Ｓ⁺ 、ρは角周波数ω（以下、単に、「周波数」と記す）に関する複素関数である。ωに関して、Ｓ^- 、Ｓ⁺ の絶対値は等しく、位相は反転関係にある。したがって、Ｓ^- 、Ｓ⁺ は相互に複素共役関数である。Ｓ^- （ｔ）、Ｓ⁺ （ｔ）は、それぞれ、Ｓ^- 、Ｓ⁺ のフーリエ逆変換であり、時間に関する複素関数である。また、Ｓ^- （ｔ）、Ｓ⁺ （ｔ）は、相互に複素共役の関係にあり、したがって、Ｓ^- （ｔ）＋Ｓ⁺ （ｔ）は実関数である。ω_c は、変調時の搬送波の周波数、ω_c ＋ω_n は上側帯波に重畳した雑音の周波数である。 Next, the operation of this apparatus will be described.
It is assumed that noise is superimposed on the upper sideband band from the broadcasting station to the receiving device. The reception signal r (t) output from the antenna 11 is expressed by equation (1).

A spectrum that is a Fourier transform of the received signal r (t) is as shown in FIG. 2A, and has an upper sideband and a lower sideband. S ⁻ is the spectrum of the lower sideband, S ⁺ is the spectrum of the upper sideband, A is the amplitude of the carrier wave, ρ is the transmission path from the broadcasting station to the receiving apparatus before the demodulation of the RF band and IF band It is the spectrum of the noise superimposed by. A is a real number, S ⁻ , S ⁺ , and ρ are complex functions relating to an angular frequency ω (hereinafter simply referred to as “frequency”). With respect to ω, the absolute values of S ⁻ and S ⁺ are equal, and the phases are in an inversion relationship. Therefore, S ⁻ and S ⁺ are mutually complex conjugate functions. S ⁻ (t) and S ⁺ (t) are inverse Fourier transforms of S ⁻ and S ⁺ , respectively, and are complex functions related to time. Further, S ⁻ (t) and S ⁺ (t) are in a complex conjugate relationship with each other, and therefore S ⁻ (t) + S ⁺ (t) is a real function. ω _c is the frequency of the carrier wave during modulation, and ω _c + ω _n is the frequency of the noise superimposed on the upper sideband.

A wave propagating in space is represented by the real part of r (t). Therefore, the sampled received signal (data) output from the A / D converter 13 is a real number sequence. Next, the received signal is demodulated orthogonally. Since there is no orthogonal component of the signal component, in the complex space, the orthogonal demodulation is equivalent to performing an operation of multiplying the received signal of the real part of the complex function expressed by equation (1) by exp (−jω _c t). Therefore, the demodulated baseband signal is expressed by equation (2). Since the demodulation result has a factor of 1/2, x (t) is defined as twice the result of the orthogonal demodulation in order to simplify the expression. That is, the output signal x (t) of the mixer 21 can be expressed by equation (2), and its spectrum is as shown in FIG. 2 (b), which has a baseband positive frequency band and a negative frequency band. Yes. Noise exists only in the positive frequency band.

すなわち、同相成分抽出部２３の出力信号ｘ_r （ｔ）が（３）式で、直交成分抽出部２４の出力信号ｘ_i （ｔ）が（４）式で、表現される。同相成分には信号成分と雑音成分が存在するが、直交成分には、信号成分が存在せず、雑音成分のみが存在する。同相成分（３）式のスペクトルは、図３（ａ）に示すようになる。正周波数帯域には、信号成分のスペクトルＳ⁺ と雑音成分のスペクトル（ρ／２）が現れ、負周波数帯域には、信号成分のスペクトルＳ^- と雑音成分のスペクトル（ρ^* ／２）が現れている。ρ^* はρの複素共役で、ρの位相を反転したスペクトルである。同相成分ｘ_r （ｔ）も、直交成分ｘ_i （ｔ）も実関数である。 The real part of equation (2) is the in-phase component in quadrature demodulation, and the imaginary part is the quadrature component in quadrature demodulation.
The in-phase component is expressed by equation (3), and the quadrature component is expressed by equation (4).

That is, the output signal x _r (t) of the in-phase component extraction unit 23 is expressed by equation (3), and the output signal x _i (t) of the quadrature component extraction unit 24 is expressed by equation (4). The in-phase component has a signal component and a noise component, but the quadrature component has no signal component and only a noise component. The spectrum of the in-phase component (3) is as shown in FIG. In the positive frequency band, the signal component spectrum S ⁺ and the noise component spectrum (ρ / 2) appear, and in the negative frequency band, the signal component spectrum S ⁻ and the noise component spectrum (ρ ^* / 2) appear. ing. ρ ^* is a complex conjugate of ρ and a spectrum obtained by inverting the phase of ρ. Both the in-phase component x _r (t) and the quadrature component x _i (t) are real functions.

The spectrum of the orthogonal component x _i (t) in the equation (4) is as shown in FIG. In the positive frequency band, a spectrum (−jρ / 2) of the noise component of the orthogonal component appears. That is, the noise component has the same amplitude as the noise component of the in-phase component, but the phase is rotated by −π / 2 with respect to the in-phase component (delayed by π / 2 on the time axis). The spectrum (jρ ^* / 2) of the noise component of the orthogonal component appears in the negative frequency band. That is, the noise component has the same amplitude as the noise component of the in-phase component, but the phase is rotated by π / 2 with respect to the in-phase component (delayed by π / 2 on the time axis). Further, in both the in-phase component and the quadrature component, the spectra in the positive frequency band and the negative frequency band are in a complex conjugate relationship, that is, a relationship in which the phases are inverted.

Next, the filter 31 extracts an in-phase component x _r ⁺ (t) (one-band signal component) in the baseband positive frequency band. x _r ⁺ (t) is expressed by equation (5), and its spectrum is as shown in FIG. Further, the filter 32 extracts a noise component x _i ⁺ (t) (one-band noise component) of an orthogonal component in the baseband positive frequency band. x _i ⁺ (t) is expressed by equation (6), and its spectrum is as shown in FIG. x _r ⁺ (t) and x _i ⁺ (t) are complex functions.

Next, the multiplier 41 of the phase rotation unit 40 multiplies the noise component x _i ⁺ (t) output from the filter 32 by −j to output the phase rotation noise component (−jx _i ⁺ (t)). Is done. In this process, the phase of the noise component in the positive frequency band of the quadrature component is delayed by π / 2 (rotated by −π / 2) so that the phase difference from the noise component in the positive frequency band of the in-phase component becomes π. This means that the phase is controlled. That is, the quadrature noise component is controlled to have an inverted waveform of the in-phase noise component. Then, in the synthesizing unit 60, the phase rotation noise component (−jx _i ⁺ (t)) is added to the in-phase component x _r ⁺ (t) in the positive frequency band. The output of the synthesis unit 60 is expressed by equation (7).

That is, as shown in the spectra of FIGS. 4A and 4C, the in-phase noise component (ρ / 2) in the positive frequency band is eliminated, and the signal component A + S ⁺ (t) of the in-phase component is obtained. A + S ⁺ (t) is a complex function. Since double-sideband broadcasting can be demodulated only in one band, the AM broadcast can be demodulated by the real part of this signal component S ⁺ (t).
The above processing procedure in the case where noise is superimposed only on the upper side band is easy to understand from the input signal and the mathematical expression and spectrum of the output signal shown in FIG.

In the RF band, when noise is superimposed only on the lower sideband band, x _r (t) and x _i (t) are expressed as ω _n = −ω _n in the expressions (3) and (4). It is expressed by a set expression. However, in order to make the expressions of the expressions (3) and (4) the same as when noise is superimposed on the upper sideband, ω _n = −ω _{n is set} , and ρ is ρ ^* and ρ ^* is ρ Replace with. That is, ρ in Equation (1) when noise is superimposed on the lower sideband band is ρ ^* . By defining the noise spectrum in this manner, the noise component of the in-phase component is the same as that in FIG. 3A, and is ρ ^* / 2 in the negative frequency band, and ρ / 2 in the positive frequency band. They are in a complex conjugate relationship.

The noise component of the orthogonal component is a spectrum (jρ / 2) in the positive frequency band and a spectrum (−jρ ^* / 2) in the negative frequency band. That is, in the positive frequency band and the negative frequency band, the phase of the noise component of the quadrature component is advanced by π / 2 on the time axis with respect to the noise component of the in-phase component. Therefore, in order to remove the noise component (ρ / 2) of the in-phase component in the positive frequency band, the noise component (jρ / 2) of the quadrature component in the positive frequency band is multiplied by (j) and added to the in-phase component. There is a need to. In this case, the calculation by the multiplier 41 is multiplication of (j). In this way, the sign of the multiplication factor (j) by the multiplier 41 differs between when noise is superimposed on the upper sideband and when noise is superimposed on the lower sideband. The changeover switch 42 is operated so that the listener of the AM broadcast judges the degree of noise mixing in the reproduced sound and the noise is eliminated by the listener.

In the above, the case where the noise component of the quadrature component in the positive frequency band is extracted has been described. However, the quadrature component in the negative frequency band is extracted, and the noise component of the in-phase component is removed from the in-phase component in the negative frequency band. Also good. When noise is superimposed on the upper side band, since the spectrum of the noise component of the orthogonal component in the negative frequency band is (jρ ^* / 2), the multiplication factor in the multiplier 41 is (j). Further, when noise is superimposed on the lower sideband, the spectrum of the noise component of the orthogonal component in the negative frequency band is (−jρ ^* / 2), so the multiplication factor in the multiplier 41 is (−j ) Thereby, A + S ⁻ (t) is output from the synthesizing unit 60, and an AM audio signal can be obtained from the real part.
In addition, although the term of the noise component in the equations (1) to (6) is expressed by a single spectrum, it is actually expressed by an integral of the frequency ω _n .

In the first embodiment, the sign of the multiplication factor (j) in the multiplier 41 is switched by the setting switch 42. In the second embodiment, the sign of the multiplication factor (j) in the multiplier 41 is automatically determined by the cross-correlation operation between the in-phase component (signal component + noise component) in the positive frequency band and the noise component of the orthogonal component. It is the device which was made. The signal processing apparatus of this embodiment is configured as shown in FIG. Parts having the same functions as those in FIG. 1 of the first embodiment are denoted by the same reference numerals. The only difference is the configuration of the phase rotation unit 40 with respect to FIG. The complex conjugate computing unit 43 computes the complex conjugate of the in-phase component x _r ⁺ (t) in the positive frequency band of the expression (5) output from the filter 31, and the multiplier 44 outputs the complex conjugate and the output of the filter 32. The product of the orthogonal component of the positive frequency band in equation (8) and the noise component x _i ⁺ (t) is calculated.

Since x _i ⁺ (t) has a different sign when noise is superimposed only on the upper sideband band and when noise is superimposed only on the lower sideband band, in equation (8), both include ± It is expressed with the sign of. (+) Represents the upper sideband band, and (−) represents the case where noise is superimposed on the lower sideband band.

（９）式の第１行における雑音成分は、周波数ω_n の単一周波数の雑音を示しているが、実際には雑音成分のスペクトルρはω_n の関数ρ（ω_n ）である。雑音成分は一定幅の帯域を有しているので、正確には、（９）式の積のそれぞれの雑音成分は、ω_n に関する積分表示となる。したがって、（９）式の最終行の第２項は、正確には、ω_n に関する積分表示となる。また、（９）式の｜ρ｜² は、スペクトルの振幅の２乗のω_n に関する積分であり、実数で正の値である。 The output signal of the multiplier 44 is expressed by equation (9).

(9) the noise component in the first row of the equation, while indicating noise of a single frequency of the frequency omega _n, is actually a spectrum [rho noise component functions ω _n ρ (ω _n). Since the noise component has a fixed bandwidth, precisely, each noise component of the product of the equation (9) is an integral display regarding ω _n . Therefore, the second term in the last row of the equation (9) is accurately an integral display relating to ω _n . In addition, | ρ | ^{2 in the} equation (9) is an integral with respect to the square of the spectrum amplitude ω _n and is a real positive value.

The signal of the expression (9) output from the multiplier 44 is input to the moving average calculation unit 45. Since the product of noise having different frequencies due to the moving average is 0, only the product of noise components having the same frequency remains. The second term in the last row of the equation (9) is a waveform obtained by amplitude-modulating a sine wave of noise frequency ω _n with a conjugate A + S ⁺ (t) ^* of the in-phase component in the positive frequency band when viewed in a single frequency spectrum. It becomes. Therefore, by calculating the moving average over a time sufficiently longer than the shortest cycle determined by the frequency ω _n and the maximum frequency of the in-phase component, the noise component of the frequency ω _n is eliminated. When the noise has a certain band, the second term is merely an integral display of ω _n , and thus the component that varies depending on the noise frequency of the second term can be eliminated. As a result, the output signal of the moving average calculation unit 45 is expressed by equation (10).

Equation (10) is a pure imaginary number. When the sign is (+), the noise is superimposed on the upper sideband band. When the sign is (−), the noise is lower sideband. It means that it is superimposed on the wave band. Therefore, the phase detection unit 46 determines whether the expression (10) is (−j) or (j). That is, it is determined whether the phase of the noise component of the quadrature component is in the delayed phase delayed by π / 2 or the advanced phase advanced by π / 2 with respect to the phase of the noise component of the in-phase component. Then, in the phase determination unit 47, if the sign of the expression (10) is (+), (-j) is outputted to the multiplier 41, and if the sign is (-), (j) is outputted to the multiplier 41. The phase rotation noise component is output from the synthesizer 41 and added to the in-phase component x _r (t) of the equation (5) in the synthesizer 40. As a result, even when noise is superimposed on one of the upper sideband band and the lower sideband band, the noise component (ρ / 2) of the in-phase component in the positive frequency band can be automatically removed. it can. Note that even if the impulse response of the Hilbert filter, which is one of the constituent elements of the filters 31 and 32, is approximated by a finite term, the band separation is not perfect, and even if the real part of DC is present in the equation (9) The phase rotation direction of the quadrature component can be determined by the sign of the imaginary part of equation (9).

  The above processing is the same when the noise component of the negative component in the negative frequency band is extracted. The filter 31 in FIG. 6 may be a filter that extracts the negative frequency band of the in-phase component, and the filter 32 may be a filter that extracts the negative frequency band of the quadrature component.

In the first and second embodiments, noise removal and demodulation are performed for one of the positive frequency band and the negative frequency band. In the third embodiment, as shown in FIG. 7, the signal processing is performed independently for each of the positive frequency band and the negative frequency band. That is, similarly to the processing in the positive frequency band in the first embodiment, the noise component of the orthogonal component in the negative frequency band is extracted, and the noise component superimposed on the in-phase component is removed. In FIG. 7, an upper side band processing unit U that removes noise in the positive frequency band and a lower side band processing unit L are provided. The upper side band processing unit U has the same configuration as that of FIG. 1, and the lower side band processing unit L has the same configuration as that of FIG. 1, but the filter 31L extracts the negative frequency band of the in-phase component and the filter 32L has the negative frequency of the quadrature component. The only difference is that the band is extracted. Then, S _U (t) output from the upper side band processing unit U and S _L (t) output from the lower side band processing unit L are combined by the combining unit 61, and the double side band S (t) is generated. can get.

As described in the second embodiment, S _U (t) output from the upper sideband processing unit U is expressed by Expression (11) including the case where noise is superimposed on the upper sideband or the lower sideband. .

For the negative frequency band, the waveform of the in-phase component is the same regardless of whether noise is superimposed on the upper sideband wave or noise on the lower sideband wave, and the output x _r ⁻ (t) of the filter 31L is _{expressed by} equation (12). Given in.

よって、（１３）式の符号が（−）であれば直交成分の雑音成分はπ／２の遅れ位相にあるので、ｘ_i ^- （ｔ）に（ｊ）を乗算して（π／２回転させて）、同相成分の雑音成分に対してπの遅れ位相にして、同相成分の雑音成分ｘ_r ^- （ｔ）に加算すれば、負周波数帯域の同相成分から雑音を除去することができる。 On the other hand, when the noise component of the quadrature component is superimposed only on the upper sideband wave, it is in a delayed phase of π / 2 with respect to the noise component of the in-phase component, and the noise is superimposed only on the lower sideband wave. In this case, the phase is π / 2 in advance with respect to the noise component of the in-phase component. Therefore, the output x _i ⁻ (t) of the filter 32L that extracts the negative frequency band of the orthogonal component is expressed by the following equation (13).

Therefore, if the sign of equation (13) is (−), the noise component of the orthogonal component is in a delayed phase of π / 2, so x _i ⁻ (t) is multiplied by (j) (π / 2 rotation). Therefore, if the phase is delayed by π with respect to the noise component of the in-phase component and added to the noise component x _r ⁻ (t) of the in-phase component, the noise can be removed from the in-phase component in the negative frequency band.

On the other hand, if the sign of equation (13) is (+), the noise component of the quadrature component is in the lead phase of π / 2, so x _i ⁻ (t) is multiplied by (−j) (−π / (2 rotations), if the phase of the noise component of the in-phase component is π and the phase is added to the noise component x _r ⁻ (t) of the in-phase component, the noise can be removed from the in-phase component of the negative frequency band. it can. This process is realized by the phase rotation unit 40L and the synthesis unit 60L. That is, the signal S _L (t) in the negative frequency band is obtained from the equation (14).

本実施例の場合に、位相回転部４０Ｕと位相回転部４０Ｌにおける位相回転は、実施例１と同様に共通の切換スイッチ４２により、位相回転の符号を反転させて、ＡＭ放送の聴者が雑音の少ない方に設定することになる。 Then, as shown in the equation (15), the combining unit 61 adds the output S _U (t) of the combining unit 60U and the output S _L (t) of the combining unit 60L, thereby removing the noise-removed both side bands. A wave signal S (t) can be obtained.

In the case of this embodiment, the phase rotation in the phase rotation unit 40U and the phase rotation unit 40L is reversed by the common changeover switch 42 in the same manner as in the first embodiment, so that the AM broadcast listener can hear noise. It will be set to the smaller one.

  In the third embodiment, the second embodiment that automatically detects the sign of the rotational phase may be used. That is, the phase rotation unit 40U of the upper side band processing unit U in FIG. 7 of the third embodiment and the phase rotation unit 40L of the lower side band processing unit L are used in the phase rotation unit 40 shown in FIG. Also good. According to this configuration, it is possible to obtain a baseband double sideband wave in which noise is automatically removed from the in-phase component. Since the phase rotation code determined for the positive frequency band and the phase rotation code determined for the negative frequency band are only different from each other, the phase rotation code determined for one frequency band is different. Can be used to multiply the code by (−1) to obtain a phase rotation code in the other frequency band. In this case, the phase rotation part 40 of FIG. 6 should exist only in the processing system of one frequency band.

In this embodiment, x _r ⁺ (t) and x _r ⁻ (t) are the first half-band signal component, the second half-band signal component, and x _i ⁺ (t) and x _i ⁻ (t) are, respectively. , Corresponding to the first single-band noise component and the second single-band noise component, respectively. The filters 31U and 31L correspond to first in-phase component band extraction means and second in-phase component band extraction means, and the filters 32U and 32L correspond to first quadrature component band extraction means and second quadrature component band extraction means. To do. The phase rotation unit 40U corresponds to the first phase rotation unit, and the phase rotation unit 40L corresponds to the second phase rotation unit. The combining unit 60U corresponds to the first combining unit, and the combining unit 60L corresponds to the second combining unit.

  In the first to third embodiments, it has been described that the demodulated carrier wave is synchronized with the modulated carrier wave at the time of demodulation. This embodiment is an embodiment in which the influence on noise removal due to the synchronization error is eliminated. FIG. 8 shows a configuration of a device that realizes synchronous demodulation in the signal processing device of the first embodiment. In FIG. 8, the feature of the present embodiment is that the quadrature demodulation unit 20 is provided with a phase synchronization processing unit 70. The phase synchronization processing unit 70 receives a baseband signal and calculates a moving average of the moving average calculating unit 71, a complex conjugate calculating unit 72 that calculates the complex conjugate of the output, and an amplitude that normalizes the amplitude of the output. A normalization unit 73 and a multiplication unit 74 that multiplies the output by the baseband signal x (t).

受信信号ｒ（ｔ）は（１）式の実部で表されるので、ミキサー２１の出力するベースバンド信号ｘ（ｔ）は、（２）式と同様にして、（１７）式で表される。すなわち、ベースバンド信号に、ｅｘｐ（Δωｔ）の因子が現れる。

このベースバンド信号ｘ（ｔ）が移動平均演算部７１によりその移動平均が演算される。移動平均の結果は、（１８）式で与えられる。

すなわち、移動平均により、（１７）式の最終項の周波数は大きいので、移動平均により、この項は０となる。 The frequency of the demodulated carrier wave output from the demodulated carrier wave generator 22 is assumed to be larger by Δω than the frequency ω _{c of the} modulated carrier wave. That is, the demodulated carrier wave L (t) is expressed by equation (16).

Since the received signal r (t) is represented by the real part of the equation (1), the baseband signal x (t) output from the mixer 21 is represented by the equation (17) in the same manner as the equation (2). The That is, a factor of exp (Δωt) appears in the baseband signal.

The moving average of the baseband signal x (t) is calculated by the moving average calculator 71. The result of the moving average is given by equation (18).

That is, since the frequency of the final term of the equation (17) is large due to the moving average, this term becomes 0 due to the moving average.

Next, the complex conjugate of the equation (18) is obtained by the complex conjugate computing unit 72, and the normalized signal of the equation (19) is obtained by the amplitude normalizing unit 73. Since A + S ⁺ (t) + S ⁻ (t) in equation (18) is a real number, −jωt is obtained from equation (18) by −jωt = tan ⁻¹ (real part / imaginary part), so exp ( jωt).

この処理により、復調搬送波の周波数が変調搬送波の周波数に対して偏差Δωを有していても、その偏差による影響を排除することができる。
なお、上記の説明では、受信信号に含まれる復調搬送波と、変調搬送波との位相差Δφは、明示していないが、（１６）〜（１８）式におけるｊΔωｔをｊΔω＋ｊΔφとおいて、位相誤差Δφを考慮して、（２０）式を演算すると、Δφは消去されるので、Δφが存在しても、（２０）式が得られる。すなわち、周波数誤差だけでなく位相誤差も、補償されることになる。 Next, the multiplier 74 can multiply the baseband signal by the normalized signal to obtain the synchronized baseband signal x _sync (t) of the equation (20).

By this processing, even if the frequency of the demodulated carrier wave has a deviation Δω with respect to the frequency of the modulated carrier wave, the influence of the deviation can be eliminated.
In the above description, the phase difference Δφ between the demodulated carrier wave and the modulated carrier wave included in the received signal is not clearly shown, but jΔωt in equations (16) to (18) is set to jΔω + jΔφ, and the phase error Δφ is calculated. Considering this, if the equation (20) is calculated, Δφ is deleted, so that even if Δφ exists, the equation (20) can be obtained. That is, not only the frequency error but also the phase error is compensated.

The apparatus of the fourth embodiment is an example in which the phase synchronization processing unit 70 is used in the configuration of FIG. 1 of the first embodiment, but the quadrature demodulation unit 20 of FIG. 70 may be provided. The configuration is shown in FIG.
As for synchronous demodulation, the in-phase component is demodulated by cos [(ω _c + Δω) t + φ], and the quadrature component is demodulated by sin [(ω _c + Δω) t + φ] so that the quadrature component becomes 0 (minimum). The frequency ω _c + Δω and the phase φ of the demodulated carrier wave may be synchronously demodulated by a PLL circuit using VOC, that is, a Costas loop.

  In this embodiment, in the configuration of the third embodiment shown in FIG. 7, the quadrature demodulation unit 20 is provided with the phase synchronization processing unit 70 of the fourth embodiment shown in FIG. This is an example in which the configuration in which the signs of different phase rotations are automatically detected according to the second embodiment is employed when the is superimposed. The configuration is shown in FIG. That is, in the upper side band processing unit U and the lower side band processing unit L in FIG. 10, the same configuration as the phase rotation unit 40 in FIG. 6 is used as the phase rotation unit 40U and the phase rotation unit 40L. In this embodiment, the phase rotation code of the quadrature component in the positive frequency band is determined by the cross-correlation between the in-phase component and the quadrature component in the positive frequency band, and the cross-correlation between the in-phase component and the quadrature component in the negative frequency band. The phase rotation code of the orthogonal component in the negative frequency band is determined. Of course, a code obtained by inverting the phase rotation code determined from one band may be used as the phase rotation code in the other band.

In this embodiment, the complex conjugate of the noise component of the in-phase component obtained in one frequency band is calculated and synthesized with the in-phase component of the other frequency band, thereby superimposing it on the in-phase component of both frequency bands. This is an example of removing noise. The configuration is shown in FIG.
The configuration of the phase rotation unit related to the positive frequency band in the upper side band processing unit U is the same as that of the phase rotation unit 40U of the upper side band processing unit U of FIG. The lower sideband processing unit L is provided with a phase inversion unit 401L and a synthesis unit 60L in place of the phase rotation unit for the negative frequency band. And the phase inversion part 401L has the complex conjugate calculating part 43L. The output of the multiplication unit 41U of the upper side band processing unit U is input to the complex conjugate calculation unit 43L. The case where noise is superimposed only on the upper side band will be described. The output x _ia ⁺ (t) of the multiplication unit 41U is expressed by Expression (21) obtained by multiplying Expression (6) by −j.

複素共役演算部４３Ｌの出力（ｘ_ia ⁺ （ｔ））^* は、ｘ_ia ⁺ （ｔ）の複素共役であり、（２２）式で表される。

合成部６０Ｌで、複素共役演算部４３Ｌの出力である（２２）の（ｘ_ia ⁺ （ｔ））^* と、フィルタ３１Ｌから出力される（１２）式のｘ_r ^- （ｔ）を加算合成すれば、負周波数帯域の同相雑音が除去される。すなわち、（２３）式により、Ｓ_L （ｔ）を得ることができる。

したがって、合成部６１の出力は、（１５）式のＳ（ｔ）となり、同相成分から雑音が除去されたＤＳＢ信号が得られる。

The output (x _ia ⁺ (t)) ^* of the complex conjugate calculation unit 43L is a complex conjugate of x _ia ⁺ (t), and is expressed by equation (22).

The synthesizer 60L adds and synthesizes (x _ia ⁺ (t)) ^{* of} (22), which is the output of the complex conjugate calculator 43L, and x _r ⁻ (t) of equation (12) output from the filter 31L. For example, in-phase noise in the negative frequency band is removed. That is, S _L (t) can be obtained from the equation (23).

Therefore, the output of the synthesizing unit 61 is S (t) in the equation (15), and a DSB signal from which noise is removed from the in-phase component is obtained.

したがって、乗算部４１Ｕでの乗算は（ｊ）となり、乗算部４１Ｕの出力ｘ_ia ⁺ （ｔ）は、上側帯波帯域に雑音が重畳した場合の（２１）式と同一となる。よって、複素共役演算部４３Ｌの出力（ｘ_ia ⁺ （ｔ））^* は、（２２）式と同一となる。同相成分については、上側帯波帯域にのみ雑音が重畳した場合と同一であるから、フィルタ３１Ｌの出力ｘ_r ^- （ｔ）も（１２）式と同一である。したがって、合成部６０Ｌの出力Ｓ_L （ｔ）も、（２３）式となり、合成部６１の出力Ｓ（ｔ）も（１５）式となる。このようにして、下側帯波帯域にのみ雑音が重畳された場合にも、同相成分から雑音が除去されたＤＳＢ信号が得られる。 When noise is superimposed only on the lower sideband, as described above, the in-phase component x _r (t) is the same as the in-phase component when noise is superimposed only on the upper sideband, so that (3) It is expressed by a formula. The quadrature component is an equation obtained by inverting the sign of the quadrature component in the equation (4) when noise is superimposed only on the upper sideband. Therefore, the orthogonal component x _i (t) is expressed by the equation (24).

Therefore, the multiplication in the multiplication unit 41U is (j), and the output x _ia ⁺ (t) of the multiplication unit 41U is the same as the equation (21) when noise is superimposed on the upper side band. Therefore, the output (x _ia ⁺ (t)) ^* of the complex conjugate calculation unit 43L is the same as the expression (22). Since the in-phase component is the same as when noise is superimposed only on the upper sideband, the output x _r ⁻ (t) of the filter 31L is also the same as the equation (12). Accordingly, the output S _L (t) of the synthesizing unit 60L is also expressed by the equation (23), and the output S (t) of the synthesizing unit 61 is also expressed by the equation (15). In this way, even when noise is superimposed only on the lower sideband, a DSB signal from which noise has been removed from the in-phase component is obtained.

  The present invention can be used in an apparatus for removing periodic noise from an input signal.

20 ... Quadrature demodulation unit 23 ... In-phase component extraction unit 24 ... Quadrature component extraction unit 31, 32 ... Filter U ... Upper side band processing unit L ... Lower side band processing unit

Claims (14)

In a signal processing device that receives a double sideband signal and removes noise superimposed only on one sideband in the RF band,
Demodulating means for orthogonally demodulating the both sideband signals and demodulating into a baseband signal having a positive frequency band and a negative frequency band;
A signal processing apparatus comprising: a noise removing unit that removes a noise component included in an in-phase component output from the demodulating unit based on a quadrature component output from the demodulating unit. An orthogonal component band extracting means for extracting a single band noise component of one of the positive frequency band and the negative frequency band among the orthogonal components output from the demodulation means;
The noise removing means includes
Phase rotation means for rotating the phase of the one-band noise component extracted by the orthogonal component band extraction means by −π / 2 or π / 2;
Combining a phase rotation noise component output from the phase rotation unit with the in-phase component output from the demodulation unit and removing the noise component included in the in-phase component. Item 2. The signal processing device according to Item 1. In-phase component band extraction means for extracting one of the positive frequency band and the negative frequency band among the in-phase component output from the demodulation means,
The phase rotation means includes cross-correlation calculation means for calculating a cross-correlation value between the one-band noise component extracted by the quadrature component band extraction means and the one-band signal component extracted by the in-phase component band extraction means. The signal processing apparatus according to claim 2, further comprising: determining a phase rotation direction based on a sign of a value output from the cross-correlation calculating unit.   The cross-correlation calculating means calculates a product of one and the other complex conjugate of the one-band signal component output by the in-phase component band extracting means and the one-band noise component extracted by the quadrature component band extracting means. The signal processing apparatus according to claim 3, wherein the cross-correlation value is calculated based on a moving time average.   The said synthetic | combination means synthesize | combines the said phase rotation noise component and the said one band signal component extracted by the said in-phase component band extraction means, The any one of Claim 3 thru | or 4 characterized by the above-mentioned. Signal processing equipment. The in-phase component band extracting unit extracts a first in-phase component band extracting unit that extracts the one-band signal component as a first one-band signal component, and a second of the other frequency band with respect to the first one-band signal component. Second in-phase component band extracting means for extracting one-band signal component;
The phase rotation unit includes a first phase rotation unit that rotates the phase of the one-band noise component extracted by the quadrature component band extraction unit by −π / 2 or π / 2, and an output of the first phase rotation unit. Phase inversion means for calculating the complex conjugate of the signal to be
The synthesizing unit removes noise superimposed on the first single band signal component from the first single band signal component based on the first phase rotation noise component output from the first phase rotation unit. Synthesizing means; and second synthesizing means for removing noise superimposed on the second one-band signal component from the second one-band signal component based on the phase-inverted noise component output from the phase inverting means. The signal processing device according to claim 3, wherein the signal processing device is a signal processing device. The in-phase component band extracting unit extracts a first in-phase component band extracting unit that extracts the one-band signal component as a first one-band signal component, and a second of the other frequency band with respect to the first one-band signal component. Second in-phase component band extracting means for extracting one-band signal component;
The quadrature component band extracting means extracts first quadrature component band extracting means for extracting a noise component in the frequency band of the same sign as the first single band signal component as a first single band noise component; On the other hand, the second orthogonal component band extracting means for extracting the second single band noise component of the other frequency band,
The phase rotation means includes first phase rotation means for rotating the phase of the first one-band noise component extracted by the first orthogonal component band extraction means by −π / 2 or π / 2, and the second orthogonal rotation. Second phase rotation means for rotating the phase of the second one-band noise component extracted by the component band extraction means by π / 2 or −π / 2,
The synthesizing unit removes noise superimposed on the first single band signal component from the first single band signal component based on the first phase rotation noise component output from the first phase rotation unit. A second combining unit that removes noise superimposed on the second one-band signal component from the combining unit and the second one-band signal component based on the second phase rotation noise component output from the second phase rotating unit; The signal processing apparatus according to any one of claims 3 to 5, wherein the signal processing apparatus includes:   The sign of the rotational phase by the first phase rotating means and the second phase rotating means is determined by the sign of the cross-correlation value determined by one of the phase rotating means and the sign of the phase rotation by the other phase rotating means. The signal processing apparatus according to claim 7, wherein: The first phase rotation means includes first cross-correlation calculation means as the cross-correlation calculation means for calculating the cross-correlation value from the first single-band noise component and the first single-band signal component, Determining the rotational direction of the phase based on the first value output from the first cross-correlation calculating means;
The second phase rotation means includes second cross-correlation calculation means as the cross-correlation calculation means for calculating a cross-correlation value between the second half-band noise component and the second half-band signal component, The signal processing apparatus according to claim 7, wherein the phase rotation direction is determined based on the second value output from the two cross-correlation calculating means.   The demodulation means has a phase-locked loop unit that controls the frequency and phase of the demodulated carrier so that the beat signal of the error frequency of the demodulated carrier with respect to the modulated carrier contained in the quadrature component after quadrature demodulation is zero. The signal processing device according to claim 1, wherein the signal processing device is a signal processing device.   The demodulating means obtains a beat signal of an error frequency of the demodulated carrier wave with respect to the modulated carrier wave from the moving average of the baseband signal, and newly corrects the fluctuation of the baseband signal due to the beat signal based on the beat signal. The signal processing apparatus according to claim 1, further comprising a synchronization unit configured as a baseband signal. In a signal processing method for receiving a double sideband signal and removing noise superimposed on only one sideband in the RF band,
Quadrature demodulation of the both sideband signals, demodulated into a baseband signal having a positive frequency band and a negative frequency band,
A signal processing method comprising removing a noise component included in a demodulated in-phase component based on the demodulated quadrature component. Extracting the noise component of one of the positive frequency band and the negative frequency band in the orthogonal component as a single band noise component,
Rotate the phase of the extracted one-band noise component by −π / 2 or π / 2,
The signal processing according to claim 12, wherein noise superimposed on the signal component is removed based on a phase rotation noise component obtained by phase-rotating the demodulated in-phase component baseband signal component. Method. The signal component of one of the positive frequency band and the negative frequency band in the in-phase component is extracted as a single band signal component,
14. The signal processing method according to claim 12, wherein the sign of phase rotation is determined based on a cross-correlation value between the one-band signal component and the one-band noise component.

Images