CN112910333A - Control method, device and circuit for motor drive and variable frequency air conditioner - Google Patents

Control method, device and circuit for motor drive and variable frequency air conditioner Download PDF

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CN112910333A
CN112910333A CN202110164992.1A CN202110164992A CN112910333A CN 112910333 A CN112910333 A CN 112910333A CN 202110164992 A CN202110164992 A CN 202110164992A CN 112910333 A CN112910333 A CN 112910333A
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pwm
power factor
inverter
time
switching tube
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CN112910333B (en
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盛爽
郑长春
黄勇
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Guangdong Xita Frequency Conversion Technology Co ltd
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Guangdong Xita Frequency Conversion Technology Co ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/28—Arrangements for controlling current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)
  • Inverter Devices (AREA)

Abstract

The invention relates to a control method, a device, a circuit and a variable frequency air conditioner for driving a motor, wherein an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter is obtained, and a power factor PWM time sequence of a switching tube of a PFC (power factor correction) circuit is determined according to the inversion PWM time sequence, wherein the conduction time of the switching tube in a power factor PWM signal is at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM, and finally the switching tube of the PFC circuit is controlled to work according to the power factor PWM corresponding to the power factor PWM time sequence, so that the magnitude of a capacitance current can be effectively reduced, the volume of a filtered electrolytic capacitor in a filter module is reduced, the volume of the whole control circuit board is reduced, and the cost is reduced.

Description

Control method, device and circuit for motor drive and variable frequency air conditioner
Technical Field
The invention relates to a control method, a control device and a control circuit for motor driving and a variable frequency air conditioner, and belongs to the technical field of air conditioners.
Background
The alternating current-direct current-alternating current voltage type frequency converter is generally used in the frequency conversion air conditioner, a direct current energy storage filtering link exists in the middle of the frequency conversion air conditioner, a large-capacity electrolytic capacitor is generally adopted to carry out filtering on direct current bus voltage, the large-capacity electrolytic capacitor is easy to generate text wave current during working, and in order to solve the problem, the ripple current generated by a single large electrolytic capacitor is reduced by adopting a mode that a plurality of small-capacity electrolytic capacitors are connected in parallel, but the size of a controller can be increased by the scheme, so that the problems of installation difficulty and the like are increased.
Disclosure of Invention
The invention discloses a control method, a control device and a control circuit for motor driving and a variable frequency air conditioner, and aims to solve the problem of ripple current generated when an electrolytic capacitor works.
The invention provides a control method for motor drive, a control circuit of the motor drive comprises a controller and an inverter, and the control method comprises the following steps:
acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter;
determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in the power factor PWM signal is controlled to be at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM;
and controlling the switch tube of the PFC circuit to work according to the power factor PWM corresponding to the power factor PWM time sequence.
Optionally, a center time of a conduction time of a switching tube of the PFC circuit is aligned with a center time of the inverter PWM.
Optionally, the determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing includes:
generating a sawtooth carrier corresponding to the power factor PWM, wherein the period of the sawtooth carrier is half of a triangular carrier of the inverter PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as the starting time or the peak time of the triangular carrier;
comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a first power factor PWM;
comparing the value corresponding to the remaining half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a second power factor PWM;
and synthesizing the first PWM signal and the second PWM signal into power factor PWM.
The present invention also proposes a control device for motor drive, the control device comprising:
the PFC module is used for correcting the power factor of the input pulsating direct current to generate smooth direct current;
the inverter is used for converting the input direct current into three-phase alternating current so as to drive the motor to run;
the controller is configured to:
acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter;
determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in the power factor PWM signal is controlled to be at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM;
and controlling the PFC module to operate according to the power factor PWM corresponding to the power factor PWM time sequence.
Optionally, a center time of a conduction time of a switching tube of the PFC circuit is aligned with a center time of the inverter PWM.
Optionally, when determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing, the controller is further configured to:
generating a sawtooth carrier corresponding to the power factor PWM, wherein the period of the sawtooth carrier is half of a triangular carrier of the inverter PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as that of the triangular carrier;
comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a first power factor PWM;
comparing the value corresponding to the remaining half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a second power factor PWM;
and synthesizing the first PWM signal and the second PWM signal into power factor PWM.
The invention also provides a control circuit for the variable frequency air conditioner, which comprises a rectification module, a PFC module and a filtering module and is characterized by also comprising the control device for driving the motor;
the rectification module is used for rectifying alternating current input into the motor driving circuit and outputting pulsating direct current;
the PFC module is connected with the rectifying module to correct the power factor of the pulsating direct current;
the filtering module is connected with the PFC module and used for filtering the direct current output by the FC module and outputting smooth direct current, and the filtering module is connected with the direct current bus and supplies power to the control device through the direct current bus.
The invention also provides a variable frequency air conditioner which is characterized by being provided with the control circuit for the variable frequency air conditioner.
According to the technical scheme, the control method for driving the motor comprises the steps of obtaining an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter, determining a power factor PWM time sequence of a switching tube of a PFC (power factor correction) circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in a power factor PWM signal is at least partially overlapped with the time of a zero vector of the inverter corresponding to the inversion PWM, and finally controlling the switching tube of the PFC circuit to work according to the power factor PWM corresponding to the power factor PWM time sequence, so that the size of a capacitance current can be effectively reduced, the size of an electrolytic capacitor of filtering in a filtering module is reduced, the size of the whole control circuit board is reduced, and the cost is reduced.
Drawings
FIG. 1 is a circuit and block diagram of a motor drive circuit according to an embodiment of the present invention;
FIG. 2A shows a composite schematic of the space vectors of a six-phase step wave mode of the switching tubes of an inverter;
FIG. 2B shows a schematic diagram of current states corresponding to the six-phase space vector of FIG. 2A;
fig. 3 is a schematic diagram of a PWM waveform and a power factor PWM waveform of the inverter and a current waveform of each branch in the PFC circuit according to the embodiment of the present invention;
FIG. 4 is a topological circuit equivalent diagram of the switching state of the IGBT switching tube in the PFC circuit;
FIG. 5A is a block diagram of power factor PWM signal generation according to an embodiment of the present invention;
FIG. 5B is a block diagram of inverter PWM generation according to an embodiment of the present invention;
fig. 6 is a flowchart of a control method for motor driving according to an embodiment of the present invention;
fig. 7 is a schematic diagram of PWM waveforms and power factor PWM waveforms of the inverter and current waveforms of each branch in the PFC circuit according to another embodiment of the present invention.
Detailed Description
It is to be noted that the embodiments and features of the embodiments may be combined with each other without conflict in structure or function. The present invention will be described in detail below with reference to examples.
The invention provides a control method for motor driving. As shown in fig. 1, the motor driving circuit includes a rectifying module 8, a filtering module 9, a PFC module 7, a controller 1, and an inverter 3, wherein the rectifying module 8 rectifies an input ac power into a pulsating dc power, and the circuit may be a bridge rectifier circuit in fig. 1; the PFC module 7 is used for correcting power factors of the pulsating direct current output by the rectification module 8, the filtering module 9 filters the pulsating direct current output by the rectification module 8 and converts the pulsating direct current into smooth direct current, and the filtering module 9 is mainly composed of a large-capacity electrolytic capacitor (such as 400uF/450V) and supplies power to the inverter 3 by connecting a direct current bus; the system can further comprise a bus voltage sampling module 6 which is used for collecting the direct current bus voltage Vdc and outputting the direct current bus voltage Vdc to the controller 1, a current sampling module 5 which is connected in series in a direct current supply loop of the inverter 3 and used for collecting the working current of the inverter 3 and outputting the working current to the controller 1, and the controller 1 generates the three-phase current of the three-phase winding of the driving motor 4 of the inverter 3 through calculation; the controller 1 performs vector control according to the dc bus voltage Vdc and the phase current, and finally generates a PWM signal for driving the six switching tubes of the inverter 3, so as to control the inverter 3 to drive the motor 4 to operate.
Wherein some of the processing modules internal to the controller 1 are prior art, such as a speed regulator, a current regulator, a speed/position observer, respective coordinate converters including a Clarke converter, a Park converter and a Park inverse converter, and an SVPWM voltage modulator.
Fig. 4 shows a flowchart of a control method for motor driving according to an embodiment of the present invention, the control method including:
s100, acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter;
step S200, determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in the power factor PWM signal is controlled to be at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM;
and step S300, controlling the switch tube of the PFC circuit to work according to the power factor PWM corresponding to the power factor PWM time sequence.
In step S100, fig. 2A shows a synthetic diagram of space vectors of a six-phase step wave mode of switching tubes of an inverter, fig. 2B shows a diagram of a current state corresponding to the six-phase space vectors of fig. 2A, the two diagrams show that any one vector corresponding to the switch tube of the inverter is synthesized by two adjacent vectors, specifically one vector Vref is synthesized by two adjacent vectors V100 and V110 shown in FIG. 2A, the waveforms of the two vectors corresponding to the inverted PWM when they are combined are shown in fig. 3, where SA, SB, SC show the inverted PWM waveform diagrams of the switching tubes of the three legs of the inverter, which is a combination of two adjacent vectors V100 and V110, plus a zero vector, to form these three PWM waveforms, and correspondingly outputting three values to the triangular wave where the carrier wave of the PWM wave is positioned for comparison so as to generate the three paths of PWM waveforms. The inversion PWM sequence is formed by the start and end moments of the high level and the low level in the three PWM waveforms.
In step S200, the power factor PWM signal of the switching tube of the PFC circuit is generated as S in fig. 3IGBTAnd (4) waveform. Fig. 4 shows an equivalent circuit diagram of a topology circuit of the PFC circuit in the on and off states of the switching tube, i.e. the IGBT, as can be seen from fig. 4, when the IGBT of the PFC module 7 is on, the power supply charges the inductor of the PFC module 7, the capacitor of the filter module 9 discharges the inverter, the capacitor current is equal to the dc side current of the inverter, and when the IGBT is off, the capacitor current is equal to the difference between the inductor current and the dc side current. Namely, the following formula holds:
Figure BDA0002937425170000051
wherein IC is electrolytic capacitor current, IDCIs the DC side current of the inverter, IL is the inductive current, I is when the IGBT is ON, i.e. is turned ONFRDI.e. the current of the inverter diode is zero, while the IGBT is OFF, i.e. OFFFRDEqual to IL.
Further, the relationship between the dc-side current IDC and the switching state S, which can be combined with the switching states of the switching tubes of the inverter of fig. 2B, is as follows:
Figure BDA0002937425170000052
wherein IA, IB and IC are currents output by the inverter to A, B, C phase windings of the motor.
The two formulas are combined to obtain:
IC＝IFRD-IDC
＝(1-SIGBT)IL-[-(1-SA)IA-(1-SB)IB-(1-SC)IC]
wherein SIGBTA 1 indicates that the IGBT is on, and 0 indicates off; sxAnd (x is A/B/C) represents the state of the switching tube on the x-phase bridge arm of the inverter, wherein the switching tube on the phase is turned on when the value is 1, and the switching tube on the phase is turned off when the value is 0.
From the above formula, the electrolytic capacitor current IC and the IGBT switch state SIGBTInductor current IL and switching state S of inverterxIn this regard, the inductor current IL is generally associated with the PFC algorithm and is generally relatively fixed, while the IGBT switch state SIGBTAnd the switching state S of the inverterxThe correspondence relationship between the power factor PWM signal and the inverter PWM signal, i.e. the time sequence of the two signals, of the switching tube of the PFC circuit has a significant influence on the electrolytic capacitor current IC, and the following situations specifically exist:
first, the IGBT is aligned with the inverter PWM at the center moment of turn-on of the IGBT, as shown in the figure3, power factor PWM, i.e. IGBT tube PWM signal SIGBTThe center time of the middle conduction time is aligned with the center time of the inversion PWM, namely SA, SB and SC. When the IGBT is turned on, the inverter vector corresponding to the corresponding inverter PWM at this time is 111, i.e., the zero vector at this time, and thus the dc side current IDCIs zero, and the electrolytic capacitor current IC is equal to the direct current side current I when the IGBT is conductedDCTherefore, the electrolytic capacitor current IC is zero at this time. When the IGBT is turned off, the turn-off time period of the IGBT is aligned with the non-zero vectors of the inverter, i.e., the time periods T1 and T2 in fig. 3, and at this time, the capacitor current is the difference between the inductor current and the dc side current, and since the dc side current is positive in the motor electromotive state, the difference is smaller than the inductor current. In this case, as can be seen from fig. 3, the inverter zero vector corresponding to the inverted PWM, that is, the time periods other than T1 and T2 in one cycle of the inverted PWM, completely includes the on time period in the IGBT tube PWM signal. As can be seen from fig. 3, the capacitance current IC mainly consists of zero current during the on period of the IGBT, the inductance current IL during the off period of the IGBT, and the difference between the inductance current IL and the dc side current IDC, so the effective value of the capacitance current IC is certainly smaller than the inductance current IL, and may even be smaller than the current of the motor winding.
Secondly, at the turn-off time of the IGBT, the power factor PWM is aligned with the center of the inverter PWM, as shown in FIG. 7, namely the PWM signal S of the IGBT tubeIGBTThe center time of the middle off time is aligned with the center time of the inversion PWM, i.e., SA, SB, SC. When the IGBT is turned on, the inverter vector corresponding to the corresponding inverter PWM at this time is a non-zero vector, i.e., time periods T1 and T2 in fig. 7, and the capacitance current IC is equal to the inverted dc side current IDC at this time, and is a negative value; when the IGBT is turned off, the inverter vector corresponding to the inverter PWM corresponding to the time period of the turn-off time is 111, that is, the inverter vector is a zero vector at this time, so the dc side current IDC is zero, and the capacitive current IC at this time is equal to the inductive current IL. As can be seen from fig. 7, the non-zero vector of the inverter corresponding to the inverted PWM completely corresponds to the on-time of the PWM signal of the IGBT. As can be seen from fig. 7, the capacitive current IC mainly consists of the inductive current IL and the inverted dc side current IDC, and the effective value of the capacitive current IC is the inductive currentIL and the motor winding current are superimposed and therefore the magnitude of the current is significantly greater than in the first case.
Thirdly, the central moment of the IGBT conduction is between the inversion PWM center and the inversion PWM starting moment, and the IGBT tube PWM signal S is at the momentIGBTThe part of the time of the on-state coincides with the zero vector of the inverted PWM. The magnitude of the effective value of its capacitive current IC lies between the first and second cases.
From the above analysis, the capacitance current IC is the largest in the second case, and the capacitance current IC is smaller in the first and third cases, so if the inverter and the PFC circuit operate in the first and third modes, the venturi current on the filter module can be effectively reduced, and the reduction of the capacitance current enables the electrolytic capacitor in the filter module with smaller capacity to be adopted, so that the volume of the electrolytic capacitor occupies a larger space of the whole control circuit, and the reduction of the volume of the electrolytic capacitor can effectively reduce the volume of the control circuit board, thereby reducing the cost.
In the first and third modes, the zero vector of the inverse PWM is at least partially overlapped with the on time of the IGBT in the power factor PWM, wherein the whole on time of the IGBT in the first condition is within the zero vector time of the inverse PWM. Therefore, after the time sequence of the inverter PWM is obtained, the time sequence of the power factor PWM can be determined according to the time sequence, so that at least part of the zero vector of the inverter PWM is overlapped with the conduction time of the IGBT in the power factor PWM, the capacitance current IC is reduced, the size and the cost of the control circuit board are finally reduced, and the working reliability of the whole control circuit is improved.
In step S300, the switching tube of the PFC circuit operates according to the determined timing of the power factor PWM, so as to control the operation of the PFC circuit, thereby improving the operational reliability thereof.
The specific process of generating the duty ratio value in the power factor PWM of the switching tube is the prior art, and a single-cycle or average current scheme may be referred to, which is not described in detail herein, and the generation method of the duty ratio corresponding to the inverter PWM is also the prior art, which is not described in detail herein.
According to the control method for the motor drive, the inversion PWM time sequence of three bridge arms of the inverter is obtained, the power factor PWM time sequence of the switching tube of the PFC circuit is determined according to the inversion PWM time sequence, wherein the conduction time of the control switching tube in the power factor PWM signal is at least partially overlapped with the time of the inverter zero vector corresponding to the inversion PWM, and finally the switching tube of the PFC circuit is controlled to work according to the power factor PWM corresponding to the power factor PWM time sequence, so that the size of the capacitor current can be effectively reduced, the size of the filtered electrolytic capacitor in the filter module is reduced, the size of the whole control circuit board is reduced, and the cost is reduced.
In some embodiments of the present invention, determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing comprises:
step S210, generating a sawtooth carrier corresponding to the power factor PWM, wherein the cycle of the sawtooth carrier is half of a triangular carrier of the inverse PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as the starting time or the peak time of the triangular carrier;
step S220, comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a first power factor PWM;
step S230, comparing a value corresponding to the remaining half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a second power factor PWM;
in step S240, the first PWM signal and the second PWM signal are combined into a power factor PWM.
Compared with the previous embodiment, the embodiment provides one implementation mode of determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing.
In step S210, as shown in fig. 3, the period of the sawtooth wave carrier IGBT CNT is half of the triangular carrier PWM CNT of the inverter PWM of the three-way bridge arm, and the start time of the sawtooth wave carrier IGBT CNT is the same as the peak time of the triangular carrier PWM CNT, or the start time of the sawtooth wave carrier IGBT CNT is the same as the peak time of the triangular carrier PWM CNT. As shown in fig. 3, the start time of the first sawtooth wave carrier IGBT CNT from the left is the same as the start time of the triangular carrier PWM CNT, and the start time of the second sawtooth wave carrier IGBT CNT is the same as the peak time of the triangular carrier PWM CNT.
In step S220, a value Duty corresponding to a Duty ratio of a switching tube, i.e., an IGBT, of the PFC circuit is divided into two parts, wherein a half Duty/2 is compared with a sawtooth wave carrier IGBT CNT to generate a first power factor PWM, such as a PWM waveform S1 in fig. 3IGBT。
In step S230, the remaining half 1-Duty/2 of the Duty ratio Duty of the switching tube, i.e. IGBT of the PFC circuit is compared with the sawtooth wave carrier IGBT CNT to generate a second power factor PWM, such as the PWM waveform S2 in fig. 3IGBT。
In step S240, the first PWM signal and the second PWM signal are superimposed to form a power factor PWM, where the two PWM signals are superimposed through an or gate to finally generate a PWM with a power factor higher than the power factor. As shown in fig. 5A, the Duty ratio of the IGBT is divided into two paths, one of the two paths is divided into Duty/2 by the 1/2 proportional amplifier, and the Duty/2 and the triangular carrier PWM CNT are sent to the first comparator to generate the first power factor PWM, the other path is divided into the remaining 1-Duty/2 by the subtractor, and the remaining 1-Duty/2 and the triangular carrier PWM CNT are sent to the second comparator to generate the second power factor PWM, and finally the two paths of power factor PWM are sent to the or gate to be superimposed to generate the power factor PWM signal.
The conducting center time in the generated power factor PWM is aligned with the center of the inverter PWM, so that when an IGBT switching tube for controlling PFC works, the current of an electrolytic capacitor current IC is minimum, the capacity of the electrolytic capacitor is effectively reduced, the volume of the whole control circuit board is reduced, and the cost is reduced.
The invention also provides a control device for motor drive, the motor drive where the control device is located is shown in fig. 1, the control device comprises a rectifying module 8, a PFC module 7, a filtering module 9 and a controller, wherein the rectifying module 8 rectifies input alternating current into pulsating direct current, and the circuit can be a bridge rectifier circuit in fig. 1; the PFC module 7 is configured to perform power factor correction on the pulsating direct current output by the rectification module 8, the filtering module 9 is configured to filter the pulsating direct current output by the rectification module 8 and convert the pulsating direct current into a smooth direct current, the filtering module 9 is mainly configured to supply power to the inverter 3 by connecting a large-capacity electrolytic capacitor (e.g., 400uF/450V) and by connecting a direct-current bus, and may further include a bus voltage sampling module 6 configured to collect the direct-current bus voltage Vdc and output the collected direct-current bus voltage Vdc to the controller 1, the current sampling module 5 is connected in series to a direct-current power supply loop of the inverter 3 and configured to collect a working current of the inverter 3 and output the working current to the controller 1, and the controller 1 generates a three-phase current of a three-phase winding; the controller 1 performs vector control according to the dc bus voltage Vdc and the phase current, and finally generates a PWM signal for driving the six switching tubes of the inverter 3, so as to control the inverter 3 to drive the motor 4 to operate. Wherein some of the processing modules internal to the controller 1 are prior art, such as a speed regulator, a current regulator, a speed/position observer, respective coordinate converters including a Clarke converter, a Park converter and a Park inverse converter, and an SVPWM voltage modulator.
Wherein the controller is configured to: acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of an inverter; determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in the power factor PWM signal is controlled to be at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM; and controlling the PFC module to operate according to the power factor PWM corresponding to the power factor PWM time sequence.
Fig. 2A shows a resultant schematic of space vectors of a six-phase step wave mode of switching tubes of an inverter, fig. 2B shows a schematic of a current state corresponding to the six-phase space vector of fig. 2A, the two diagrams show that any one vector corresponding to the switch tube of the inverter is synthesized by two adjacent vectors, specifically one vector Vref is synthesized by two adjacent vectors V100 and V110 shown in FIG. 2A, the waveforms of the two vectors corresponding to the inverted PWM when they are combined are shown in fig. 3, where SA, SB, SC show the inverted PWM waveform diagrams of the switching tubes of the three legs of the inverter, which is a combination of two adjacent vectors V100 and V110, plus a zero vector, to form these three PWM waveforms, and correspondingly outputting three values to the triangular wave where the carrier wave of the PWM wave is positioned for comparison so as to generate the three paths of PWM waveforms. Specifically, a device shown as 5B may be provided inside the controller, wherein the device includes a symmetrical triangular wave generator such as a symmetrical triangular wave generator of 6KHZ, and three comparators, wherein duty ratio values DutyA, DutyB, and DutyC corresponding to three PWM waveforms are respectively input to one input end of the three comparators, and a carrier wave output by the symmetrical triangular wave generator is respectively input to the other input end of the three comparators, and the three paths of PWM waveforms TA, TB, and TC are respectively output from output ends of the three comparators, that is, corresponding to SA, SB, and SC in fig. 3. The inversion PWM sequence is formed by the start and end moments of the high level and the low level in the three PWM waveforms.
The generation of the power factor PWM signal of the switch tube of the PFC circuit is S in FIG. 3IGBTAnd (4) waveform. Fig. 4 shows an equivalent circuit diagram of a topology circuit of the PFC circuit in the on and off states of the switching tube, i.e. the IGBT, as can be seen from fig. 4, when the IGBT of the PFC module 7 is on, the power supply charges the inductor of the PFC module 7, the capacitor of the filter module 9 discharges the inverter, the capacitor current is equal to the dc side current of the inverter, and when the IGBT is off, the capacitor current is equal to the difference between the inductor current and the dc side current. Namely, the following formula holds:
Figure BDA0002937425170000091
wherein IC is electrolytic capacitor current, IDCIs the DC side current of the inverter, IL is the inductive current, I is when the IGBT is ON, i.e. is turned ONFRDI.e. the current of the inverter diode is zero, while the IGBT is OFF, i.e. OFFFRDEqual to IL.
Further, the relationship between the dc-side current IDC and the switching state S, which can be combined with the switching states of the switching tubes of the inverter of fig. 2B, is as follows:
Figure BDA0002937425170000101
wherein IA, IB and IC are currents output by the inverter to A, B, C phase windings of the motor.
The two formulas are combined to obtain:
IC＝IFRD-IDC
＝(1-SIGBT)IL-[-(1-SA)IA-(1-SB)IB-(1-SC)IC]
wherein SIGBTA 1 indicates that the IGBT is on, and 0 indicates off; sxAnd (x is A/B/C) represents the state of the switching tube on the x-phase bridge arm of the inverter, wherein the switching tube on the phase is turned on when the value is 1, and the switching tube on the phase is turned off when the value is 0.
From the above formula, the electrolytic capacitor current IC and the IGBT switch state SIGBTInductor current IL and switching state S of inverterxIn this regard, the inductor current IL is generally associated with the PFC algorithm and is generally relatively fixed, while the IGBT switch state SIGBTAnd the switching state S of the inverterxThe correspondence relationship between the power factor PWM signal and the inverter PWM signal, i.e. the time sequence of the two signals, of the switching tube of the PFC circuit has a significant influence on the electrolytic capacitor current IC, and the following situations specifically exist:
first, the center of the IGBT is aligned with the center of the inverter PWM, as shown in FIG. 3, the power factor PWM is the PWM signal S of the IGBTIGBTThe center time of the middle conduction time is aligned with the center time of the inversion PWM, namely SA, SB and SC. When the IGBT is turned on, the inverter vector corresponding to the corresponding inverter PWM at this time is 111, i.e., the zero vector at this time, and thus the dc side current IDCIs zero, and the electrolytic capacitor current IC is equal to the direct current side current I when the IGBT is conductedDCTherefore, the electrolytic capacitor current IC is zero at this time. When the IGBT is turned off, the turn-off time period of the IGBT is aligned with the non-zero vectors of the inverter, i.e., the time periods T1 and T2 in fig. 3, and at this time, the capacitor current is the difference between the inductor current and the dc side current, and since the dc side current is positive in the motor electromotive state, the difference is smaller than the inductor current. In this case, as can be seen from fig. 3, the inverter zero vector corresponding to the inverted PWM, that is, the time periods other than T1 and T2 in one cycle of the inverted PWM, completely includes the on time period in the IGBT tube PWM signal.As can be seen from fig. 3, the capacitance current IC mainly consists of zero current during the on period of the IGBT, the inductance current IL during the off period of the IGBT, and the difference between the inductance current IL and the dc side current IDC, so the effective value of the capacitance current IC is certainly smaller than the inductance current IL, and may even be smaller than the current of the motor winding.
Secondly, at the turn-off time of the IGBT, the power factor PWM is aligned with the center of the inverter PWM, as shown in FIG. 7, namely the PWM signal S of the IGBT tubeIGBTThe center time of the middle off time is aligned with the center time of the inversion PWM, i.e., SA, SB, SC. When the IGBT is turned on, the inverter vector corresponding to the corresponding inverter PWM at this time is a non-zero vector, i.e., time periods T1 and T2 in fig. 7, and the capacitance current IC is equal to the inverted dc side current IDC at this time, and is a negative value; when the IGBT is turned off, the inverter vector corresponding to the inverter PWM corresponding to the time period of the turn-off time is 111, that is, the inverter vector is a zero vector at this time, so the dc side current IDC is zero, and the capacitive current IC at this time is equal to the inductive current IL. As can be seen from fig. 7, the non-zero vector of the inverter corresponding to the inverted PWM completely corresponds to the on-time of the PWM signal of the IGBT. As can be seen from fig. 7, the capacitance current IC is mainly composed of the inductance current IL and the inverse dc side current IDC, and the effective value of the capacitance current IC is the superposition of the inductance current IL and the motor winding current, so that the current magnitude thereof is significantly larger than that in the first case.
Thirdly, the central moment of the IGBT conduction is between the inversion PWM center and the inversion PWM starting moment, and the IGBT tube PWM signal S is at the momentIGBTThe part of the time of the on-state coincides with the zero vector of the inverted PWM. The magnitude of the effective value of its capacitive current IC lies between the first and second cases.
From the above analysis, the capacitance current IC is the largest in the second case, and the capacitance current IC is smaller in the first and third cases, so if the inverter and the PFC circuit operate in the first and third modes, the venturi current on the filter module can be effectively reduced, and the reduction of the capacitance current enables the electrolytic capacitor in the filter module with smaller capacity to be adopted, so that the volume of the electrolytic capacitor occupies a larger space of the whole control circuit, and the reduction of the volume of the electrolytic capacitor can effectively reduce the volume of the control circuit board, thereby reducing the cost.
In the first and third modes, the zero vector of the inverse PWM is at least partially overlapped with the on time of the IGBT in the power factor PWM, wherein the whole on time of the IGBT in the first condition is within the zero vector time of the inverse PWM. Therefore, after the time sequence of the inverter PWM is obtained, the time sequence of the power factor PWM can be determined according to the time sequence, so that at least part of the zero vector of the inverter PWM is overlapped with the conduction time of the IGBT in the power factor PWM, the capacitance current IC is reduced, the size and the cost of the control circuit board are finally reduced, and the working reliability of the whole control circuit is improved.
And the switching tube of the PFC circuit corresponding to the determined time sequence of the power factor PWM works to control the operation of the PFC circuit, so that the working reliability of the PFC circuit is improved.
The specific process of generating the duty ratio value in the power factor PWM of the switching tube is the prior art, and a single-cycle or average current scheme may be referred to, which is not described in detail herein, and the generation method of the duty ratio corresponding to the inverter PWM is also the prior art, which is not described in detail herein.
According to the control device for driving the motor, the controller obtains the inversion PWM time sequence of the three bridge arms of the inverter, determines the power factor PWM time sequence of the switch tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the control switch tube in the power factor PWM signal is at least partially overlapped with the time of the inverter zero vector corresponding to the inversion PWM, and finally controls the switch tube of the PFC circuit to work according to the power factor PWM corresponding to the power factor PWM time sequence, so that the size of the capacitor current can be effectively reduced, the size of the filtered electrolytic capacitor in the filter module is reduced, the size of the whole control circuit board is reduced, and the cost is reduced.
In some embodiments of the present invention, when determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing, the controller is further configured to:
generating a sawtooth carrier corresponding to the power factor PWM, wherein the period of the sawtooth carrier is half of a triangular carrier of the inverter PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as that of the triangular carrier;
comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a first power factor PWM;
comparing the value corresponding to the remaining half of the duty ratio of a switching tube of the PFC circuit with a sawtooth carrier to generate a second power factor PWM;
and synthesizing the first PWM signal and the second PWM signal into power factor PWM.
As shown in fig. 3, the period of the sawtooth wave carrier IGBT CNT is half of the triangular carrier PWM CNT of the inverter PWM of the three-way bridge arm, and the start time of the sawtooth wave carrier IGBT CNT is the same as the start time of the triangular carrier PWM CNT, or the start time of the sawtooth wave carrier IGBT CNT is the same as the peak time of the triangular carrier PWM CNT. As shown in fig. 3, the start time of the first sawtooth wave carrier IGBT CNT from the left is the same as the start time of the triangular carrier PWM CNT, and the start time of the second sawtooth wave carrier IGBT CNT is the same as the peak time of the triangular carrier PWM CNT.
Dividing the value Duty corresponding to the Duty ratio of the switching tube, i.e. IGBT, of the PFC circuit into two parts, wherein half Duty/2 is compared with the sawtooth wave carrier IGBT CNT to generate a first power factor PWM, such as the PWM waveform S1 in FIG. 3IGBT。
Comparing the remaining half 1-Duty/2 of the value Duty corresponding to the Duty ratio of the switching tube, i.e. IGBT, of the PFC circuit with the sawtooth wave carrier IGBT CNT to generate a second power factor PWM, such as the PWM waveform S2 in FIG. 3IGBT。
And superposing the first PWM signal and the second PWM signal to synthesize the power factor PWM, wherein the two signals are superposed through an OR gate to finally generate the PWM with the power factor higher than the power factor. As shown in fig. 5A, the Duty ratio of the IGBT is divided into two paths, one of the two paths is divided into Duty/2 by the 1/2 proportional amplifier, and the Duty/2 and the triangular carrier PWM CNT are sent to the first comparator to generate the first power factor PWM, the other path is divided into the remaining 1-Duty/2 by the subtractor, and the remaining 1-Duty/2 and the triangular carrier PWM CNT are sent to the second comparator to generate the second power factor PWM, and finally the two paths of power factor PWM are sent to the or gate to be superimposed to generate the power factor PWM signal.
The conducting center time in the generated power factor PWM is aligned with the center of the inverter PWM, so that when an IGBT switching tube for controlling PFC works, the current of an electrolytic capacitor current IC is minimum, the capacity of the electrolytic capacitor is effectively reduced, the volume of the whole control circuit board is reduced, and the cost is reduced.
The invention also provides a control circuit for the inverter air conditioner, as shown in fig. 1, the control circuit comprises a rectifying module 8, a PFC module 7 and a filtering module 9, and the control circuit further comprises the control device for driving the motor mentioned in the above embodiment.
The rectifying module 8 is used for rectifying alternating current input into the motor driving circuit and outputting pulsating direct current;
the PFC module 7 is connected with a rectification module 8 to correct the power factor of the pulsating direct current;
the filtering module 9 is connected with the PFC module 7 and used for filtering the direct current output by the FC module and outputting smooth direct current, and the filtering module 9 is connected with the direct current bus and supplies power to the control device through the direct current bus.
The invention further provides a variable frequency air conditioner which is provided with the control circuit for the variable frequency air conditioner.
In the description herein, references to the description of the term "one embodiment," "some embodiments," "an example," "a specific example," or "some examples," etc., mean that a particular feature, structure, material, or characteristic described in connection with the embodiment or example is included in at least one embodiment or example of the invention. In this specification, the schematic representations of the terms used above do not necessarily refer to the same embodiment or example. Furthermore, the particular features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.
In the description of the present invention, it is to be understood that the terms "central," "longitudinal," "lateral," "length," "width," "thickness," "upper," "lower," "front," "rear," "left," "right," "vertical," "horizontal," "top," "bottom," "inner," "outer," "clockwise," "counterclockwise," "axial," "radial," "circumferential," and the like are used in the orientations and positional relationships indicated in the drawings for convenience in describing the invention and to simplify the description, and are not intended to indicate or imply that the referenced devices or elements must have a particular orientation, be constructed and operated in a particular orientation, and are therefore not to be considered limiting of the invention.
Furthermore, the terms "first", "second" and "first" are used for descriptive purposes only and are not to be construed as indicating or implying relative importance or implicitly indicating the number of technical features indicated. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one such feature. In the description of the present invention, "a plurality" means at least two, e.g., two, three, etc., unless specifically limited otherwise.
In the present invention, unless otherwise expressly stated or limited, the terms "mounted," "connected," "secured," and the like are to be construed broadly and can, for example, be fixedly connected, detachably connected, or integrally formed; can be mechanically or electrically connected; they may be directly connected or indirectly connected through intervening media, or they may be connected internally or in any other suitable relationship, unless expressly stated otherwise. The specific meanings of the above terms in the present invention can be understood by those skilled in the art according to specific situations.
In the present invention, unless otherwise expressly stated or limited, the first feature "on" or "under" the second feature may be directly contacting the first and second features or indirectly contacting the first and second features through an intermediate. Also, a first feature "on," "over," and "above" a second feature may be directly or diagonally above the second feature, or may simply indicate that the first feature is at a higher level than the second feature. A first feature being "under," "below," and "beneath" a second feature may be directly under or obliquely under the first feature, or may simply mean that the first feature is at a lesser elevation than the second feature.
Although embodiments of the present invention have been shown and described above, it is understood that the above embodiments are exemplary and should not be construed as limiting the present invention, and that variations, modifications, substitutions and alterations can be made to the above embodiments by those of ordinary skill in the art within the scope of the present invention.

Claims (8)

1. A control method for a motor drive including a PFC circuit, a controller, and an inverter, characterized by comprising:
acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of the inverter;
determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in a control switching tube in a power factor PWM signal is at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM;
and controlling the switch tube of the PFC circuit to work according to the power factor PWM corresponding to the power factor PWM time sequence.
2. The control method for motor drive according to claim 1, wherein a center timing of a conduction time of a switching tube of the PFC circuit is aligned with a center timing of the inverter PWM.
3. The control method for motor drive according to claim 1, wherein the determining the power factor PWM timing of the switching tube of the PFC circuit according to the inverted PWM timing comprises:
generating a sawtooth carrier corresponding to the power factor PWM, wherein the period of the sawtooth carrier is half of a triangular carrier of an inverted PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as the starting time or the peak time of the triangular carrier;
comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with the sawtooth carrier to generate a first power factor PWM;
comparing the value corresponding to the remaining half of the duty ratio of the switching tube of the PFC circuit with the sawtooth carrier to generate a second power factor PWM;
and synthesizing the first PWM signal and the second PWM signal into power factor PWM.
4. A control device for a motor drive, the control device comprising:
the PFC module is used for correcting the power factor of the input pulsating direct current to generate smooth direct current;
the inverter is used for converting the input direct current into three-phase alternating current so as to drive the motor to operate;
the controller is configured to:
acquiring an inversion PWM (pulse width modulation) time sequence of three bridge arms of the inverter;
determining a power factor PWM (pulse width modulation) time sequence of a switching tube of the PFC circuit according to the inversion PWM time sequence, wherein the conduction time of the switching tube in a control switching tube in a power factor PWM signal is at least partially overlapped with the time of an inverter zero vector corresponding to the inversion PWM;
and controlling the PFC module to operate according to the power factor PWM corresponding to the power factor PWM time sequence.
5. The control device of claim 4, wherein a center time of a conduction time of a switching tube of the PFC circuit is aligned with a center time of the inverter PWM.
6. The control apparatus of claim 4, wherein when determining the power factor PWM timing for the switching tubes of the PFC circuit according to the inverted PWM timing, the controller is further configured to:
generating a sawtooth carrier corresponding to the power factor PWM, wherein the period of the sawtooth carrier is half of a triangular carrier of an inverted PWM of the three bridge arms, and the starting time of the sawtooth carrier is the same as that of the triangular carrier;
comparing a value corresponding to half of the duty ratio of a switching tube of the PFC circuit with the sawtooth carrier to generate a first power factor PWM;
comparing the value corresponding to the remaining half of the duty ratio of the switching tube of the PFC circuit with the sawtooth carrier to generate a second power factor PWM;
and synthesizing the first PWM signal and the second PWM signal into power factor PWM.
7. A control circuit for a variable frequency air conditioner, the control circuit comprises a rectification module and a filtering module, and is characterized by further comprising the control device for the motor drive of any one of claims 4 to 6;
the rectification module is used for rectifying alternating current input into the motor driving circuit and outputting pulsating direct current;
the filtering module is connected with the PFC equipment and used for filtering the direct current output by the PFC equipment and outputting smooth direct current, and the filtering module is connected with a direct current bus and supplies power to the control device through the direct current bus.
8. An inverter air conditioner, characterized in that the inverter air conditioner is provided with the control circuit for the inverter air conditioner as claimed in claim 7.
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