CN109936318B - Optimization method for reducing electromagnetic loss of motor - Google Patents
Optimization method for reducing electromagnetic loss of motor Download PDFInfo
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- CN109936318B CN109936318B CN201910333145.6A CN201910333145A CN109936318B CN 109936318 B CN109936318 B CN 109936318B CN 201910333145 A CN201910333145 A CN 201910333145A CN 109936318 B CN109936318 B CN 109936318B
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- 238000000034 method Methods 0.000 title claims abstract description 27
- 238000005457 optimization Methods 0.000 title claims abstract description 15
- 230000000694 effects Effects 0.000 claims abstract description 21
- 238000004364 calculation method Methods 0.000 claims abstract description 16
- 230000004907 flux Effects 0.000 claims description 50
- XEEYBQQBJWHFJM-UHFFFAOYSA-N Iron Chemical compound [Fe] XEEYBQQBJWHFJM-UHFFFAOYSA-N 0.000 claims description 34
- 238000004804 winding Methods 0.000 claims description 30
- 229910052742 iron Inorganic materials 0.000 claims description 17
- RYGMFSIKBFXOCR-UHFFFAOYSA-N Copper Chemical compound [Cu] RYGMFSIKBFXOCR-UHFFFAOYSA-N 0.000 claims description 13
- 229910052802 copper Inorganic materials 0.000 claims description 13
- 239000010949 copper Substances 0.000 claims description 13
- 238000004088 simulation Methods 0.000 claims description 5
- 238000006880 cross-coupling reaction Methods 0.000 claims description 4
- 238000004458 analytical method Methods 0.000 claims description 3
- 230000035699 permeability Effects 0.000 claims description 3
- 229910000976 Electrical steel Inorganic materials 0.000 claims description 2
- 230000004323 axial length Effects 0.000 claims description 2
- 230000005284 excitation Effects 0.000 claims description 2
- 230000005415 magnetization Effects 0.000 claims description 2
- 230000003313 weakening effect Effects 0.000 claims description 2
- 230000005389 magnetism Effects 0.000 claims 1
- 230000008859 change Effects 0.000 description 11
- 230000009467 reduction Effects 0.000 description 5
- 230000001360 synchronised effect Effects 0.000 description 4
- 230000005672 electromagnetic field Effects 0.000 description 3
- 230000006872 improvement Effects 0.000 description 2
- 230000008569 process Effects 0.000 description 2
- 238000011160 research Methods 0.000 description 2
- 230000001133 acceleration Effects 0.000 description 1
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- 239000004020 conductor Substances 0.000 description 1
- 238000011217 control strategy Methods 0.000 description 1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/02—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for optimising the efficiency at low load
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
- H02P21/28—Stator flux based control
- H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
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- Engineering & Computer Science (AREA)
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- Control Of Ac Motors In General (AREA)
- Iron Core Of Rotating Electric Machines (AREA)
Abstract
The invention provides an optimization method for reducing the electromagnetic loss of a motor, which establishes a calculation model of the electromagnetic loss of the motor by considering an SVPWM control model under a magnetic saturation effect, thereby obtaining the optimization method for reducing the electromagnetic loss in a low-speed and large-torque area of the motor, providing a theoretical basis for improving the peak torque of the motor, and realizing a plurality of effects which are not achieved in the prior art.
Description
Technical Field
The application relates to the field of motor optimization design, in particular to an optimization method for reducing electromagnetic loss of a motor.
Background
The peak torque of the motor directly influences the acceleration capability of the motor and plays a vital role in the dynamic performance of the motor. The generation of the electromagnetic loss of the motor is closely related to the interaction of an electromagnetic field inside the motor, and is an important factor for restricting the peak torque of the motor. The electromagnetic loss of the motor mainly comprises winding copper loss, stator iron loss and rotor eddy current loss, and the eddy current loss in the rotor iron loss is far greater than the magnetic hysteresis loss, so that the rotor iron loss and the permanent magnet eddy current loss are called as rotor eddy current loss together.
At present, SVPWM control is widely applied to motor control technology. The motor control generates a large amount of current harmonics, so that electromagnetic field harmonic components are increased, and electromagnetic loss is increased. Compared with a traditional power frequency motor, the wide speed regulation range of the vehicle motor deepens the influence of a motor control mode on iron loss, so that the electromagnetic loss under the SVPWM control of the motor is necessary to be calculated, the magnetic saturation effect of a motor stator causes the nonlinear change of electromagnetic parameters, particularly in a low-speed large-torque area, the magnetic saturation effect is more serious, the current harmonic generated by the motor in the SVPWM control process is changed, and the electromagnetic loss generated by the motor is further influenced. Meanwhile, the electromagnetic loss of a low-speed large-torque area of the motor is high, so that the problem of temperature rise of the motor is obvious, and the improvement of the peak torque of the motor is limited.
Therefore, how to provide an optimization method for reducing the electromagnetic loss in the low-speed large-torque region of the motor on the basis of considering the SVPWM control model of the motor under the magnetic saturation effect provides a theoretical basis for improving the peak torque of the motor, and is a problem to be solved in the field.
Disclosure of Invention
Aiming at the technical problems in the prior art, the invention provides an optimization method for reducing the electromagnetic loss of a motor, which specifically comprises the following steps:
step one, dq axis flux linkage data of the motor under different currents are obtained, and a motor voltage, flux linkage and electromagnetic torque equation considering a magnetic saturation effect is established according to the relation between the dq axis flux linkage and the currents.
And step two, establishing a motor SVPWM control model under the nonlinear flux linkage according to the equation established in the step, and obtaining the phase current of the motor under the working condition of low speed and large torque.
And step three, establishing an electromagnetic loss calculation model of the motor under SVPWM control in consideration of the magnetic saturation effect based on the phase current obtained in the step two.
And step four, analyzing the influence of the tooth width of the motor stator and the winding number of the conducting wires on the electromagnetic loss under the working condition of low speed and large torque by using the electromagnetic loss calculation model established in the step three, and implementing an optimization strategy for reducing the electromagnetic loss.
Further, the dq-axis flux linkage data obtained in the first step is obtained by performing nonlinear simulation on the dq-axis flux linkage of the motor under different currents based on a frozen permeability method.
Further, the dq-axis flux linkage in the first step has the following fitting relationship with the current:
wherein,
respectively d and q axis flux linkage, KLd、KsdTo account for the curve shape coefficient of the d-axis total flux linkage under the effect of magnetic saturation, KsqdIs the cross-coupling influence coefficient of q-axis current on d-axis total flux linkage, KLq、KsqTo account for the curve shape coefficient of the total flux linkage of the q axis under the effect of magnetic saturation, KsdqIs the cross-coupling coefficient of influence of the d-axis current on the q-axis total flux linkage,
for a fixed value of d-axis flux linkage at different q-axis currents, I0D-axis current, i, corresponding to a fixed d-axis flux linkaged、iqD-axis and q-axis currents, respectively.
Further, the motor voltage, flux linkage and electromagnetic torque equation considering the magnetic saturation effect established in the first step specifically includes:
voltage equation:
the flux linkage equation:
electromagnetic torque equation:
in the formula, ω is an electrical angular velocity, vd、vqAre d and q axis voltages, RsP is the number of pole pairs for phase resistance.
Further, in the motor SVPWM control model in the second step, maximum torque current ratio control and field weakening control are respectively adopted according to working conditions, and the motor stator dq axis current obtained according to the flux linkage equation is:
the motor SVPWM system mainly comprises a control module and a motor module, wherein the motor control module calculates a command value of the dq axis voltage of the motor through a rotating speed command value, an actual rotating speed and the dq axis current, and obtains a control signal of an IGBT switch of the inverter through SVPWM modulation. The motor module obtains the working state of the motor according to the actual voltage and the actual rotating speed of the dq axis by using the mechanical and electrical characteristics of the motor module, and stator phase currents at different working points required by the analysis of the iron loss of the motor stator are obtained.
Further, the establishing the electromagnetic loss calculation model in the third step specifically includes:
the copper loss of the winding is calculated based on the harmonic current of each time of the motor:
wherein m is the number of motor phases, IkIs the effective value of the kth harmonic current, RdcIs a winding phase resistance;
the method comprises the steps of establishing a finite element model of the motor according to the geometric structure and the size of the motor, dividing a motor stator into N subdivision units, obtaining radial tangential magnetic flux density of each subdivision unit of the stator under the common excitation of a permanent magnet and phase current through finite element simulation, obtaining stator iron loss of the motor by utilizing a segmented variable coefficient iron loss calculation model of the motor, and simultaneously referring to a stator iron loss calculation method by a rotor iron loss calculation method. The calculation model of the stator iron loss is as follows:
wherein, PFeFor stator core losses, PhFor hysteresis loss, PeFor eddy current losses, PecxTo add losses, khα is a hysteresis loss coefficient, keIs the eddy current loss coefficient, kecxTo add a loss factor, kh、ke、kecxα are the loss coefficient of classical loss separation model, and can be obtained by fitting measured loss data, wherein f is the motor frequency, and k isrTo the spin magnetization loss coefficient, BnkIs the k-th harmonic amplitude of magnetic flux density, n1kBnk β1kAdding a low order term of flux density, n, to hysteresis losses2kBnk β2kAdding a higher order term of magnetic flux density, n, to the eddy current losses1k、β1k、n2k、β2kFor additional loss factor, L can be obtained by fitting the measured orthogonal loss data at different frequenciesaIs the axial length of the stator, and rho is the density of the silicon steel sheets of the stator, Ph (i)、Pe (i)、Pecx (i)Hysteresis, eddy current and additional loss density, delta, of the ith stator unit, respectivelys (i)Is the area of the ith cell, NsThe number of the subdivision units.
According to the phase current of the motor at the working point of low speed and large torque, the eddy current loss of the permanent magnet is accurately calculated by a three-dimensional time-step finite element method:
where J is the current density, σ is the conductivity, and V is the volume of the permanent magnet block.
Further, the electromagnetic loss under the working condition of low speed and large torque in the fourth step is optimized, including the appropriate reduction of the tooth width and the increase of the number of the parallel wires in the stator slot.
According to the method provided by the invention, the SVPWM control model under the magnetic saturation effect is considered, and the calculation model of the electromagnetic loss of the motor is established, so that the optimization method for reducing the electromagnetic loss in the low-speed large-torque area of the motor is obtained, a theoretical basis is provided for the improvement of the peak torque of the motor, and a plurality of effects which are not achieved in the prior art are realized.
Drawings
FIG. 1 is a flow chart of a method provided by the present invention
FIG. 2 is a graph of the total flux linkage of the dq axis of the motor as a function of the current in the dq axis
FIG. 3 is a motor correction module in an SVPWM control model under consideration of magnetic saturation effect of a motor
FIG. 4 is a stator structure view of an interior permanent magnet synchronous motor
FIG. 5 is a phase current distribution of different tooth width structures of the interior permanent magnet synchronous motor at the peak working point
FIG. 6 shows electromagnetic loss distribution of an interior permanent magnet synchronous motor before optimization under different tooth widths
FIG. 7 shows the optimized electromagnetic loss distribution of the PMSM with different tooth widths
Detailed Description
The following describes an optimized method for reducing electromagnetic loss in a low-speed and high-torque region of a motor according to the present invention in detail with reference to the accompanying drawings.
As shown in fig. 1, the method provided by the present invention specifically includes the following steps:
step one, dq axis flux linkage data of the motor under different currents are obtained, and a motor voltage, flux linkage and electromagnetic torque equation considering a magnetic saturation effect is established according to the relation between the dq axis flux linkage and the currents.
And step two, establishing a motor SVPWM control model under the nonlinear flux linkage according to the equation established in the step, and obtaining the phase current of the motor under the working condition of low speed and large torque.
And step three, establishing an electromagnetic loss calculation model of the motor under SVPWM control in consideration of the magnetic saturation effect based on the phase current obtained in the step two.
And step four, analyzing the influence of the tooth width of the motor stator and the winding number of the conducting wires on the electromagnetic loss under the working condition of low speed and large torque by using the electromagnetic loss calculation model established in the step three, and implementing an optimization strategy for reducing the electromagnetic loss.
In a preferred embodiment of the present application, the dq-axis flux linkage data can be obtained by performing a nonlinear simulation on dq-axis flux linkages of the motor at different currents based on a frozen permeability method, and a fitting relationship between the flux linkages and the currents is shown in table 1 and fig. 2.
TABLE 1
The motor module in the SVPWM control model taking the magnetic saturation effect into consideration of the motor is modified as shown in FIG. 3, and the control strategy is kept unchanged. And obtaining the stator phase current of the motor at a low-speed large-torque working condition point according to the SVPWM model.
In a preferred embodiment of the present application, a 370kW interior permanent magnet synchronous motor is taken as a research object, the stator structure of the motor is shown in fig. 4, and the influence of the tooth width and winding parameters of the stator on the phase current, electromagnetic field and electromagnetic loss of the motor is studied by taking the peak operating point (1000rpm, 1660Nm) as an example. When analyzing the tooth width, the tooth width is set to be 4-10 mm, and the height h of the motor yoke is ensuredjConstant and stator bottom and top tooth width ratio bt1/bt2And is not changed. Based on the principle, stator slot parameters of the motor under different tooth widths can be obtained. And then, obtaining the phase currents of the motors with different tooth widths at the peak working point according to the SVPWM control model, as shown in FIG. 5. The inductance and flux linkage parameters of the motor change along with the change of the tooth width, so that the phase current difference of the motor with different tooth widths at the peak working point is larger. Under the condition of ensuring that winding parameters are not changed, the smaller the tooth width is, the larger the phase current is needed under the same peak working point.
According to the phase currents of the motors with different tooth widths at the peak working point, the electromagnetic losses of the motors with different tooth widths are respectively calculated by using formulas about all losses in an electromagnetic loss calculation model, as shown in fig. 6. It can be seen from the figure that the total loss of the motor increases with the reduction of the tooth width, wherein the copper loss accounts for a large proportion of the total loss, and when the tooth width is smaller, the higher phase current causes the copper loss of the motor to be larger, and the variation range of the stator iron loss and the rotor eddy current loss under the same load working condition is not large. Meanwhile, when the tooth width is small, the slot filling rate is reduced, and at the moment, enough stator slot space can be provided for optimizing winding parameters, so that the reduction of copper loss is realized, and the reduction of the electromagnetic loss of the motor is realized.
In order to research the influence of winding parameters on the electromagnetic loss of the motor, the optimization of the winding parameters in the stator slot is realized by changing the number of winding turns of the lead and the number of winding roots, the number of winding turns and the wire diameter of the motor are ensured to be unchanged in the analysis process, and the influence of the voltage drop on phase resistance and the change of the number of winding roots of the lead on induced electromotive force is ignored. Wound aroundThe change of the number of the parallel winding of the group of conducting wires does not affect the phase current, but the change of the number of the parallel winding of the conducting wires changes the sectional area of the winding and is in inverse proportion to the phase resistance, and the phase resistance is correspondingly reduced along with the increase of the number of the parallel winding of the conducting wires. Therefore, the change of the number of the conducting wires in the winding can change the copper loss of the winding, but the influence on the iron loss of the stator and the eddy current loss of the rotor can be ignored, and the total loss of the motor is influenced. Suppose the number of parallel windings of the conductor is Nt1Phase resistance R and stator slot fullness Sf1Now change the number of parallel winding of the wire to Nt2When the stator phase resistance R' is equal to (N)t1/Nt2) R, tank fullness Sf2＝(Nt2/Nt1)Sf1The winding copper loss of the machine can be expressed as:
according to the formula, the change curve of the winding copper loss with the tooth width under different stator slot filling ratios of the motor can be obtained, and on the basis, the change curve of the total loss with the tooth width under different slot filling ratios of the motor is obtained, as shown in fig. 7. The stator tooth width is properly reduced for a motor with a certain inner diameter and outer diameter, and the number of the conducting wires wound around the motor can be increased due to enough space in the stator slot, so that the phase resistance is reduced, the copper consumption of the winding is reduced, and the total loss of the motor in a low-speed large-torque area is reduced. In the same stator slot, the number of the conducting wires in parallel winding is increased, so that the slot fullness rate is increased, the phase resistance is reduced, the copper loss and the total loss of the motor are reduced, but the stator slot fullness rate is not too large for the convenience of wire embedding. Under the same stator slot filling rate, the stator tooth width influences the loss of the motor at the peak working point. With the reduction of the tooth width, the larger stator slot space enables the number of the parallel wound wires to be increased, the phase resistance is reduced, and the copper loss is reduced. Therefore, the tooth width of the stator is properly reduced, the number of the conducting wires wound around the winding is increased, and the copper loss and the total loss of the motor in a low-speed large-torque area can be reduced.
Although embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that changes, modifications, substitutions and alterations can be made in these embodiments without departing from the principles and spirit of the invention, the scope of which is defined in the appended claims and their equivalents.
Claims (6)
1. An optimization method for reducing electromagnetic loss of a motor is characterized by comprising the following steps: the method specifically comprises the following steps:
step one, dq axis flux linkage data of a motor under different currents are obtained, and a motor voltage, flux linkage and electromagnetic torque equation considering a magnetic saturation effect is established according to the relation between the dq axis flux linkage and the currents; the dq-axis magnetic linkage and the current have the following fitting relation:
wherein,
respectively d and q axis flux linkage, KLd、KsdTo account for the curve shape coefficient of the d-axis total flux linkage under the effect of magnetic saturation, KsqdIs the cross-coupling influence coefficient of q-axis current on d-axis total flux linkage, KLq、KsqTo account for the curve shape coefficient of the total flux linkage of the q axis under the effect of magnetic saturation, KsdqIs the cross-coupling coefficient of influence of the d-axis current on the q-axis total flux linkage,
for a fixed value of d-axis flux linkage at different q-axis currents, I0D-axis current, i, corresponding to a fixed d-axis flux linkaged、iqD and q axis currents respectively;
step two, establishing a motor SVPWM control model under a nonlinear flux linkage according to the equation established in the step one to obtain phase current of the motor under a low-speed large-torque working condition;
step three, establishing an electromagnetic loss calculation model of the motor under SVPWM control in consideration of a magnetic saturation effect based on the phase current obtained in the step two;
and step four, analyzing the influence of the tooth width of the motor stator and the winding number of the conducting wires on the electromagnetic loss under the working condition of low speed and large torque by using the electromagnetic loss calculation model established in the step three, and implementing an optimization strategy for reducing the electromagnetic loss.
2. The method of claim 1, wherein: and performing nonlinear simulation on the dq-axis flux linkage of the motor under different currents based on the frozen permeability method to obtain the dq-axis flux linkage data obtained in the first step.
3. The method of claim 2, wherein: the motor voltage, flux linkage and electromagnetic torque equation which is established in the first step and takes the magnetic saturation effect into consideration specifically comprises the following steps:
voltage equation:
the flux linkage equation:
electromagnetic torque equation:
in the formula, ω is an electrical angular velocity, vd、vqAre d and q axis voltages, RsP is the number of pole pairs for phase resistance.
4. The method of claim 3, wherein: in the motor SVPWM control model in the second step, maximum torque current ratio control and flux weakening control are respectively adopted according to working conditions, and based on the flux linkage equation, the current of a dq axis of a motor stator can be obtained as follows:
and obtaining the working state of the motor according to the actual voltage and the actual rotating speed of the dq axis, and obtaining stator phase currents at different working points required by the analysis of the iron loss of the motor stator.
5. The method of claim 4, wherein: establishing the electromagnetic loss calculation model in the third step specifically comprises the following steps:
the copper loss of the winding is calculated based on the harmonic current of each time of the motor:
wherein m is the number of motor phases, IkIs the effective value of the kth harmonic current, RdcIs a winding phase resistance;
establishing a finite element model of the motor according to the geometric structure and the size of the motor, dividing a motor stator into a plurality of subdivision units, obtaining the radial tangential magnetic flux density of each subdivision unit of the stator under the common excitation of a permanent magnet and phase current through finite element simulation, and obtaining the stator iron loss of the motor by utilizing a segmented variable coefficient iron loss calculation model of the motor as follows:
wherein, PFeFor stator core losses, PhFor hysteresis loss, PeFor eddy current losses, PecxTo add losses, khα is a hysteresis loss coefficient, keIs the eddy current loss coefficient, kecxFor additional loss factor, f is the motor frequency, krTo the spin magnetization loss coefficient, BnkIs the k-th harmonic amplitude of magnetic flux density, n1kBnk β1kAdding a low order term of flux density, n, to hysteresis losses2kBnk β2kAdding magnetism to eddy current lossesFlux density higher order term, LaIs the axial length of the stator, and rho is the density of the silicon steel sheets of the stator, Ph (i)、Pe (i)、Pecx (i)Hysteresis, eddy current and additional loss density, delta, of the ith stator unit, respectivelys (i)Is the area of the ith cell, NsThe number of the subdivision units; calculating the rotor iron loss by referring to the stator iron loss;
according to the phase current of the motor at the working point of low speed and large torque, the eddy current loss of the permanent magnet is accurately calculated by a three-dimensional time-step finite element method:
where J is the current density, σ is the conductivity, and V is the volume of the permanent magnet block.
6. The method of claim 1, wherein: and in the fourth step, the electromagnetic loss is optimized under the working condition of low speed and high torque, including properly reducing the tooth width and increasing the number of the parallel wires in the stator slot.
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