JP4564508B2 - Electric vehicle control device - Google Patents

Description

  The present invention relates to an electric vehicle control device for reducing noise generated from a main motor of an electric vehicle.

  In an inverter that drives a main motor of an electric vehicle, when the output voltage is high during high-speed running, one-pulse control that reduces loss and maximizes the voltage utilization rate of the inverter is employed. This one-pulse control is a voltage fixed mode control that outputs a voltage of one pulse in a half cycle of a sine wave. FIG. 7 shows a conventional electric vehicle control apparatus for an AC train.

  In the figure, the configuration of an electric vehicle control device for an AC train that is fed from an AC overhead line 17 and drives the main motor 3 is shown. The AC overhead line 17 is connected to the rail 21 through the pantograph 18, the primary winding of the main transformer 16, and the wheels 20. A single-phase NPC converter 15 (Neutral Point Clamped Converter) is connected to the secondary winding of the main transformer 16. Two filter capacitors 2 are connected in series on the DC side of the single-phase NPC converter 15 to smooth the DC voltage. An NPC inverter 1 having a three-phase variable voltage variable frequency output is connected to the two series filter capacitors 2, and the main motor 3 is driven and controlled. Here, an induction motor is used as the main motor 3.

  As the motor control method, vector control is applied in which the torque and speed of the motor are controlled by controlling various electric quantities on the DQ axis rotation coordinate system. Currents (phase currents) Iu and Iw flowing through the main motor 3 are detected by the current detector 4 and converted to DQ axis currents Id and Iq by the coordinate converter 7. The voltage calculation unit 5 calculates and outputs the DQ axis output voltage commands Vd * and Vq * so that the excitation current command IdRef and the excitation current Id match the torque current command IqRef and the torque current Iq, respectively. In the coordinate conversion unit 6, the DQ axis output voltage commands Vd * and Vq * are converted into a polar coordinate system. That is, the magnitude V1 * of the DQ axis output voltage command and the phase difference Gamma from the D axis to the output voltage vector are generated and output. The adder 24 adds the phase angle θ from the stationary coordinate system A axis to the rotating coordinate system D axis and the phase difference Gamma, and calculates the phase angle θV from the stationary coordinate system A axis to the output voltage vector.

  The modulation factor calculator 22 receives the output voltage magnitude V1 * generated by the coordinate converter 6 and the DC voltage Vdc detected by the voltage detector 19 to calculate the modulation factor AL0 *. In the limiter 23, the modulation rate AL0 * is limited by the lower limit value 0 and the upper limit value 1, and is output as the corrected modulation rate AL *.

  The slip frequency calculation unit 11 calculates and outputs the slip frequency ωs based on the excitation current command IdRef and the torque current command IqRef. The adder 13 adds the rotation speed ωr of the main motor detected by the speed detector 10 and the slip frequency ωs *, and outputs the inverter output frequency reference ω1 *.

  On the other hand, the DC voltage Vdc of the inverter 1 detected by the voltage detector 19 is input to the beatless control unit 12. The beatless control unit 12 extracts pulsation from the DC voltage, and the inverter output frequency correction amount ω1 cmp is calculated and output in accordance with the pulsation. This beatless control method is described in Japanese Patent Publication No. 7-46918. In the technology described in this publication, since the single-phase AC power supply is converted to DC, the pulsation of the frequency component twice the power supply frequency (100 Hz pulsation when the power supply is 50 Hz) is superimposed on the DC side of the converter / inverter. Is done. The torque pulsation of the electric motor caused by the DC pulsation can be suppressed by the inverter output frequency correction ω1 cmp which is the output of the beatless control.

  In the adder 14, the inverter output frequency reference ω1 * and the inverter output frequency correction amount ω1cmp are added to generate the inverter output frequency ω1. The integrator 8 integrates the inverter output frequency ω1, and generates and outputs the D axis phase angle θ (relative to the reference axis A axis of the AB axis stationary coordinate system) used in the coordinate converters 6 and 7.

Here, the gate command calculation unit 9 generates and outputs a gate command based on the modulation factor AL * and the phase angle θV from the stationary coordinate system A-axis to the output voltage vector so that the fundamental voltage becomes equal. In particular, when the electric vehicle is traveling at high speed and the output voltage is large, the inverter can output the maximum voltage and can perform one-pulse control with low loss.
Japanese Examined Patent Publication No. 61-48356 (beatless control for correcting the modulation rate) Japanese Patent Publication No. 7-46918 (Beatless control to correct the frequency) Japanese Patent No. 3226258 (one pulse control: strictly speaking, voltage fixed control)

  The waveform at the time of one-pulse control in the conventional example is as shown in FIG. This figure shows the U-phase output voltage command and the U-phase output voltage. In FIG. 7, the U-phase output voltage command is not described, but can be converted by multiplying the output of the phase modulation rate command calculation unit in FIG. 2 described later by a predetermined gain (DC voltage of the inverter). Ideally, it is desirable that the U-phase output voltage is output in the same sine wave form as the U-phase output voltage command. In this case, no voltage harmonic or current harmonic is generated.

  However, actually, as shown in FIG. 3B, there is a large difference between the U-phase output voltage command and the U-phase output voltage. In particular, in the vicinity of the zero-cross of the U-phase output voltage command, the difference between the output voltage command and the output voltage is large, and this part is the largest cause of harmonic generation. As a result, extremely large harmonics are generated in the state quantity on the DQ axis rotation coordinate system and the state quantity on the DC input side of the inverter. The frequency of this harmonic is six times the inverter output frequency, and the frequency of the harmonic changes as the inverter output frequency increases or decreases.

  On the other hand, the main motor noise is dominated by 12 times the inverter output frequency. It is considered that this noise has a strong correlation with the fluctuation frequency of the amplitude of the absolute value of the current value, and therefore a 12-fold component that is twice as much as 6 times the inverter output frequency appears greatly. That is, the noise of the main motor is partly due to harmonics when the inverter applies one-pulse control.

  The present invention has been made in view of such a conventional problem, and an object of the present invention is to provide an electric vehicle control device capable of reducing noise generated from a main motor during high-speed traveling.

In order to achieve the above object, the present invention provides a converter connected to an AC overhead line via a main transformer, an inverter connected to the converter for converting a direct current into a three-phase alternating current and outputting it to a main motor , Based on inverter output frequency calculation means for calculating the output frequency of the voltage output from the inverter, inverter output voltage calculation means for calculating the magnitude of the voltage output from the inverter, and the frequency and magnitude of the voltage output from the inverter Gate command calculation means for generating a gate command for the inverter, voltage detection means for detecting a DC side voltage of the inverter, and the voltage detection value of the voltage detection means so that the voltage detection value matches a predetermined DC voltage command value. DC voltage control means for controlling the converter, and the DC side voltage of the inverter at a frequency six times the inverter output frequency. DC pulsating means for moving the frequency, and the DC pulsating means receives the inverter output frequency and calculates a frequency six times that of the inverter output frequency, and a predetermined amplitude and phase shift at the calculated six times the frequency. DC voltage pulsation command calculation means for calculating a DC voltage pulsation command having a quantity, and correction means for correcting the DC voltage command value by the calculated DC voltage pulsation command, each phase constituting the three-phase AC The phase shift amount is adjusted so that the DC voltage decreases near the zero crossing of the output voltage and the DC voltage increases near the peak of each phase output voltage .
The present invention also includes a converter connected to an AC overhead line via a main transformer, an inverter connected to the converter for converting a direct current into a three-phase alternating current and outputting it to a main motor, and a voltage output from the inverter. Inverter output frequency calculation means for calculating the output frequency, inverter output voltage calculation means for calculating the magnitude of the voltage output by the inverter, and a gate command for the inverter based on the frequency and magnitude of the voltage output by the inverter. Detects a voltage on the DC side of the inverter, and an energy storage conversion means comprising a gate command calculation means to be generated, a chopper means connected to the DC side of the inverter, and an energy storage means connected to the other of the chopper means And the voltage detection value of the voltage detection means coincides with a predetermined DC voltage command value. The DC voltage control means for controlling the chopper means constituting the energy storage conversion means, and the DC pulsation means for pulsating the DC side voltage of the inverter at a frequency 6 times the inverter output frequency. Includes a frequency calculation means for inputting the inverter output frequency and calculating a frequency six times that of the inverter, and a DC voltage pulsation for calculating a DC voltage pulsation command having a predetermined amplitude and phase shift amount at the calculated six times the frequency. Command calculation means, and correction means for correcting the DC voltage command value by the calculated DC voltage pulsation command, so that the DC voltage drops near the zero cross of each phase output voltage constituting the three-phase AC. In addition, the phase shift amount is adjusted so that the DC voltage increases near the peak of each phase output voltage.

  According to the present invention, the DC voltage of the inverter can be pulsated at 6 times the inverter frequency. As a result, even if the inverter uses one-pulse control, the output voltage of each phase is close to a sine wave. Thus, the harmonic current that is six times the inverter frequency can be reduced. Thereby, the electric vehicle control apparatus which can reduce the noise which arises from a main motor during high-speed driving | running | working can be provided.

<First Embodiment>
FIG. 1 is a block diagram showing the configuration of the first exemplary embodiment of the present invention. The same components as those in the conventional example shown in FIG.

  The first embodiment shown in FIG. 1 shows the configuration of an electric vehicle control device for an AC train that is fed from an AC overhead line 17 and drives the main motor 3, as in the conventional example shown in FIG. 7.

  That is, the AC overhead line 17 is connected to the rail 21 via the pantograph 18, the primary winding of the main transformer 16, and the wheels 20. A single-phase NPC converter 15 (Neutral Point Clamped Converter) is connected to the secondary winding of the main transformer 16. Two filter capacitors 2 are connected in series on the DC side of the single-phase NPC converter 15 to smooth the DC voltage. An NPC inverter 1 having a three-phase variable voltage variable frequency output is connected to the two series filter capacitors 2, and the main motor 3 is driven and controlled. Here, an induction motor is used as the main motor 3.

As an electric motor control system, vector control for controlling the torque and speed of the electric motor by controlling various electric quantities on the DQ axis rotation coordinate system is applied. Currents Iu and Iw flowing through the main motor 3 are detected by the current detector 4 and converted into DQ axis currents Id and Iq by the coordinate converter 7. In the voltage calculation unit 5, the excitation current command IdRef and the excitation current Id are converted into the torque current command IqRe.
And DQ axis output voltage commands Vd * and Vq * are calculated and output so that the torque current Iq and the torque current Iq coincide with each other. In the coordinate conversion unit 6, the DQ axis output voltage commands Vd * and Vq * are converted into a polar coordinate system. That is, the magnitude V1 * of the DQ axis output voltage command and the phase difference Gamma from the D axis to the output voltage vector are generated and output. The adder 24 adds the phase angle θ from the stationary coordinate system A axis to the rotating coordinate system D axis and the phase difference Gamma, and calculates the phase angle θV from the stationary coordinate system A axis to the output voltage vector.

  The modulation factor calculator 22 receives the output voltage magnitude V1 * generated by the coordinate converter 6 and the DC voltage Vdc detected by the voltage detector 19 to calculate the modulation factor AL0 *. In the limiter 23, the modulation rate AL0 * is limited by the lower limit value 0 and the upper limit value 1, and is output as the corrected modulation rate AL *.

  The limit upper limit value ALmax of the limiter 23 is switched by a modulation factor limit command Flg_AllimCmd.

[Equation 1]
ALmax = 1.0 if Flg_AllimCmd = 0
= β if Flg_AllimCmd = 1
Here, β is a predetermined set value and is defined by β <1.0.

  The modulation factor limit command is generated by the modulation factor limit command calculation unit 31 according to the torque current command IqRef as shown in the following equation.

[Equation 2]
Flg_AllimCmd = 1 if | IqRef | <γ
= 0 if | IqRef |> = γ
Here, γ is a predetermined set value.

  The slip frequency calculation unit 11 calculates and outputs the slip frequency ωs based on the excitation current command IdRef and the torque current command IqRef. The adder 13 adds the rotation speed ωr of the main motor detected by the speed detector 10 and the slip frequency ωs *, and outputs the inverter output frequency reference ω1 *.

  On the other hand, the DC voltage Vdc of the inverter 1 detected by the voltage detector 19 is input to the beatless control unit 12. The beatless control unit 12 extracts pulsation from the DC voltage, and the inverter output frequency correction amount ω1 cmp is calculated and output in accordance with the pulsation. This beatless control method is described in Japanese Patent Publication No. 7-46918. In the technology described in this publication, since the single-phase AC power supply is converted to DC, the pulsation of the frequency component twice the power supply frequency (100 Hz pulsation when the power supply is 50 Hz) is superimposed on the DC side of the converter / inverter. Is done. The torque pulsation of the electric motor caused by the DC pulsation can be suppressed by the inverter output frequency correction ω1 cmp which is the output of the beatless control.

  In the adder 14, the inverter output frequency reference ω1 * and the inverter output frequency correction amount ω1cmp are added to generate the inverter output frequency ω1. The integrator 8 integrates the inverter output frequency ω1, and generates and outputs the D axis phase angle θ (relative to the reference axis A axis of the AB axis stationary coordinate system) used in the coordinate converters 6 and 7.

  As shown in FIG. 2, the gate command calculation unit 9 includes a 1-pulse control unit 25, a 1-dash (hereinafter “1 ′”) pulse control unit 26, a triangular wave comparison PWM control unit 27, a switch 28, The control mode selection unit 29 and the phase modulation rate command calculation unit 30 are provided, and the fundamental voltage is equal based on the input modulation rate AL * and the phase angle θV from the stationary coordinate system A axis to the output voltage vector. A gate command is generated and output as follows.

  There are various PWM control methods for outputting a fundamental wave voltage that matches the output voltage command. Here, triangular wave comparison PWM control, one-dash pulse control, and one-pulse control are employed. Triangular wave comparison PWM control is a well-known technique, and a description thereof will be omitted. One pulse control and one dash pulse control as points will be described below. As will be described later, “1-dash pulse control” is different from 1-pulse control in that a pause interval (zero interval) is provided when the polarity is switched.

  First, FIGS. 3A and 3B show a U-phase output voltage command and a U-phase output voltage. FIG. 5B shows a waveform of one-pulse control. The U-phase gate is turned on with the zero-cross up edge of the U-phase voltage command, and the U-phase gate is turned off with the zero-cross down edge. One-pulse control is the maximum mode that the inverter can output, and there is no parameter that can be arbitrarily adjusted. On the other hand, FIG. 3A shows a waveform of one dash pulse control. The difference from the 1-pulse mode is that there is a section α in which 0 potential is output in the vicinity of the zero cross of the U-phase output voltage. The fundamental wave voltage can be adjusted and controlled by this α.

  In the first embodiment, a gate command to the inverter 1 is generated by switching these three control modes by the control mode selection unit 29 and the switch 28. In the control mode selection unit 29, the modulation rate AL * is input, and the number Pnum corresponding to each control mode is output by the following equation.

[Equation 3]
Pnum = 1 if AL * <x1
= 2 if x1 <= AL * <x2
= 3 if AL * = 1
Here, x1 and x2 are predetermined set values and have a relationship of x1 <x2.

  With the above configuration, when the modulation factor limit flag Flg_AllimCmd is set to 1, the upper limit of the modulation factor AL * is β (defined as β <1.0). Limited. In one-pulse control, a current harmonic that is six times the fundamental frequency is greatly generated and causes noise. In control modes other than this one-pulse control, at least six times larger current than one-pulse control. Harmonics are not generated. Thereby, a reduction in noise can be expected.

  In the first embodiment, the 1-dash control mode is employed as a control mode other than the 1-pulse control, but other control modes may be used.

  However, comparing the case where 1 pulse control is selected with the case where 1 dash pulse control is selected, in order to obtain the same output, the output voltage is 1 dash pulse while the output voltage is larger in the case of 1 pulse control. Control is greater. For this reason, the one-dash pulse control is not preferable because various losses increase. On the other hand, if the output current is constant, the output will decrease as the output voltage is lower.

  Therefore, only when the absolute value of the torque, that is, the torque current command is small, the modulation factor limit command calculation unit 31 sets the modulation factor limit flag Flg_AllimCmd = 1 so as not to use one-pulse control for noise reduction. By setting the modulation factor limit flag Flg_AllimCmd = 0 so that one-pulse control is performed only when the absolute value of the torque current command that needs to be output is large, noise can be partially reduced.

  In addition, regarding the problem that the output is reduced, since a single train is provided with a plurality of the electric vehicle control devices, there is no problem in the system if the output of the overall train is ensured. Therefore, only for electric vehicle control devices installed in vehicles that require high quality, such as Shinkansen green cars, the modulation rate limit flag Flg_AllimCmd = 1 is set for normal cars, and Flg_AllimCmd = 0 is set for the total number of cars in the organization. It is possible to take a trade-off between output reduction and quality improvement.

  Furthermore, there is a beat phenomenon of the main motor as a problem peculiar to the AC train. In the first embodiment, a method is shown in which a beat phenomenon caused by a DC voltage pulsating at twice the power supply frequency in an AC train is suppressed by superimposing a compensation amount of the same frequency on the inverter frequency. . On the other hand, Japanese Patent Publication No. 61-48356 discloses a method of suppressing the beat phenomenon by superimposing a compensation amount on the modulation rate of the inverter. Comparing both effects, the first embodiment only suppresses the beat phenomenon, but the one described in the publication can completely cancel the beat phenomenon in principle. Therefore, even in the conventional control, the latter modulation rate compensation method is adopted in a region where the modulation rate is less than 1. However, in the modulation rate compensation method, since the modulation rate increases or decreases from the average every moment, there is a timing at which a waveform close to one-pulse control that causes noise is output. For this reason, even in the operation in which the modulation rate is limited as in the first embodiment, further noise reduction can be expected by applying beatless control for compensating the frequency.

<Second Embodiment>
FIG. 4 is a block diagram showing the configuration of the second exemplary embodiment of the present invention. Since the configuration of the modulation factor limit command calculation unit 31 is different from that of the first embodiment shown in FIG.

  In the modulation factor limit command calculation unit 31 in the second embodiment, current values Id and Iq detected by the current detector 4 and coordinate-converted by the coordinate conversion unit 7 are input. The modulation factor limit command calculation unit 31 calculates a loss in the motor and predicts the motor temperature based on the loss. Based on this motor temperature, the modulation factor limit flag Flg_AllimCmd is switched. Here, for example, the loss Ploss in the motor can be simply calculated by the following equation.

[Equation 4]
Ploss = R1 × (Id 2 + Iq 2) + (M / L2) 2 × R2 × Iq 2
Here, R1: primary resistance, R2: secondary resistance, M: mutual inductance, L2: secondary self-inductance.

  Further, the predicted temperature value Tmot can be easily predicted by the following equation.

[Equation 5]
Tmot = K / (τ × s + 1) × Ploss + Tmot0
Where K is a conversion factor from heat loss to temperature, τ is a thermal time constant, s is a Laplace operator,
Tmot0 is the initial temperature value.

  The modulation factor limit flag Flg_AllimCmd is determined by the following equation based on the predicted motor temperature value Tmot.

[Equation 6]
Flg_AllimCmd = 1 if Tmot <Tmot_Set
= 0 if Tmot> = Tmot_Set
As described above, whether to use one-pulse control or not to use one-pulse control is determined based on the temperature of the motor. As described above, the one-pulse control has a low loss, but the loss increases in other control modes. Therefore, when the motor temperature is low, a control mode that does not use one-pulse control capable of reducing noise is selected, and when the motor temperature rises, one pulse with less loss is used to suppress the motor temperature rise. Let the control be selected. Therefore, noise can be reduced as much as possible without causing deterioration of reliability such as motor burnout or protection due to temperature rise.

  In the second embodiment, the method of simply predicting the motor loss from the current is shown. However, the temperature may be detected more strictly or by providing a temperature sensor. In addition, since an increase in inverter element temperature, etc., may be considered in addition to a temperature increase on the motor side, the element temperature of the inverter is predicted or detected by the same method, and the modulation rate limit is based on that temperature. The flag may be controlled.

<Third Embodiment>
FIG. 5 is a block diagram showing the configuration of the third exemplary embodiment of the present invention.

  As shown in the figure, the electric vehicle control apparatus of the third embodiment includes an inverter control unit 32 and a converter control unit 33. The inverter control unit 32 is the same as the conventional example shown in FIG. Therefore, the converter control unit 33 will be mainly described.

  The converter control unit 33 includes a zero cross detection unit 36, a phase synchronization control unit 37, an adder 38, a DC pulsation command calculation unit 39, a subtractor 40, a voltage control unit 41, a subtractor 42, and a current control. A unit 43, an adder 44, and a PWM control unit 46 are provided. Further, the converter control unit 33 includes an output current Is which is an AC side current of the converter 15 detected by the current detector 35, a DC voltage Vdc detected by the voltage detector 19 on the output side of the converter 15, and a voltage The primary voltage Vs that is the voltage of the first winding of the main transformer 16 detected by the detector 34 is input. As a detection point of the primary voltage Vs, there is an example in which an electric vehicle includes a third winding in the main transformer 16, and this tertiary voltage is used as an alternative to the primary voltage, that is, the overhead wire voltage Vs. It may be a simple configuration.

  The zero cross detection unit 36 detects the zero cross point based on the primary voltage, that is, the overhead wire voltage Vs. The phase synchronization control unit 37 calculates the overhead line voltage phase θs from the zero cross point of the overhead line voltage. Here, the zero point of the phase θs is defined as the zero cross point of the overhead line voltage.

  The adder 38 adds a DC voltage command VdcRef0 and VdcRefCmp which is an output of a DC pulsation command calculation unit 39 described later to generate a corrected DC pulsation command VdcRef. In the subtractor 40, the DC voltage Vdc between PN detected by the voltage detector 19 is subtracted from the corrected DC pulsation command VdcRef. The voltage controller 41 receives a deviation between the corrected DC pulsation command VdcRef and the detected DC voltage Vdc between the PNs, and the overhead line voltage generated by the phase synchronization controller so that the deviation becomes zero. Based on phase θs, converter current command value IsRef is calculated.

[Equation 7]
IsRef = [(Kp ・ s + Ki) / s] ・ (VdcRef−Vdc) ・ sin (θs)
The current control unit 43 calculates the output voltage correction amount Vccmp as in the following equation so that the detected output current Is matches the output current command value IsRef.

[Equation 8]
Vccmp = Kp ・ (IsRef−Is)
Also, the feedforward term VsFF of the overhead wire voltage is calculated by the power supply voltage FF calculation unit 45 by the following equation. Here, Vs * is a converted value of the rated voltage of the overhead wire to the converter AC side.

[Equation 9]
VsFF = Vs * ・ sin (θs)
The adder 44 can calculate the output voltage command value VcRef of the converter by adding the overhead wire voltage feedforward term VsFF and the output voltage correction amount Vccmp. In the PWM control unit 46, a gate command to the converter 15 is generated by triangular wave comparison PWM control or the like so as to obtain an output voltage that matches the output voltage command value Vc * of the converter.

  Details of the DC pulsation command calculation unit 39, which is a point of the third embodiment, will be described. The DC pulsation command calculation unit 39 receives the inverter output voltage phase angle θV, which is the output of the adder 24 of the inverter control unit 32, and calculates a DC voltage command correction amount VdcRefCmp as shown in the following equation.

[Equation 10]
VdcRefCmp = K2 ・ sin (6 × θv + Δ)
Here, K2 is a predetermined gain, and Δ is a predetermined phase shift amount.

  According to the above configuration, the DC voltage of the inverter can be pulsated in synchronism with the phase angle of the inverter output voltage, six times that is, that is, six times the inverter frequency. By adjusting the amount of phase shift so that the DC voltage decreases near the zero-cross of the U-phase output voltage and the DC voltage increases near the peak of the U-phase output voltage, the inverter controls one pulse. Even if the output voltage of each phase is close to a sine wave, a harmonic current that is six times the inverter frequency can be reduced. Thereby, a reduction in noise can be expected.

<Fourth Embodiment>
FIG. 6 is a block diagram showing the configuration of the fourth exemplary embodiment of the present invention.

  Compared to the conventional example shown in FIG. 7, the inverter control unit 32 is the same, and the energy storage conversion means and its control unit are additionally provided.

  Energy storage conversion means 47 is provided on the DC side of the inverter 1. The energy storage conversion means 47 includes an energy storage unit 51, a reactor 49, and an energy conversion unit 48. As the energy storage unit 51, a large-capacity capacitor, a battery, or the like is used, but a small-capacity capacitor may be used only for the purpose of the fourth embodiment. The energy conversion unit 48 has two switching elements coupled in series. That is, it constitutes a DC chopper circuit.

  As a control unit of the energy storage conversion means 47, there is an energy storage conversion control unit 52. Since this configuration is almost the same as that of the third embodiment shown in FIG. 5, detailed description thereof is omitted. In short, it is important to have a voltage control system in which the DC voltage of the inverter 1 matches the command value. Similarly, the DC pulsation command calculation unit 39 receives the output voltage phase angle θV of the inverter control unit 32 and superimposes a compensation amount of 6 times the frequency on the DC voltage command.

  With the above configuration, as in the third embodiment, the DC voltage of the inverter can be pulsated at a frequency that is six times the frequency of the output voltage. Therefore, even if the inverter uses one-pulse control, the harmonic current that is six times the inverter frequency can be reduced by making the output voltage of each phase close to a sine wave. Thereby, a reduction in noise can be expected.

<Other embodiments>
In addition, the point of 3rd Embodiment and 4th Embodiment is making the DC voltage of an inverter pulsate and making each phase output voltage close to a sine wave even if 1 pulse control is used. In the third and fourth embodiments, the configuration using the converter and the energy storage conversion means 47 is shown, but the inverter itself deliberately pulsates current such as torque current and excitation current at 6 times the inverter output frequency, The inverter DC voltage can be pulsated. However, when the torque current is pulsated, the same component appears in the torque, which is not preferable. On the other hand, even if the exciting current is pulsated at 6 times the inverter output frequency, the vibration of the torque is hardly affected. The exciting current controls the magnetic flux, but the response time constant of the magnetic flux is several hundred ms. Since 6 times the inverter output frequency is on the order of several hundred Hz, the magnetic flux hardly reacts. Therefore, it is possible to pulsate the inverter DC voltage at the same frequency by keeping the torque current constant and pulsating the excitation current at 6 times the inverter output frequency. Thereby, a reduction in noise can be expected.

<Fifth Embodiment>
FIG. 8 is a block diagram showing the configuration of the fifth exemplary embodiment of the present invention.

  The fifth embodiment differs from the first embodiment shown in FIG. 1 in two points: a modulation rate limit command calculation unit 31 and a beatless control unit. For this reason, the description of the parts common to the first embodiment is omitted.

  The first point in the fifth embodiment is in the modulation rate limit command calculation unit 31.

  The motor speed detected by the speed detector 10 is input to the modulation rate limit command calculation unit 31. The limit value of the modulation rate, that is, the limit value of the magnitude of the output voltage is a function corresponding to the motor rotation speed as shown in FIG. 9, for example.

  It is known that motor noise in high-speed running is greatly affected by the fifth and seventh harmonics of the motor current. FIG. 10 shows the reduction rate with respect to the fifth and seventh harmonic currents at the time of one pulse with respect to the values of the fifth and seventh harmonic currents when the modulation rate is limited and the operation is performed with the one dash pulse. Is. Thus, by limiting the modulation rate and not operating with one pulse, the fifth and seventh current harmonics can be greatly reduced, and a reduction in noise can be expected.

  Further, as can be understood from the increase rate of the motor current simultaneously shown in FIG. 10, in order to obtain the same torque, the current is increased and the predetermined torque is output by the amount corresponding to the limit (decrease) of the modulation rate, that is, the output voltage. There is a need. As described above, although the harmonic current is slightly reduced and the current RMS is reduced, the motor current RMS is increased as shown in FIG. An increase in the motor current RMS results in a reduction in the efficiency of the motor and the inverter, and also causes an increase in the temperature of the device. In this case, it is indispensable to strengthen the cooler, which leads to an increase in size, weight, and cost of the device. In some cases, overcurrent may cause operation to stop. Therefore, it is desirable not to increase the motor current as much as possible. From the figure, the current increase rate is small on the low speed side of the field weakening control region (region in which the modulation rate (output voltage) is limited), and conversely, the current increase rate is large on the high speed side where the motor enters the characteristic region. Therefore, on the lower side of the field-weakening control region, the modulation factor limit value is lowered to a level acceptable for the system to reduce noise, and conversely, the modulation factor limit value is set to speed so that it falls within the current allowable value range in the high speed region. By increasing according to this, it is possible to reduce noise and avoid an increase in equipment.

  In the fifth embodiment, the modulation factor limit value is set as a function depending on the speed. However, it can be set according to the output. The larger the output, the larger the basic current, the more severe the thermal conditions of the motor and inverter, and the smaller the overcurrent tolerance. In the case of such a large output, it is necessary to give priority to driving continuity, not noise reduction.

  The second point of the fifth embodiment is that a beatless control unit 53 is provided in place of the beatless control unit 12 described above. In the beatless control unit 12 in the first embodiment, the phase angle of the inverter output voltage is corrected in accordance with the pulsation of the DC voltage, whereas in the beatless control unit 53 in the fifth embodiment, the DC voltage The modulation rate is corrected according to the pulsation.

  The DC voltage detected by the voltage detector 19 is input to the beatless control unit 53. The divider 54 divides the target voltage Vdc * of the DC voltage by the detected DC voltage and calculates power. The output of the divider 54 is multiplied by a modulation factor limit value, which is the output of the limit 23, in a multiplier 55.

  In this manner, the beatless control unit 53 calculates the pulsation rate of the DC voltage and multiplies the modulation rate to correct it. Since the magnitude of the output voltage is instantaneous modulation factor × DC voltage, the same operation can maintain the output voltage constant and suppress the beat phenomenon even when the DC voltage pulsates.

  In the beatless control for correcting the modulation rate, the output voltage is made constant as described above, so that the motor current does not increase. On the other hand, in the beatless control for correcting the output voltage phase angle of the first embodiment, the output voltage pulsates according to the DC voltage. However, the torque is made constant by controlling the phase angle. In this case, the motor current increases. Therefore, although a modulation rate correction method is desirable, a control margin for the modulation rate is necessary, so that it is not currently applied in the field-weakening control region.

  In the fifth embodiment, in order to reduce the motor noise in consideration of a slight increase in motor current, the modulation rate is limited, and it is preferable to apply a beatless control method for correcting the modulation rate. .

  In general, when the beat is completely suppressed by the modulation rate, a variation range of the modulation rate corresponding to the pulsation rate of the DC voltage is required. The current Shinkansen is planned to have a DC voltage pulsation rate of approximately 5% when the maximum load is applied. In this case, when the beatless control for correcting the modulation rate is applied, the average modulation rate (that is, the modulation rate limit value) is desirably 95% or less. However, as described above, limiting with a low modulation rate is undesirable because it increases the motor current. Therefore, as in the speed dependence characteristic of the modulation rate limit in FIG. 9, only the speed band in which the beat phenomenon is noticeable (near the inverter output frequency coincides with twice the power supply frequency, that is, the 100 Hz or 120 Hz band) In order to limit the modulation rate to less than%, give priority to low noise and beatless control effects, and suppress the increase in motor current in other areas, a method of increasing the modulation rate limit is effective.

The block diagram which shows the structure of the 1st Embodiment of this invention. The block diagram which shows the structure of a gate command calculating part. Explanatory drawing which shows a U-phase output voltage waveform. The block diagram which shows the structure of the 2nd Embodiment of this invention. The block diagram which shows the structure of the 3rd Embodiment of this invention. The block diagram which shows the structure of the 4th Embodiment of this invention. The block diagram which shows the structure of a prior art example. The block diagram which shows the structure of the 5th Embodiment of this invention. Explanatory drawing which shows the operation example of a modulation factor limit command calculating part. The characteristic view which shows the relationship between the reduction rate of the 5th-7th harmonic of a motor current, and the increase rate of a motor current.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... NPC inverter 2 ... Filter capacitor 3 ... Main motors 4, 35, 50 ... Current detector 5 ... Voltage calculation part 6, 7 ... Coordinate conversion part 8 ... Integrator 9 ... Gate command calculation part 10 ... Speed detection part 11 ... Slip frequency calculation unit 12, 53 ... Beatless control unit 13, 14, 24, 38, 44 ... Adder 15 ... Converter 16 ... Main transformer 17 ... Overhead wire 18 ... Pantograph 19, 34 ... Voltage detector 20 ... Wheel 21 ... Rail 22 ... Modulation rate calculation unit 23 ... Limiter 25 ... 1 pulse control unit 26 ... 1 dash pulse control unit 27 ... Triangular wave comparison PWM control unit 28 ... Switch 29 ... Control mode selection unit 30 ... Phase voltage command calculation unit 31 ... Modulation rate Limit command calculation unit 32 ... Inverter control unit 33 ... Converter control unit 36 ... Zero cross detection unit 37 ... Phase synchronization control unit 39 ... DC pulsation command calculation units 40, 42 ... Decrease Unit 41 ... Voltage control unit 43 ... Current control unit 45 ... Power supply voltage FF calculation unit 46 ... PWM control unit 47 ... Energy storage conversion means 48 ... Energy conversion unit 49 ... Reactor 51 ... Energy storage unit 52 ... Energy storage conversion control unit 54 ... Divisor 55 ... Multiplier

Claims (2)

A converter connected to the AC overhead line via a main transformer;
An inverter connected to the converter for converting direct current to three-phase alternating current and outputting it to the main motor ;
Inverter output frequency calculation means for calculating the output frequency of the voltage output by the inverter;
Inverter output voltage calculation means for calculating the magnitude of the voltage output from the inverter;
Gate command calculation means for generating a gate command for the inverter based on the frequency and magnitude of the voltage output from the inverter;
Voltage detecting means for detecting the voltage on the DC side of the inverter;
DC voltage control means for controlling the converter so that a voltage detection value of the voltage detection means matches a predetermined DC voltage command value;
DC pulsating means for pulsating the DC side voltage of the inverter at a frequency six times the inverter output frequency,
The direct current pulsation means includes:
A frequency calculating means for inputting the inverter output frequency and calculating a frequency six times the inverter output frequency;
DC voltage pulsation command calculation means for calculating a DC voltage pulsation command having a predetermined amplitude and phase shift amount at the calculated six times frequency;
Correction means for correcting the DC voltage command value by the calculated DC voltage pulsation command, so that the DC voltage decreases near the zero cross of each phase output voltage constituting the three-phase AC, and An electric vehicle control apparatus , wherein the phase shift amount is adjusted so that a DC voltage increases near a peak of a phase output voltage . A converter connected to the AC overhead line via a main transformer;
An inverter connected to the converter for converting direct current to three-phase alternating current and outputting it to the main motor ;
Inverter output frequency calculation means for calculating the output frequency of the voltage output by the inverter;
Inverter output voltage calculation means for calculating the magnitude of the voltage output from the inverter;
Gate command calculation means for generating a gate command for the inverter based on the frequency and magnitude of the voltage output from the inverter;
Energy storage conversion means comprising chopper means connected to the DC side of the inverter and energy storage means connected to the other of the chopper means;
Voltage detecting means for detecting the voltage on the DC side of the inverter;
DC voltage control means for controlling chopper means constituting the energy storage conversion means so that the voltage detection value of the voltage detection means matches a predetermined DC voltage command value;
DC pulsating means for pulsating the DC side voltage of the inverter at a frequency six times the inverter output frequency,
The direct current pulsation means includes:
A frequency calculating means for inputting the inverter output frequency and calculating a frequency six times the inverter output frequency;
DC voltage pulsation command calculation means for calculating a DC voltage pulsation command having a predetermined amplitude and phase shift amount at the calculated six times frequency;
Correction means for correcting the DC voltage command value by the calculated DC voltage pulsation command, so that the DC voltage decreases near the zero cross of each phase output voltage constituting the three-phase AC, and An electric vehicle control apparatus , wherein the phase shift amount is adjusted so that a DC voltage increases near a peak of a phase output voltage .

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