JP4240606B2 - Voltage resonant switching power supply circuit - Google Patents

Description

により表すことができる。
そして、巻線Ｎ3及び一次巻線Ｎ1のインダクタンスとしてＬ3＝Ｌ1の関係が得られるようにし、整流平滑電圧Ｅｉ、ブースト用ダイオードＤBの降下電圧ＶF、及びスイッチング素子Ｑ1の飽和電圧Ｖ(SAT)についてＥｉ≫ＶF，Ｖ(SAT)の関係が成立しているとすると、ブースト平滑電圧ＥBは上記（数１）に基づいて、
【数２】

により示されることになる。この場合、図３に示す電源回路では、例えば、直交型トランスＰＲＴの被制御巻線ＮRのインダクタンスＬRを、０．１×Ｌ1〜１．２×Ｌ1の範囲で変化させることで、ブースト平滑電圧ＥBについて、ほぼＥｉ〜２Ｅｉ（Ｅｉは、平滑コンデンサＣｉの両端に得られる整流平滑電圧レベルに相当する）の範囲で可変することが可能とされる。
【００３３】
このようにしてブースト回路が備えられる場合、本実施の形態の一次側の並列共振コンデンサＣｒは、ブースト回路の平滑コンデンサＣｉBの正極（ブースト平滑電圧ＥBのライン）と、スイッチング素子Ｑ1のコレクタ間に対して接続される。このような接続形態によっても、並列共振コンデンサＣｒの両端に得られる共振電圧Ｖｃｒはブースト平滑電圧ＥBによって基準電位がシフトされるので１４００Ｖよりも低い電圧が得られ、並列共振コンデンサＣｒとしては、１本のコンデンサとすることができる。つまり、本実施の形態の構成によっても並列共振コンデンサＣｒの小型化が図られる。
【００３４】
また、図３に示す絶縁コンバータトランスＰＩＴの二次側では、二次巻線Ｎ2にセンタータップが設けられたうえで、整流ダイオードＤO1，ＤO2，ＤO3，ＤO4及び平滑コンデンサＣO1，ＣO2を図のように接続することで、［整流ダイオードＤO1，ＤO2，平滑コンデンサＣO1］の組と、［整流ダイオードＤO3，ＤO4，平滑コンデンサＣO2］の組とによる、２組の全波整流回路が設けられる。
［整流ダイオードＤO1，ＤO2，平滑コンデンサＣO1］から成る全波整流回路は直流出力電圧ＥO1を生成し、［整流ダイオードＤO3，ＤO4，平滑コンデンサＣO2］から成る全波整流回路は直流出力電圧ＥO2を生成する。この二次側出力電圧は後段の負荷（図示しない）に供給される。
【００３５】
本実施の形態の絶縁コンバータトランスＰＩＴは、図４に示すように、例えばフェライト材によるＥ型コアＣＲ１、ＣＲ２を互いの磁脚が対向するように組み合わせたＥＥ型コアが備えられ、このＥＥ型コアの中央磁脚に対して、ボビンＢを利用して一次巻線Ｎ1（及びＮ3，ＮB） と、二次巻線Ｎ2をそれぞれ分割した状態で巻装している。そして、本実施の形態では、中央磁脚に対しては図のようにギャップＧを形成するようにしている。これによって、所要の結合係数による疎結合が得られるようにしている。
【００３６】
絶縁コンバータトランスＰＩＴにおいては、一次巻線Ｎ1 、二次巻線Ｎ2 の極性（巻方向）と整流ダイオードＤO （ＤO1，ＤO2／ＤO3，ＤO4）の接続との関係によって、一次巻線Ｎ1 のインダクタンスＬ1と二次巻線Ｎ2 のインダクタンスＬ2 との相互インダクタンスＭについて、＋Ｍとなる場合と−Ｍとなる場合とがある。
例えば、図５（ａ）に示す接続形態を採る場合に相互インダクタンスは＋Ｍとなり、図５（ｂ）に示す接続形態を採る場合に相互インダクタンスは−Ｍとなる。
これを、上述した本実施の形態の二次側の動作に対応させてみると、例えば二次巻線Ｎ2に得られる交番電圧が正極性のときに整流ダイオードＤO1，ＤO3に整流電流が流れる動作は＋Ｍの動作モード（フォワード方式）とみることができ、逆に、二次巻線Ｎ2に得られる交番電圧が負極性のときに整流ダイオードＤO2，ＤO4に流れる整流電流は−Ｍの動作モード（フライバック方式）であるとみることができる。即ち、本実施の形態では、二次巻線に得られる交番電圧が正／負となるごとに、相互インダクタンスが＋Ｍ／−Ｍのモードで動作することになる。
【００３７】
このようにして、二次側において全波整流回路を設けた場合、本実施の形態としての二次側並列共振コンデンサは、直流出力電圧Ｅ01を生成する整流回路に関すれば、図のように、整流ダイオードＤO1，ＤO2に対して、それぞれ二次側並列共振コンデンサＣ2A，Ｃ2Bを並列に接続するようにされる。これにより、整流ダイオードＤO1のオフ時には、平滑コンデンサＣO1を介して二次側並列共振コンデンサＣ2A及び二次巻線Ｎ2とにより並列共振回路を形成し、整流ダイオードＤO2のオフ時には、平滑コンデンサＣO2を介して二次側並列共振コンデンサＣ2B及び二次巻線Ｎ2とにより並列共振回路を形成することになる。
このような接続形態によっても、先の実施の形態と同様に、二次側並列共振コンデンサＣ2A，Ｃ2Bとにより二次側の整流ダイオードの逆回復時間時における輻射ノイズの発生が抑えられるため、セラミックコンデンサＣｐを設ける必要はなくなる。
【００３８】
この図に示す電源回路のように、二次側並列共振コンデンサＣ2A，Ｃ2Bを設けて二次側並列共振回路を形成した構成とすると、この二次側並列共振回路の作用によって負荷側に電力が供給されるため、二次側並列共振コンデンサを設けない場合よりも、最大負荷電力が増加するのは前述したとおりである。
これに加えて、この図に示す回路のようにして、二次側並列共振回路に対して全波整流回路を接続した場合、前述のように、相互インダクタンスが＋Ｍ／−Ｍの両方の動作モードで交互に整流電流が流れるようにされる。つまり、交番電圧が正極と負極との両期間において整流出力が得られるようにされるので、それだけ負荷側に供給される電力も増加して、最大負荷電力の増加率も向上する。
【００３９】
この場合の制御回路１としては、直流出力電圧ＥO1が一定となるように直交型トランスＰＲＴの制御巻線ＮCに対して、直流出力電圧ＥO1の変動に応じたレベルの制御電流を流して、被制御巻線ＮRのインダクタンスＬRを可変するように動作する。これにより、前述したインダクタンス制御方式による定電圧制御が行われる。
また、この場合には、インダクタンスＬRの可変によって（数１）により表されるブースト平滑電圧ＥBを一定とするように制御することになるが、この作用によって、二次側の直流出力電圧を一定とするように動作する。
【００４０】
例えば、先に本出願人が提案したソフトスイッチング電源回路の構成では、例えば交流入力電圧ＡＣ１００Ｖ系で、かつ最大負荷電力１５０Ｗ〜１６０Ｗ以上に対応する構成を採る場合には、倍電圧整流回路により交流入力電圧のほぼ２倍のレベルに対応する整流平滑電圧を得ていた。これにより、スイッチングコンバータに入力される直流電圧レベルを増加させて比較的高負荷の条件に対応していたものである。
【００４１】
これに対して図３に示す電源回路では、上述のようにして最大負荷電力の増加を図ることで、直流入力電圧（整流平滑電圧）を生成する整流平滑回路としては倍電圧整流方式を採って負荷電力をカバーする必要はなくなる。この結果、図３に示すようにして、例えばブリッジ整流回路による通常の等倍電圧整流回路の構成を採ることができるものである。
これにより、例えば図３に示す電源回路では、交流入力電圧ＶAC＝１４４Ｖ時における整流平滑電圧Ｅｉは２００Ｖ程度となる。ここで、スイッチング素子Ｑ1のコレクタ−エミッタ間電圧ＶCPは、整流平滑電圧Ｅｉに対して一次側の並列共振回路が作用することで、スイッチング素子Ｑ1のオフ時に発生するが、図３の回路では、上記のように整流平滑電圧Ｅｉが倍電圧整流時の約１／２とされることになる。但し、図３に示す構成では、この整流平滑電圧Ｅｉに対してブースト電圧ＶBを重畳してブースト平滑電圧ＥBが発生するため、共振電圧Ｖｃｒはブースト平滑電圧ＥBのレベルに依存するのであるが、それでも共振電圧Ｖｃｒは１２００Ｖ程度にまで抑えられる。
従って、図３に示す回路においては、スイッチング素子Ｑ1については、１２００Ｖの耐圧品を選定すればよいことになる。
【００４２】
また、上述したように、二次側において全波整流回路を設け、二次巻線Ｎ2Aの交番電圧が正負の両期間において整流電流が流れるようにしたことで、二次側の共振電圧Ｖ2は正負の両期間において共に整流平滑電圧Ｅｉと同等のレベルにまで抑制されることになる。
これにより、二次側の全波整流回路を形成する整流ダイオード（ＤO1〜Ｄ04）としては、整流平滑電圧Ｅｉのレベルにほぼ対応する耐圧品を選定すればよいことになる。
【００４３】
このように、図３に示す回路では、並列共振コンデンサＣｒが小型となり、スイッチング素子Ｑ1、及び二次側の全波整流回路を形成する整流ダイオードについて低耐圧品を用いることができるため、素子としてはそれだけ安価となる。このため、特にコストアップを考慮することなく、例えばスイッチング素子Ｑ1及び二次側の全波整流回路を形成する整流ダイオードについて特性の向上されたもの（スイッチング素子Ｑ1であれば、飽和電圧ＶCE(SAT)、蓄積時間ｔSTG、下降時間ｔｆ、電流増幅率ｈFE等の特性の良好なもの、また、整流ダイオードであれば順方向電圧降下ＶF、逆回復時間ｔｒｒ等の特性の良好なもの）を選定することができ、それだけ電力損失の低減が促進されることにもなる。つまり、倍電圧整流回路により直流入力電圧を生成する構成よりも、低コスト或いはほぼ同等のコストでありながら電力変換効率の向上を図ることが可能になる。また、電力変換効率の向上により、例えば倍電圧整流回路の整流ダイオードの放熱のために必要であった放熱板等も不要となる。
【００４４】
また、電圧共振形コンバータの構成として、例えば１００ＫＨｚ程度の高いスイッチング周波数を設定するようにすれば、上記各種部品の小型・軽量化も図られることになる。ここで、実際に対応すべき最大負荷電力に応じて、ブースト電圧ＶBが最適となるように巻線Ｎ3を選定すれば、更なる各種部品の小型・軽量化を実現できる。
更に、電源回路の小型・軽量化の観点からすれば、直流入力電圧の生成のために倍電圧整流回路を備える構成では、それぞれ２組の整流ダイオードと平滑コンデンサが必要とされたのであるが、図３に示す回路では、例えば通常のブリッジ整流回路による全波整流回路とされるため、１組のブロック型の平滑コンデンサとブリッジ整流ダイオードを採用することができるので、この点でも、コストの削減及び部品の小型化が図られるものである。つまり、図３に示す回路では、絶縁コンバータトランスＰＩＴ、直交型制御トランスＰＲＴを含む各種部品の小型化が図られる。また、電力変換効率も向上が図られることが実験により分かっている。
【００４５】
図６は、本発明の第３の実施の形態としての電源回路の構成を示す回路図である。この図に示す電源回路は、商用交流電源がＡＣ１００Ｖ系で、かつ、最大負荷電力１６０Ｗ以上の比較的重負荷に対応可能な構成とされる。なお、図１及び図３と同一部分については同一符号を付して説明を省略する。
【００４６】
本実施の形態の場合、交流入力電圧ＶACを入力して整流平滑電圧Ｅｉを得るための整流回路としては、ブリッジ整流回路Ｄｉ及び１本の平滑コンデンサＣｉにより形成される。
これは、次に説明するようにして、２石のスイッチング素子によるプッシュプルのスイッチング動作により、スイッチング出力の増加を図ることで負荷電力を賄うようにされるため、特に、直流入力電圧については２Ｅｉのレベルが要求されないことに因る。
【００４７】
また、一次巻線Ｎ1にはセンタータップが設けられて、一次巻線Ｎ1A（インダクタンスＬ1A），Ｎ1B（インダクタンスＬ1B）に分割される。このセンタータップ端子は、直交型制御トランスＰＲＴの被制御巻線ＮRの直列接続を介して整流平滑電圧Ｅｉのラインと接続される。
【００４８】
この図に示す電圧共振形コンバータは、２石のスイッチング素子Ｑ1，Ｑ2を備えた他励式の構成を採っている。この場合、スイッチング素子Ｑ1には、ＭＯＳ−ＦＥＴが採用されている。
この場合、スイッチング素子Ｑ1 のドレインは一次巻線Ｎ1Aの端部と接続され、ソースは一次側アースに接地される。また、スイッチング素子Ｑのドレイン−ソース間には、スイッチングオフ時の期間電流の経路を形成するダンパーダイオードＤD1が並列に接続される。
スイッチング素子Ｑ2 のドレインは一次巻線Ｎ1Bの端部と接続され、ソースは一次側アースに接地される。また、スイッチング素子Ｑのドレイン−ソース間に対しても、ダンパーダイオードＤD2が並列に接続される。
【００４９】
そして、本実施の形態の一次側の並列共振コンデンサとしては、スイッチング素子Ｑ1，Ｑ2に対応して、それぞれ並列共振コンデンサＣｒ1，Ｃｒ2が備えられる。
並列共振コンデンサＣｒ1は、スイッチング素子Ｑ1のコレクタと平滑コンデンサＣｉの正極（整流平滑電圧Ｅｉライン）間に対して挿入されるようにして接続されることで、並列共振コンデンサＣｒ1のキャパシタンスと、インダクタンスＬ1A，インダクタンスＬRの合成インダクタンスとによって、スイッチング素子Ｑ1のスイッチング動作を電圧共振形とする並列共振回路が形成される。
同様にして、並列共振コンデンサＣｒ2は、スイッチング素子Ｑ2のコレクタと平滑コンデンサＣｉの正極（整流平滑電圧Ｅｉライン）間に対して挿入されるようにして接続されることで、並列共振コンデンサＣｒ2のキャパシタンスと、インダクタンスＬ1B，インダクタンスＬRの合成インダクタンスとによって、スイッチング素子Ｑ2のスイッチング動作を電圧共振形とする並列共振回路を形成する。なお、従来の構成では、並列共振コンデンサＣｒ1，Ｃｒ2は、それぞれスイッチング素子Ｑ1，Ｑ2のコレクタ−エミッタ間に対して並列に接続されるものである。
上記スイッチング素子Ｑ1，Ｑ2のスイッチング出力点は、一次巻線Ｎ1のセンタータップとされることになる。
つまり、本実施の形態では、スイッチング素子Ｑ1，Ｑ2をそれぞれ駆動する二系統のスイッチング回路系が備えられることになる。
【００５０】
発振ドライブ回路２は、起動抵抗ＲSにより起動されるようになっており、例えばスイッチング周波数ｆｓ＝１００ＫＨｚに対応する周波数信号を発振出力すると共に、この周波数信号に基づいてスイッチング駆動信号（駆動電圧）を生成して、スイッチング素子Ｑ1，Ｑ2のゲートに印加する。この際、スイッチング素子Ｑ1，Ｑ2のゲートに印加するスイッチング駆動信号としては、互いに反転した逆極性の信号とされる。
【００５１】
ここで、スイッチング素子Ｑ1，Ｑ2が共にスイッチング動作を行うとすると、スイッチング素子Ｑ1，Ｑ2は、例えばスイッチング周波数ｆｓ＝１００ＫＨｚで、交互にオン／オフとなるタイミングでスイッチング動作を行うことになる。
つまり、スイッチング素子Ｑ1，Ｑ2が共にスイッチングを行う場合には、いわゆるプッシュプル方式によるスイッチング動作（プッシュプル動作）となり、例えば１石のスイッチング素子による、いわゆるシングルエンド方式によるスイッチング動作（シングルエンド動作）の場合よりも大きな電力を負荷側に対して供給することができる。これにより、本実施の形態では最大負荷電力１６０Ｗ以上の重負荷の条件に対応する。
【００５２】
また、この図に示す制御回路１は、例えば先に図１及び図３に示した制御回路１と同様に、二次側の直流電圧出力（ＥO1）と基準電圧を比較してその誤差に応じた直流電流を、制御電流として直交型制御トランスＰＲＴの制御巻線ＮC に供給する誤差増幅器として構成される。
【００５３】
例えば、交流入力電圧ＶAC或いは最小負荷電力の変動に伴って二次側の直流出力電圧ＥO2が変動した時は、制御回路１によって制御巻線ＮC に流れる制御電流を例えば所定の範囲で変化させる。これにより、被制御巻線ＮR のインダクタンスＬR が所定の範囲で変化するようにされる。
【００５４】
本実施の形態の場合、前述のように、被制御巻線ＮR はスイッチング素子Ｑ1側の並列共振回路と、スイッチング素子Ｑ2側の並列共振回路とを形成するようにして挿入されている。このため、インダクタンスＬR が変化することで、スイッチング素子Ｑ1及びスイッチング素子Ｑ2のスイッチング動作時において、固定とされているスイッチング周波数に対して、この共振回路の共振条件が変化するようにされる。この共振条件の変化に応じて二次側の直流出力電圧（ＥO1，ＥO2）の出力レベルは変化することになるが、これによって二次側直流電圧（ＥO1，ＥO2）の定電圧化が図られることになる。つまり、この場合にもインダクタンス制御方式による定電圧制御動作となる。
【００５５】
図６に示す構成によっても、先の第１，第２の実施の形態と同様に、一次側の並列共振コンデンサＣｒ1，Ｃｒ2の小型化、及び二次側における輻射ノイズ吸収のためのセラミックコンデンサＣｐは省略される。
【００５６】
なお、本発明は、上記第１〜第３の実施の形態として各図に示した構成に限定されるものではなく、変更が可能である。
例えば、図３及び図６に示した構成では、例えば二次側の整流回路について、倍電圧全波整流回路を備えるように構成することも可能で、この場合には、例えば、等倍電圧の全波整流回路と同等レベルの直流出力電圧を得ようとすれば、二次巻線Ｎ2の巻数は１／２に削減することが可能になる。また、スイッチング素子についも、バイポーラトランジスタだけではなく、ＭＯＳ−ＦＥＴトランジスタをはじめとする他の種類のスイッチング素子が採用されて構わないものである。
【００５７】
【発明の効果】
以上説明したように本発明の電圧共振形コンバータは、スイッチング動作を電圧共振形とするための一次側共振回路の共振コンデンサを、直流入力電圧の正極ラインとスイッチング手段のスイッチング出力端子間に対して挿入するようにしている。これによって、一次側共振回路の作用によって共振コンデンサの両端に発生する共振電圧の基準電位は、直流入力電圧レベル分シフトすることになり、それだけスイッチング素子オフ時に共振コンデンサの両端に発生する共振電圧パルスのレベルを低下させることが可能になる。
これにより実際の回路では、製造の限界であるコンデンサ（例えばポリプロピレンフィルムコンデンサ）の耐圧内に納めることが可能となり、２本以上のコンデンサを直列接続して共振コンデンサを形成する必要が無くなる。つまり、共振コンデンサとしては、その直列接続数を減らして例えば１本のコンデンサとすることも可能であり、それだけ共振コンデンサの小型軽量化、及びコストの削減を図ることが可能になる。
【００５８】
また、二次側巻線のインダクタンス成分と共に二次側並列共振回路を形成する二次側並列共振コンデンサを、二次側の整流回路を形成する二次側整流ダイオードに対して並列に接続することで、二次側整流ダイオードの逆回復時間時に発生する輻射ノイズを解消するようにも動作するため、従来、輻射ノイズ抑制のために二次側整流ダイオードに並列に接続されていたセラミックコンデンサを省略することが可能となる。
このようにして、本発明では、電圧共振形スイッチング電源回路の回路の小型軽量化、及び低コスト化を促進することが可能になるという効果を有するものである。
【図面の簡単な説明】
【図１】本発明の第１の実施の形態の電源回路の構成例を示す回路図である。
【図２】実施の形態の電源回路の要部の動作を示す波形図である。
【図３】本発明の第２の実施の形態の電源回路の構成例を示す回路図である。
【図４】絶縁コンバータトランスの構成を示す断面図である。
【図５】相互インダクタンスが＋Ｍ／−Ｍの場合の各動作を示す説明図である。
【図６】本発明の第３の実施の形態の電源回路の構成例を示す回路図である。
【図７】従来例としての電源回路の構成例を示す回路図である。
【図８】図７に示す電源回路の要部の動作を示す波形図である。
【符号の説明】
１ 制御回路、２ 発振回路、３ ドライブ回路、Ｃｉ，ＣｉB 平滑コンデンサ、Ｃｒ，Ｃｒ1，Ｃｒ2 並列共振コンデンサ、Ｃ2，Ｃ2A，Ｃ2B （二次側）並列共振コンデンサ、Ｄｉ ブリッジ整流回路、ＤO1，ＤO2，ＤO3，ＤO4 整流ダイオード、ＰＩＴ 絶縁コンバータトランス、Ｎ1，Ｎ1A，Ｎ1B 一次巻線、Ｎ2 二次巻線、ＰＲＴ 直交型制御トランス、ＮC 制御巻線、ＮR 被制御巻線、Ｑ1，Ｑ2 スイッチング素子、ＤB ブースト用ダイオード、Ｎ3 （ブースト用）巻線、[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a voltage resonance type switching power supply circuit provided as a power source in various electronic devices.
[0002]
[Prior art]
As a so-called soft switching power supply circuit, a switching power supply circuit including a voltage resonance type switching converter is known. The voltage resonance type switching converter has a low noise because a smooth waveform is obtained with respect to the switching output voltage pulse and the switching output current flowing into the insulating converter transformer, and can be configured with a relatively small number of components.
[0003]
The circuit diagram of FIG. 7 shows an example of a voltage resonance type switching power supply circuit that can be configured based on the invention previously proposed by the present applicant.
In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC100V system, and the maximum load power corresponds to a condition of 150 W or more.
[0004]
In the switching power supply circuit shown in this figure, a so-called voltage doubler rectifier circuit including rectifier diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 is provided as a rectifying / smoothing circuit for rectifying and smoothing the AC power supply AC. In this voltage doubler rectifier circuit, for example, when a DC input voltage corresponding to one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei approximately twice that is generated. For example, if the AC input voltage VAC = 144V, the DC input voltage 2Ei is about 400V.
As described above, the voltage doubler rectifier circuit is adopted as the rectifying / smoothing circuit, as described above, in response to a relatively heavy load condition in which the AC input voltage is AC100V and the maximum load power is 150 W or more. It is for the purpose. That is, by making the DC input voltage twice that of the normal one, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. .
For the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. Yes.
[0005]
The voltage resonance type switching converter in this figure has a self-excited configuration including one switching element Q1. In this case, a high breakdown voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.
The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothing voltage 2Ei) via the starting resistor RS so that the base current at the time of starting can be obtained from the rectifying smoothing line. A resonant circuit for self-excited oscillation comprising a damping resistor RB, an inductor LB, a resonant capacitor CB, and a drive winding NB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is wound around an insulating converter transformer PIT (Power Isolation Transformer), and the required inductance for setting the switching frequency is obtained together with the inductor LB in the resonance circuit for self-excited oscillation. Has been.
A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitor Ci forms a path for a damper current that flows when the switching element Q1 is turned off. The collector of the switching element Q1 is connected to one end of the primary winding N1 of the insulating converter transformer PIT, and the emitter is grounded.
[0006]
A parallel resonant capacitor Cr is connected in parallel between the collector and emitter of the switching element Q1. This parallel resonant capacitor Cr has a combined inductance (capacitance obtained by series connection of its own capacitance, a primary winding N1 of an insulating converter transformer PIT described later, and a controlled winding N R of an orthogonal control transformer PRT (Power Regulating Transformer)). L1 + LR) forms a parallel resonance circuit of a voltage resonance type converter. Although detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcr across the resonance capacitor Cr is actually a sinusoidal pulse waveform due to the action of the parallel resonance circuit. Can be obtained.
[0007]
The parallel resonant capacitor Cr is actually configured as a set of package parts after two capacitors CrA and CrB are connected in series, for example, by a low-loss polypropylene film capacitor with respect to a high-frequency current. The reason for this will be described later.
[0008]
The insulating converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1. The other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.
[0009]
On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary resonant circuit is formed by connecting a secondary side parallel resonant capacitor C2 in parallel to the secondary winding N2.
By this parallel resonance circuit, the alternating voltage excited in the secondary winding N2 is a resonance voltage, and this resonance voltage is generated from a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and from the rectifier diode DO2 and the smoothing capacitor CO2. Are supplied to two sets of half-wave rectifier circuits. The two sets of half-wave rectifier circuits provide DC output voltages EO1 and EO2, respectively.
At this time, the secondary side parallel resonant capacitor C2 is provided as described above to form a resonance circuit, and thus the amount of current flowing on the secondary side is increased by the resonance action. As a result, the maximum load power that can be handled is increased.
The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed type for rectifying the alternating voltage of the switching period.
[0010]
The control circuit 1 is, for example, an error amplifier that compares a DC voltage output on the secondary side with a reference voltage and supplies a DC current corresponding to the error to the control winding NC of the orthogonal control transformer PRT as a control current. In this case, a DC voltage output E01 is input to the control circuit 1 as a detection voltage, and a DC output voltage E02 is input as an operating power supply.
[0011]
The orthogonal control transformer PRT is wound with a controlled winding NR so that its winding direction is orthogonal to the control winding NC, and a gap G with respect to a required portion of the magnetic leg of the core. It is comprised so that it may become a saturable reactor by forming.
[0012]
For example, when the secondary side DC output voltage E02 varies with the variation of the AC input voltage VAC or the minimum load power, the control current flowing through the control winding NC is changed within a required range by the control circuit 1. As a result, the orthogonal control transformer PRT operates so that the inductance LR of the controlled winding NR changes within a predetermined range.
[0013]
Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage resonance type switching operation as described above, the resonance of the parallel resonance circuit is fixed with respect to a fixed switching frequency. Conditions are made to change. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit The width of the resonance pulse is variably controlled by changing. That is, a PWM (Pulse Width Moduration) control operation for the resonance pulse is obtained. The PWM control of the resonance pulse width is control of the off period of the switching element Q1. In other words, this means that the on period of the switching element Q1 is variably controlled under the condition of a fixed switching frequency. . In this way, the ON period of the switching element Q1 is variably controlled, so that the switching output transmitted from the primary winding N1 forming the parallel resonant circuit to the secondary side changes, and the secondary side DC output voltage ( The output level of EO1, EO2) is also changed. As a result, the secondary side DC voltage (EO1, EO2) is made constant. Such a constant voltage control method is hereinafter referred to as an inductance control method.
[0014]
If the power supply circuit having the configuration shown in FIG. 7 is adapted to an AC 200V region such as Europe, for example, a rectifier circuit that generates a rectified and smoothed voltage is replaced by a bridge rectifier circuit, for example, A full-wave rectifier circuit may be provided so as to generate a rectified and smoothed voltage corresponding to the AC input voltage VAC.
[0015]
Here, in the power supply circuit having the configuration shown in FIG. 7 in FIG. 8, it is necessary for the switching cycle when the load is short-circuited with the AC input voltage VAC = 144 V and the rectified smoothing voltage (DC input voltage) Ei = 400 V. The operation waveform of a part is shown.
[0016]
Under this condition, the parallel resonance circuit on the primary side is stabilized by the action of the capacitance of the parallel resonance capacitor Cr, the inductance L1 of the primary winding N1, and the inductance LR of the controlled winding NR whose variable state is controlled. Voltage-resonant switching operation can be obtained. For this reason, at the both ends of the parallel connection circuit of the switching element Q1 // parallel resonant capacitor Cr, a sinusoidal wave is formed during the period when the switching element Q1 is off as shown in FIG. A resonance voltage Vcr is obtained. The resonance voltage Vcr has a peak level of about 1800V. Further, during the period when the switching element Q1 is off, the resonance current Icr flows through the parallel resonance capacitor Cr as shown in FIG. 8B.
On the secondary side, the voltage (resonance voltage) VC2 across the secondary side parallel resonant capacitor C2 is a DC output voltage during the period in which the rectifier diodes DO1 and D02 are conducting, as shown in FIG. A level of EO (EO1, EO2) is obtained, and during the period when the rectifier diodes DO1, DO2 are non-conductive, a sine wave-shaped peak level of 285 V is obtained by the resonance operation of the secondary side parallel resonance circuit. For this reason, the Peak To Peak level of the resonance voltage VC2 is about 420 Vp-p if the DC output voltage EO is about 135V.
Further, the resonance current IC2 flowing through the secondary side parallel resonance capacitor C2 is changed from a positive level to a negative level by a sinusoidal curve during a period in which the rectifier diodes DO1 and DO2 are non-conductive as shown in FIG. Inverting waveform is obtained.
[0017]
[Problems to be solved by the invention]
As can be seen from the waveform shown in FIG. 8A, a maximum voltage of 1800 V is applied to the switching element Q1 and the parallel resonant capacitor Cr. For this reason, it is necessary to use a 1800 V withstand voltage product for the switching element Q1 and the parallel resonant capacitor Cr.
However, in the case of the parallel resonant capacitor Cr, in order to provide a low loss characteristic with respect to a resonant current having a high frequency, for example, in practice, the derivative is a polypropylene film, the electrode is a metallized polyester film, and the exterior is a flame-retardant epoxy. A so-called polypropylene film capacitor made of resin is used. This polypropylene film capacitor is the limit at which a withstand voltage of 1600 V is currently manufactured.
For this reason, as shown in FIG. 7, as an actual parallel resonant capacitor Cr, for example, two polypropylene film capacitors (CrA, CrB) of 0.02 μF / 1000 V are connected in series, and these two capacitors are plastic. By molding with epoxy resin in the case, it is configured as a single capacitor of 0.01 μF / 2000V.
In this case, for example, in order to obtain the capacitance of 0.01 μF required as the parallel resonance capacitor Cr, two capacitors that are enlarged according to the double capacitance are required. The size as the capacitor Cr is increased. For example, the external dimension of the parallel resonant capacitor Cr configured as described above is about 13 × 22 × 32 mm.
[0018]
Further, as explained in FIG. 7, on the secondary side, a high-frequency alternating voltage due to the switching cycle is rectified by a half-wave rectifier circuit including a high-voltage recovery high-speed recovery type rectifier diode (DO1, DO2). Although a DC output voltage is obtained, unnecessary radiation noise is generated during the reverse recovery time of the rectifier diode. Therefore, in practice, as shown in FIG. 7, the radiant noise is absorbed by connecting small-capacity ceramic capacitors Cp and Cp in parallel to the rectifier diodes (DO1 and D02). ing.
[0019]
As a switching power supply circuit, it is preferable to reduce size and weight by reducing the number of component elements and reducing the size of the element itself, but as described above, in the circuit configuration shown in FIG. Due to the increase in size of the primary side parallel resonance capacitor Cr and the capacitors Cp and Cp added to the secondary side, there is a limit to the reduction in size and weight of the circuit board.
[0020]
[Means for Solving the Problems]
In view of the above-described problems, the present invention generates a rectified and smoothed voltage by inputting a commercial AC power supply and performing rectification and smoothing, and outputs the rectified and smoothed voltage as a DC input voltage. Switching means configured to output to the primary winding of the insulating converter transformer, at least between the inductance component including the primary winding of the insulating converter transformer, the output terminal of the switching means, and the positive line of the DC input voltage. A primary-side resonance circuit that is formed by the capacitance of the resonance capacitor inserted and has a voltage resonance type operation of the switching means, a secondary-side rectifier diode element, and a secondary-side smoothing capacitor. DC output voltage generator that generates the secondary side DC output voltage from the alternating voltage obtained in the secondary winding And a secondary side resonant circuit formed on the secondary side by the inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side parallel resonant capacitor connected in parallel to the secondary side rectifier diode element And a controlled winding provided so as to function as an inductance component of the primary side resonance circuit, and an orthogonal in which the controlled winding and a control winding whose winding directions are orthogonal to each other are wound. A constant-type voltage control for the secondary side DC output voltage by changing the inductance of the controlled winding by passing a variable control current through the control winding according to the level of the DC output voltage. The voltage resonance type switching power supply circuit is configured to include constant voltage control means configured to perform.
[0021]
According to the above configuration, in the voltage resonance type converter, the resonance capacitor of the primary side resonance circuit for switching the voltage operation to the voltage resonance type is inserted between the positive line of the DC input voltage and the switching output terminal of the switching means. As a result, the reference potential of the resonance voltage generated at both ends of the resonance capacitor by the action of the primary side resonance circuit is shifted by the DC input voltage level.
[0022]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to an embodiment of the present invention. The same parts as those in FIG. 7 are denoted by the same reference numerals, and the description of the parts having the same circuit configuration is omitted.
In the power supply circuit shown in this figure, the primary parallel resonant capacitor Cr is provided so as to be inserted between the positive electrode (rectified smoothed voltage (DC input voltage) line) of the smoothing capacitor Ci1 and the collector of the switching element. In such a connection form, it can be considered that one end of the parallel resonant capacitor Cr is connected to the primary side ground through a series connection of the smoothing capacitors Ci1 to Ci2. The parallel resonance circuit of the voltage resonance type converter is formed by the combined inductance (L1 + LR) obtained by the serial connection of the primary winding N1 and the controlled winding NR.
[0023]
On the secondary side, the secondary side parallel resonant capacitor C2 is connected in parallel to the rectifier diode D01.
[0024]
FIG. 2 shows an operation waveform diagram of the main part according to the switching cycle of the power supply circuit having such a configuration.
2A shows the collector-emitter voltage VCP of the switching element Q1, FIG. 2B shows the resonance voltage Vcr obtained across the parallel resonance capacitor Cr, and FIG. 2C shows the secondary side parallel resonance capacitor. The resonance voltage VC2 of C2 is shown. In this case, the resonance voltage Vcr shown in FIG. 2B has a waveform that becomes a sinusoidal voltage resonance pulse when the switching element Q1 is turned off by the action of the parallel resonance circuit on the primary side formed as described above. The resonance voltage VC2 shown in FIG. 2C is obtained by connecting the secondary side parallel resonance capacitor C2 to the secondary side ground through the smoothing capacitor CO1 during the period when the rectifier diode D01 is turned off. This is a waveform in which a sinusoidal voltage resonance pulse is obtained by the secondary side parallel resonance circuit formed together with the inductance L2 of the winding N2.
[0025]
As the collector-emitter voltage VCP of the switching element Q1 shown in FIG. 2A, a peak level of 1800 V appears when the switching element Q1 is turned off, similarly to the resonance voltage Vcr shown in FIG.
On the other hand, in the resonance voltage Vcr shown in FIG. 2B, the zero point of the reference potential is offset at the level of 2Ei of the rectified and smoothed voltage. For this reason, the peak level of the resonance voltage Vcr when the switching element Q1 is turned off is reduced to 1400V (= 1800V-400V). The offset of the reference potential 0 point is an operation obtained by connecting one end of the parallel resonance capacitor Cr to the positive side of the series connection of the smoothing capacitors Ci1 to Ci2 (the positive terminal of the smoothing capacitor Ci1).
[0026]
For this reason, in this embodiment, it is only necessary to obtain a withstand voltage of 1400 V for the parallel resonant capacitor Cr.
As described above, a polypropylene film capacitor has a withstand voltage of 1600 V, which is the limit of manufacturing. In this embodiment, as a parallel resonant capacitor Cr actually mounted on a circuit board, for example, Cr = 0.01 μF / If one having a characteristic of 1600 V is selected, the withstand voltage is sufficient, and a single polypropylene film capacitor is sufficient. At this time, the capacitance of the parallel resonant capacitor Cr is set to 0.01 μF as described above, so that the polypropylene film capacitors C2A and C2B provided in the circuit shown in FIG. 7 are set to 0.02 μF. As compared with the above, the parallel resonant capacitor Cr of the present embodiment is considerably small. Specifically, the external dimension of the 0.01 μF / 1600 V (DC) parallel resonant capacitor Cr is 6 × 15 × 26.5 mm, which is about ¼ the size of the parallel resonant capacitor Cr shown in FIG. It becomes.
[0027]
In addition, since the secondary side parallel resonant capacitor C2 is connected in parallel to the rectifier diode DO1, the resonant voltage VC2 of the secondary side parallel resonant capacitor C2 is, as shown in FIG. The period during which D01 is turned on is approximately 0 level, and the peak level is 285 V during the period when it is turned off. Therefore, the Peak to Peak level is approximately 285 Vp-p, and the circuit shown in FIG. 7 is 420 Vp-p. It is greatly reduced compared to what was there. For example, a polypropylene capacitor is used as the secondary side parallel resonance capacitor C2. However, even if the resonance voltage VC2 is 285 Vp-p, a polypropylene capacitor having a withstand voltage of 400 V (DC) should be selected at present. Therefore, not much downsizing can be expected. However, for example, if a smaller polypropylene capacitor having a withstand voltage of about 300 V (DC) is available, the secondary side parallel resonant capacitor C2 can be downsized.
However, in the present embodiment, the secondary side parallel resonant capacitor C2 is connected in parallel to the rectifier diode D01, so that the secondary side parallel resonant capacitor C2 is generated during the reverse recovery time of the secondary side rectifier diode. It will also work to absorb radiation noise. For this reason, in this embodiment, the ceramic capacitor Cp shown in FIG. 7 can be omitted.
[0028]
FIG. 3 is a circuit diagram showing a configuration of a power supply circuit according to the second embodiment of the present invention. The same parts as those in FIG.
[0029]
The power supply circuit shown in FIG. 3 includes a full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci as a rectifier / smoothing circuit for obtaining an AC input voltage by inputting the AC input voltage VAC. A rectified and smoothed voltage Ei corresponding to a level of 1 times VAC is generated. That is, in the present embodiment, the voltage doubler rectifier circuit is not provided as in the conventional configuration and the configuration of FIG.
In this specification, a full-wave rectifier circuit that generates a rectified and smoothed voltage Ei corresponding to one level of the AC input voltage VAC is also referred to as an “equal voltage rectifier circuit”.
[0030]
Further, in the insulating converter transformer PIT of the power supply circuit shown in FIG. 3, in addition to the primary winding N1, the secondary winding N2, and the drive winding NB, a winding N3 is provided so as to wind up the primary winding N1.
The end of this winding N3 is connected to the positive electrode of a smoothing capacitor CiB for boost voltage generation described later. The negative electrode of the smoothing capacitor CiB is connected to the positive electrode (Ei line) of the smoothing capacitor Ci.
In the power supply circuit shown in this figure, a boost diode DB is provided. The boost diode DB has an anode connected to a connection point (Ei line) between the negative electrode of the smoothing capacitor CiB and the positive electrode of the smoothing capacitor Ci, and a cathode connected via a series connection of the controlled winding N R of the orthogonal control transformer PRT. Thus, the connection is made to the connection point between the primary winding N1 and the winding N3.
According to such a connection form, the switching output voltage obtained in the winding N3 is rectified by the boost diode DB and smoothed by the smoothing capacitor CiB, thereby generating the boost voltage VB at both ends of the smoothing capacitor CiB. A circuit will be formed. However, as described above, the controlled winding NR is inserted in series in this boost circuit.
[0031]
By providing this boost circuit, the voltage resonance type switching converter including the switching element Q1 uses the boost smoothing voltage EB obtained by superimposing the boost voltage VB on the rectified smoothing voltage Ei as an operating power source (DC input). Switching is performed as a voltage). In other words, when the boost circuit operates, the apparent DC input voltage level to be supplied to the voltage resonance type switching converter is increased, and thus, for example, a configuration in which a DC input voltage is obtained by a voltage doubler rectification operation. It is possible to increase the maximum load power that can be handled to the same extent as.
[0032]
In the configuration including the boost circuit as described above, the boost smoothing voltage EB in which the boost voltage VB is superimposed on the rectified smoothing voltage Ei is obtained at both ends of the smoothing capacitor CiB and the smoothing capacitor Ci connected in series. However, this boost smoothing voltage EB is
[Expression 1]

Can be represented by
Then, the relationship of L3 = L1 is obtained as the inductance of the winding N3 and the primary winding N1, and the rectified and smoothed voltage Ei, the drop voltage VF of the boost diode DB, and the saturation voltage V (SAT) of the switching element Q1. If the relationship of Ei >> VF, V (SAT) is established, the boost smooth voltage EB is based on the above (Equation 1),
[Expression 2]

Will be shown. In this case, in the power supply circuit shown in FIG. 3, for example, the boost smoothing voltage is changed by changing the inductance LR of the controlled winding NR of the orthogonal transformer PRT in the range of 0.1 × L1 to 1.2 × L1. EB can be varied in a range of approximately Ei to 2Ei (Ei corresponds to the rectified and smoothed voltage level obtained at both ends of the smoothing capacitor Ci).
[0033]
When the boost circuit is provided in this manner, the parallel resonant capacitor Cr on the primary side of the present embodiment is between the positive electrode of the smoothing capacitor CiB (boost smoothing voltage EB line) of the boost circuit and the collector of the switching element Q1. Connected to each other. Even in such a connection form, the resonance voltage Vcr obtained at both ends of the parallel resonance capacitor Cr is shifted to a reference potential by the boost smooth voltage EB, so that a voltage lower than 1400 V is obtained. The capacitor can be a book. That is, the size of the parallel resonant capacitor Cr can be reduced also by the configuration of the present embodiment.
[0034]
Further, on the secondary side of the insulating converter transformer PIT shown in FIG. 3, a center tap is provided in the secondary winding N2, and rectifier diodes DO1, DO2, DO3, DO4 and smoothing capacitors CO1, CO2 are shown in the figure. Are connected to each other to provide two sets of full-wave rectifier circuits, each consisting of a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2].
The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and the full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. To do. The secondary output voltage is supplied to a subsequent load (not shown).
[0035]
As shown in FIG. 4, the insulating converter transformer PIT according to the present embodiment includes an EE type core in which, for example, ferrite type E cores CR1 and CR2 are combined so that their magnetic legs face each other. The primary winding N1 (and N3, NB) and the secondary winding N2 are wound around the central magnetic leg of the core by using the bobbin B, respectively. In this embodiment, a gap G is formed with respect to the central magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient is obtained.
[0036]
In the insulating converter transformer PIT, the inductance L1 of the primary winding N1 depends on the relationship between the polarity (winding direction) of the primary winding N1 and secondary winding N2 and the connection of the rectifier diode D0 (DO1, D02 / D03, D04). And the mutual inductance M of the secondary winding N2 may be + M or -M.
For example, when the connection form shown in FIG. 5A is adopted, the mutual inductance becomes + M, and when the connection form shown in FIG. 5B is adopted, the mutual inductance becomes −M.
When this is made to correspond to the operation on the secondary side of the present embodiment described above, for example, an operation in which a rectified current flows in the rectifier diodes D01 and D03 when the alternating voltage obtained in the secondary winding N2 is positive. Can be regarded as a + M operation mode (forward method), and conversely, when the alternating voltage obtained at the secondary winding N2 is negative, the rectified current flowing through the rectifier diodes DO2 and D04 is the -M operation mode ( It can be seen that it is a flyback system). That is, in this embodiment, every time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance operates in the + M / −M mode.
[0037]
In this way, when a full-wave rectifier circuit is provided on the secondary side, the secondary side parallel resonant capacitor according to the present embodiment is as shown in the figure with respect to the rectifier circuit that generates the DC output voltage E01. Secondary-side parallel resonant capacitors C2A and C2B are connected in parallel to the rectifier diodes D01 and D02, respectively. As a result, when the rectifier diode DO1 is turned off, a parallel resonant circuit is formed by the secondary side parallel resonant capacitor C2A and the secondary winding N2 via the smoothing capacitor CO1, and when the rectifier diode DO2 is turned off, the smoothing capacitor CO2 is used. Thus, a parallel resonance circuit is formed by the secondary side parallel resonance capacitor C2B and the secondary winding N2.
Even in such a connection form, generation of radiation noise during the reverse recovery time of the secondary side rectifier diode can be suppressed by the secondary side parallel resonant capacitors C2A and C2B as in the previous embodiment. There is no need to provide the capacitor Cp.
[0038]
When the secondary side parallel resonant circuit is formed by providing the secondary side parallel resonant capacitors C2A and C2B as in the power supply circuit shown in this figure, power is supplied to the load side by the action of the secondary side parallel resonant circuit. As described above, the maximum load power is increased as compared with the case where the secondary parallel resonant capacitor is not provided.
In addition, when the full-wave rectifier circuit is connected to the secondary parallel resonant circuit as in the circuit shown in this figure, as described above, the operation modes in which the mutual inductance is both + M / −M are used. The rectified current flows alternately. That is, since the rectified output is obtained in both periods of the alternating voltage between the positive electrode and the negative electrode, the power supplied to the load side is increased accordingly, and the increase rate of the maximum load power is also improved.
[0039]
As the control circuit 1 in this case, a control current having a level corresponding to the fluctuation of the DC output voltage E01 is supplied to the control winding NC of the orthogonal transformer PRT so that the DC output voltage E01 is constant. It operates so as to vary the inductance LR of the control winding NR. Thereby, the constant voltage control by the inductance control method mentioned above is performed.
In this case, the boost smoothing voltage EB represented by (Equation 1) is controlled to be constant by varying the inductance LR, and this action makes the secondary side DC output voltage constant. It works like that.
[0040]
For example, in the configuration of the soft switching power supply circuit previously proposed by the present applicant, for example, when adopting a configuration corresponding to the AC input voltage AC100V system and the maximum load power of 150 W to 160 W or more, the voltage doubler rectifier circuit is used for AC. A rectified and smoothed voltage corresponding to a level almost twice the input voltage was obtained. As a result, the DC voltage level input to the switching converter is increased to cope with a relatively high load condition.
[0041]
On the other hand, the power supply circuit shown in FIG. 3 employs a double voltage rectification method as a rectifying and smoothing circuit that generates a DC input voltage (rectified and smoothed voltage) by increasing the maximum load power as described above. There is no need to cover the load power. As a result, as shown in FIG. 3, the configuration of a normal equal voltage rectifier circuit using, for example, a bridge rectifier circuit can be adopted.
Thereby, for example, in the power supply circuit shown in FIG. 3, the rectified and smoothed voltage Ei when the AC input voltage VAC = 144 V is about 200 V. Here, the collector-emitter voltage VCP of the switching element Q1 is generated when the switching element Q1 is turned off by the parallel resonance circuit on the primary side acting on the rectified and smoothed voltage Ei. In the circuit of FIG. As described above, the rectified and smoothed voltage Ei is approximately ½ that of the double voltage rectification. However, in the configuration shown in FIG. 3, the boost voltage VB is generated by superimposing the boost voltage VB on the rectified and smoothed voltage Ei. Therefore, the resonance voltage Vcr depends on the level of the boost smoothed voltage EB. Nevertheless, the resonance voltage Vcr can be suppressed to about 1200V.
Therefore, in the circuit shown in FIG. 3, a 1200 V withstand voltage product may be selected for the switching element Q1.
[0042]
Further, as described above, a full-wave rectifier circuit is provided on the secondary side, and the rectified current flows in both periods in which the alternating voltage of the secondary winding N2A is positive and negative, so that the secondary side resonance voltage V2 is Both the positive and negative periods are suppressed to a level equivalent to the rectified and smoothed voltage Ei.
As a result, as the rectifier diodes (DO1 to D04) that form the secondary-side full-wave rectifier circuit, it is sufficient to select a withstand voltage product that substantially corresponds to the level of the rectified and smoothed voltage Ei.
[0043]
As described above, in the circuit shown in FIG. 3, the parallel resonant capacitor Cr is small, and a low breakdown voltage product can be used for the switching element Q1 and the rectifier diode forming the secondary-side full-wave rectifier circuit. Is cheaper that much. Therefore, for example, the switching element Q1 and the rectifier diode forming the secondary-side full-wave rectifier circuit having improved characteristics (for example, the switching element Q1, the saturation voltage VCE (SAT ), Those having good characteristics such as storage time tSTG, fall time tf, and current amplification factor hFE, and those having good characteristics such as forward voltage drop VF and reverse recovery time trr in the case of a rectifier diode). Therefore, reduction of power loss can be promoted accordingly. That is, it is possible to improve the power conversion efficiency at a lower cost or substantially the same cost as compared with the configuration in which the DC input voltage is generated by the voltage doubler rectifier circuit. Further, due to the improvement of the power conversion efficiency, for example, a heat radiating plate necessary for heat dissipation of the rectifier diode of the voltage doubler rectifier circuit becomes unnecessary.
[0044]
In addition, if the switching frequency of the voltage resonance type converter is set to a high switching frequency of, for example, about 100 KHz, the above-described various parts can be reduced in size and weight. Here, if the winding N3 is selected so that the boost voltage VB is optimized in accordance with the maximum load power to be actually handled, further reduction in size and weight of various parts can be realized.
Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, the configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. In the circuit shown in FIG. 3, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit is used, and thus a set of block-type smoothing capacitors and bridge rectifier diodes can be employed. In addition, the size of the parts can be reduced. That is, in the circuit shown in FIG. 3, various parts including the insulating converter transformer PIT and the orthogonal control transformer PRT can be reduced in size. Experiments have also shown that power conversion efficiency can be improved.
[0045]
FIG. 6 is a circuit diagram showing a configuration of a power supply circuit according to the third embodiment of the present invention. The power supply circuit shown in this figure has a configuration in which a commercial AC power supply is an AC100V system and can handle a relatively heavy load having a maximum load power of 160 W or more. The same parts as those in FIG. 1 and FIG.
[0046]
In the case of the present embodiment, the rectifier circuit for obtaining the rectified and smoothed voltage Ei by inputting the AC input voltage VAC is formed by the bridge rectifier circuit Di and one smoothing capacitor Ci.
This is because, as will be described below, the load power is covered by increasing the switching output by the push-pull switching operation by the two stone switching elements. In particular, the DC input voltage is 2Ei. This is because the level is not required.
[0047]
The primary winding N1 is provided with a center tap and is divided into primary windings N1A (inductance L1A) and N1B (inductance L1B). The center tap terminal is connected to the line of the rectified and smoothed voltage Ei through a series connection of the controlled winding NR of the orthogonal control transformer PRT.
[0048]
The voltage resonance type converter shown in this figure has a separately-excited configuration having two switching elements Q1 and Q2. In this case, a MOS-FET is adopted as the switching element Q1.
In this case, the drain of the switching element Q1 is connected to the end of the primary winding N1A, and the source is grounded to the primary side ground. Further, between the drain and source of the switching element Q, a damper diode DD1 that forms a current path for a period when switching off is connected in parallel.
The drain of the switching element Q2 is connected to the end of the primary winding N1B, and the source is grounded to the primary side ground. A damper diode DD2 is also connected in parallel between the drain and source of the switching element Q.
[0049]
The primary side parallel resonant capacitors of the present embodiment are provided with parallel resonant capacitors Cr1 and Cr2 corresponding to the switching elements Q1 and Q2, respectively.
The parallel resonance capacitor Cr1 is connected so as to be inserted between the collector of the switching element Q1 and the positive electrode (rectified smoothing voltage Ei line) of the smoothing capacitor Ci, thereby causing the capacitance of the parallel resonance capacitor Cr1 and the inductance L1A. , And a combined inductance of the inductance LR form a parallel resonance circuit in which the switching operation of the switching element Q1 is a voltage resonance type.
Similarly, the parallel resonance capacitor Cr2 is connected so as to be inserted between the collector of the switching element Q2 and the positive electrode of the smoothing capacitor Ci (rectified smoothing voltage Ei line), thereby the capacitance of the parallel resonance capacitor Cr2. And a combined inductance of the inductance L1B and the inductance LR form a parallel resonance circuit in which the switching operation of the switching element Q2 is a voltage resonance type. In the conventional configuration, the parallel resonant capacitors Cr1 and Cr2 are connected in parallel to the collector-emitter of the switching elements Q1 and Q2, respectively.
The switching output point of the switching elements Q1, Q2 is the center tap of the primary winding N1.
That is, in this embodiment, two switching circuit systems for driving the switching elements Q1 and Q2 are provided.
[0050]
The oscillation drive circuit 2 is activated by the activation resistor RS, and for example, oscillates and outputs a frequency signal corresponding to the switching frequency fs = 100 KHz, and generates a switching drive signal (drive voltage) based on this frequency signal. It is generated and applied to the gates of the switching elements Q1, Q2. At this time, the switching drive signals applied to the gates of the switching elements Q1 and Q2 are signals having opposite polarities that are inverted from each other.
[0051]
Here, assuming that the switching elements Q1 and Q2 perform the switching operation, the switching elements Q1 and Q2 perform the switching operation at the timing of turning on / off alternately at, for example, the switching frequency fs = 100 KHz.
In other words, when both switching elements Q1 and Q2 perform switching, a so-called push-pull switching operation (push-pull operation) is performed. For example, a single-switching operation by a single switching element (single-end operation) It is possible to supply larger power to the load side than in the case of. Thereby, in this Embodiment, it corresponds to the conditions of the heavy load more than the maximum load electric power 160W.
[0052]
Further, the control circuit 1 shown in this figure compares the DC voltage output (EO1) on the secondary side with the reference voltage and responds to the error, for example, similarly to the control circuit 1 shown in FIGS. It is configured as an error amplifier that supplies the direct current as a control current to the control winding NC of the orthogonal control transformer PRT.
[0053]
For example, when the secondary side DC output voltage E02 varies with the variation of the AC input voltage VAC or the minimum load power, the control circuit 1 changes the control current flowing in the control winding NC within a predetermined range, for example. Thereby, the inductance LR of the controlled winding NR is changed within a predetermined range.
[0054]
In this embodiment, as described above, the controlled winding NR is inserted so as to form a parallel resonant circuit on the switching element Q1 side and a parallel resonant circuit on the switching element Q2 side. Therefore, when the inductance LR changes, the resonance condition of the resonance circuit changes with respect to the fixed switching frequency during the switching operation of the switching elements Q1 and Q2. The output level of the DC output voltage (EO1, EO2) on the secondary side changes according to the change in the resonance condition. This makes the secondary DC voltage (EO1, EO2) constant. It will be. That is, even in this case, the constant voltage control operation is performed by the inductance control method.
[0055]
Also in the configuration shown in FIG. 6, as in the first and second embodiments, the ceramic capacitors Cp for reducing the size of the parallel resonant capacitors Cr1 and Cr2 on the primary side and absorbing the radiation noise on the secondary side. Is omitted.
[0056]
In addition, this invention is not limited to the structure shown to each figure as said 1st-3rd embodiment, A change is possible.
For example, in the configuration shown in FIGS. 3 and 6, for example, the secondary side rectifier circuit can be configured to include a voltage doubler full-wave rectifier circuit. If an attempt is made to obtain a DC output voltage at the same level as the full-wave rectifier circuit, the number of turns of the secondary winding N2 can be reduced to ½. As the switching element, not only bipolar transistors but also other types of switching elements such as MOS-FET transistors may be employed.
[0057]
【The invention's effect】
As described above, the voltage resonance type converter according to the present invention is configured such that the resonance capacitor of the primary side resonance circuit for switching the voltage operation to the voltage resonance type is connected between the positive line of the DC input voltage and the switching output terminal of the switching means. I try to insert it. As a result, the reference potential of the resonance voltage generated at both ends of the resonance capacitor by the action of the primary side resonance circuit is shifted by the DC input voltage level, and accordingly, the resonance voltage pulse generated at both ends of the resonance capacitor when the switching element is off. It becomes possible to reduce the level of.
As a result, in an actual circuit, it is possible to fit within the breakdown voltage of a capacitor (for example, a polypropylene film capacitor), which is the limit of production, and it is not necessary to form a resonant capacitor by connecting two or more capacitors in series. That is, as the resonant capacitor, the number of series connections can be reduced to, for example, one capacitor, and the resonant capacitor can be reduced in size and weight and the cost can be reduced accordingly.
[0058]
Further, a secondary side parallel resonant capacitor that forms a secondary side parallel resonant circuit together with an inductance component of the secondary side winding is connected in parallel to a secondary side rectifier diode that forms a secondary side rectifier circuit. In order to eliminate the radiation noise that occurs during the reverse recovery time of the secondary rectifier diode, the ceramic capacitor that has been connected in parallel to the secondary rectifier diode in order to suppress the radiation noise is omitted. It becomes possible to do.
As described above, the present invention has an effect that it is possible to promote reduction in size and weight and cost of the voltage resonance type switching power supply circuit.
[Brief description of the drawings]
FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.
FIG. 2 is a waveform diagram showing an operation of a main part of the power supply circuit according to the embodiment.
FIG. 3 is a circuit diagram showing a configuration example of a power supply circuit according to a second embodiment of the present invention.
FIG. 4 is a cross-sectional view showing a configuration of an insulating converter transformer.
FIG. 5 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.
FIG. 6 is a circuit diagram showing a configuration example of a power supply circuit according to a third embodiment of the present invention.
FIG. 7 is a circuit diagram showing a configuration example of a power supply circuit as a conventional example.
8 is a waveform diagram showing an operation of a main part of the power supply circuit shown in FIG.
[Explanation of symbols]
1 control circuit, 2 oscillation circuit, 3 drive circuit, Ci, CiB smoothing capacitor, Cr, Cr1, Cr2 parallel resonance capacitor, C2, C2A, C2B (secondary side) parallel resonance capacitor, Di bridge rectifier circuit, DO1, DO2, D03, D04 Rectifier diode, PIT isolated converter transformer, N1, N1A, N1B primary winding, N2 secondary winding, PRT orthogonal control transformer, NC control winding, NR controlled winding, Q1, Q2 switching element, DB Boost diode, N3 (boost) winding,

Claims (4)

Rectifying and smoothing means having a rectifying diode and a smoothing capacitor that generates a rectified and smoothed voltage by inputting a commercial AC power supply and rectifies and smoothes and outputs it as a DC input voltage;
Switching means adapted to intermittently output the DC input voltage to the primary winding of the insulation converter transformer;
An orthogonal control transformer in which a controlled winding is connected between a primary winding of the insulating converter transformer and a positive electrode of the smoothing capacitor;
Formed by the inductance component of the primary winding of the insulating converter transformer and the controlled winding, and the capacitance of the resonant capacitor inserted between the output terminal of the switching means and the positive electrode of the smoothing capacitor , A primary side parallel resonant circuit in which the operation of the switching means is a voltage resonant type;
DC output voltage generating means for generating a secondary DC output voltage from an alternating voltage obtained in the secondary winding of the insulating converter transformer by including a secondary rectifier diode element and a secondary smoothing capacitor;
A secondary side resonance circuit formed on the secondary side by the inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side parallel resonance capacitor;
The controlled winding and the controlled winding by supplying a variable control current in response to the level of the DC output voltage to have been the orthogonal control transformer control winding so that the winding directions are orthogonal Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage by changing the inductance of
A voltage resonance type switching power supply circuit comprising:   2. The voltage resonant switching power supply circuit according to claim 1, wherein the secondary side parallel resonant capacitor forming the secondary side resonant circuit is connected in parallel to the secondary side rectifier diode element.   2. The voltage resonance type switching power supply circuit according to claim 1, wherein two sets of the switching means are provided and are configured to intermittently output a DC input voltage by a push-pull operation.   2. The voltage resonance type switching power supply circuit according to claim 1, wherein the DC output voltage generating means is formed with a full-wave rectifier circuit.

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