JP3061034B2 - Thermal flow meter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a thermal type flow meter, and is used, for example, for measuring an intake air amount of an internal combustion engine.

[0002]

2. Description of the Related Art As a conventional hot-wire flowmeter, a heat-generating temperature-sensitive resistor is installed in a fluid flow path, and the amount of electricity is controlled by feedback so as to maintain the temperature of the heat-generating temperature-sensitive resistor at a predetermined value. In order to measure the flow rate of the fluid based on the amount of current, there is a known configuration. There is also known a device in which a temperature-sensitive resistor that generates little heat is provided in a fluid flow path, and a bridge circuit is formed together with the above-mentioned heat-generating temperature-sensitive resistor to compensate for a change in fluid temperature.

[0003] For example, a hot wire type flow meter disclosed in Japanese Patent Application Laid-Open No. 55-43447 is known. In this prior art, there are two operational amplifiers (op-amps): an operational amplifier (op-amp) for controlling the amount of current supplied to the heat-generating temperature-sensitive resistor, and an operational amplifier for controlling the amount of current supplied to the temperature-sensitive resistor for temperature compensation. By using, stable flow rate measurement is enabled.

[0004]

According to the above prior art, the flow rate in the fluid flow path can be measured. But,
In the circuit as described above, the temperature-sensitive resistor that generates heat generates heat when energized, and also generates a temperature-sensitive resistor for temperature compensation, which is not originally scheduled to generate heat, with energization. There is a problem that a measurement error occurs due to heat generation of the compensation temperature sensitive resistor. Such a measurement error can be suppressed by lowering the voltage applied to the temperature compensation resistor for temperature compensation and reducing the current flowing therethrough.

That is, in the configuration described in Japanese Patent Application Laid-Open No. 55-43447, the temperature-sensitive resistor for temperature compensation is increased by increasing the voltage dividing ratio of a voltage-dividing circuit connected in parallel with the temperature-sensitive resistor that generates heat. Voltage to be applied can be reduced. But,
When the voltage dividing ratio of the voltage dividing circuit is increased, there is a problem that the influence of the error of the input offset voltage of the operational amplifier greatly appears in the measurement error. The input offset voltage of the operational amplifier includes a drift component due to a change in temperature, a component due to a change in the diameter of the circuit, and the like.

SUMMARY OF THE INVENTION The present invention has been made in view of the above-mentioned conventional problems, and has as its object to provide a thermal type flow meter having a small measurement error.

[0007]

In order to achieve the above-mentioned object, the present invention provides a first temperature-sensitive resistor provided in a fluid flow path, and provided in series with the first temperature-sensitive resistor. A first power adjustment circuit for adjusting power supplied from a power supply, a second temperature-sensitive resistor provided in a fluid flow path, and a second temperature-sensitive resistor provided in series with the second temperature-sensitive resistor; A second power adjustment circuit that adjusts power supplied to the body; and an operational amplifier to which detection signals of the first temperature-sensitive resistor and the second temperature-sensitive resistor are input. A first power control circuit that controls the first power adjustment circuit to maintain a temperature difference from the second temperature sensitive resistor at a predetermined value, and adjusts power supplied to the first temperature sensitive resistor.
A control circuit, and controlling the second power adjusting circuit so that power supplied to the second temperature-sensitive resistor is reduced by a predetermined ratio with respect to power supplied to the first temperature-sensitive resistor; A second control circuit that adjusts power supplied to the resistance thermometer, wherein the second control circuit applies a voltage applied to both ends of the second resistance thermometer to both ends of the first resistance thermometer. Voltage (VB
(R-VM) is adopted so as to be about 1/10 to 1/20.

[0008] The second control circuit includes a voltage dividing circuit (10, 11) for detecting a voltage (VBR-VM) applied to both ends of the first temperature sensitive resistor, and a detection signal of the voltage dividing circuit. And an operational amplifier to be input, and the voltage divider circuit
The voltage applied to both ends of the second temperature-sensitive resistor is set to 1/1 of the voltage (VBR-VM) applied to both ends of the first temperature-sensitive resistor.
It can be configured by adopting the technical means of the thermal type flow meter according to claim 1, wherein the value is set to about 0 to 1/20.

[0009]

The operation of the present invention will be described. In the configuration of the present invention, the power supplied to the first temperature-sensitive resistor is controlled by the first power adjustment circuit so that the temperature difference between the first temperature-sensitive resistor and the second temperature-sensitive resistor is maintained at a predetermined temperature difference. And the first control circuit performs feedback control. Thus, the flow rate of the fluid is measured by detecting the supplied power. In addition, the second control circuit controls the second power adjustment circuit so that the power supplied to the second temperature-sensitive resistor is reduced by a predetermined ratio with respect to the power supplied to the first temperature-sensitive resistor. This makes it possible to stabilize the power supplied to the second temperature sensitive resistor, reduce errors in the second temperature sensitive resistor, and compensate for a flow rate measurement error due to a fluid temperature change with high accuracy. Further, the second control circuit converts the voltage applied to both ends of the second temperature-sensitive resistor into the voltage (VB) applied to both ends of the first temperature-sensitive resistor.
R-VM) is set to about 1/10 to 1/20. By adopting such a value, the voltage applied to both ends of the second temperature-sensitive resistor is suppressed to a sufficiently low level, and the measurement error accompanying the heat generation of the second temperature-sensitive resistor is suppressed. Moreover, since the voltage applied to both ends of the second temperature-sensitive resistor is not excessively reduced, the influence on the measurement error of the input offset voltage of the operational amplifier included in the control circuit can be suppressed. That is, by employing the above numerical range, it is possible to suppress both the measurement error due to the heat generation of the second temperature sensitive resistor and the measurement error due to the input offset voltage of the operational amplifier.

The voltage applied to both ends of the second temperature-sensitive resistor is equal to the voltage (VBR-V) applied to both ends of the first temperature-sensitive resistor.
M) can be set by a voltage divider circuit that detects

[0011]

Embodiments of the present invention will be described below. In this specification, a circuit including a differential amplifier circuit and an output circuit thereof is referred to as an operational amplifier or an operational amplifier. FIGS. 1 and 2 are circuit diagrams of the first embodiment, and show a series of circuit diagrams extending from the left end AA line in FIG. 1 to the right end AA line in FIG. In the first embodiment, a temperature-sensitive resistor is provided in a bypass passage provided in an intake passage of an internal combustion engine, and the amount of intake air of the internal combustion engine is measured.

1 and 2, reference numeral 1 denotes a vehicle-mounted battery,
Reference numeral 2 denotes a noise removing capacitor, and reference numeral 3 denotes a resistor. Reference numeral 4 denotes a protection diode for protecting the below-described temperature-sensitive resistor 7 from burning when the power supply line of the device has a negative voltage with respect to the ground when the terminals of the battery are reversely connected.
Reference numeral 5 denotes a two-stage power transistor, which constitutes an emitter follower circuit. Reference numeral 6 denotes a resistor provided at the base of the power transistor 5. Reference numerals 7 and 8 denote temperature-sensitive resistors, which are arranged in the air flow path. In addition, the temperature-sensitive resistor 7,
Reference numeral 8 denotes an identical bobbin formed by winding a platinum wire around an insulating bobbin made of ceramic or the like, having substantially the same resistance value, and having substantially the same heat capacity and shape. The first fixed resistor 9 is
The current flowing through the first temperature-sensitive resistor 7 is converted into a voltage and detected. The resistors 10 and 11 constitute a voltage dividing circuit for dividing the voltage between both ends of the first temperature-sensitive resistor 7. Reference numeral 12 denotes resistors 13 and 14, capacitors for removing noise, and reference numerals 15 and 16 denote resistors. Reference numeral 17 denotes a second fixed resistor, and reference numeral 18 denotes a third fixed resistor. 19 and 20 are noise removing capacitors, 21 is a circuit protection resistor, 22 and 23 are resistors, 2
Reference numerals 4 and 25 denote an output voltage Vre of a reference voltage source 104 described later.
It is a voltage dividing resistor for dividing f.

Reference numeral 100 denotes a monolithic IC, and reference numerals 101, 102, and 103 denote Zener diodes for protecting the monolithic IC 100 against overvoltage of the battery voltage. 104 is a reference voltage source,
This is a band gap type reference voltage source circuit. Reference numeral 105 denotes an operational amplifier (hereinafter referred to as an operational amplifier), reference numeral 106 denotes a transistor, reference numerals 107 to 110 denote diodes, reference numerals 111 and 112 denote resistors, reference numeral 113 denotes diodes, and reference numerals 105 to 113 denote an output voltage Vref of the reference voltage source 104. A constant voltage power supply circuit is configured as a reference.

Reference numeral 200 denotes a range of the operational amplifier. The operational amplifier 200 includes resistors 201 and 202, a pair of PNP transistors 203 and 204, a pair of NPN transistors 205 and 206, a constant current source 207, a pair of PNP transistors 208 and 209, and a pair of constant current sources 210 and 21.
1, current source 212, pair of transistors 213, 21
4. Phase compensation capacitor 215, resistors 216 and 21
7, 218 are provided with constant current sources 217, 218, an NPN transistor 219 for voltage inversion amplification, resistors 220, 221 and NPN transistors 222, 223. The operational amplifier 200 has a voltage follower circuit in which the emitter of the transistor 223, which is the output terminal of the operational amplifier 200, is connected to the base of the transistor 203 via the resistor 201. Further, the current sources 207 and 21
0, 211, 212, 217 and 218 are power supply voltages (V
A constant current that does not depend on the change in B) is supplied, and a current mirror circuit using a PNP transistor is used.

Reference numeral 300 denotes a range of the operational amplifier. The operational amplifier 300 includes resistors 301 and 302, a pair of PNP transistors 303 and 304, resistors 305 and 306, a constant current source 307, and a pair of NPN transistors 308 and 30.
9, a pair of PNP transistors 310 and 311, a pair of constant current sources 312 and 313, a constant current source 314, a pair of transistors 315 and 316, and a capacitor 31 for phase compensation
7, a resistor 318, constant current sources 319 and 320, an NPN transistor 321 for voltage inversion amplification, PNP transistors 322 to 326, and constant current sources 327 to 331. Then, the resistors 305 and 306 divide the reference voltage Vref of the reference voltage source 104, and
The offset voltage is applied to the operational amplifier 300 by providing the resistor 305 to the emitter 08. The current sources 312, 313, 314, 319, and 320 supply a constant current that does not depend on a change in the power supply voltage (VB).
Specifically, it is composed of a current mirror circuit using PNP transistors.

Reference numerals 400 and 500 denote operational amplifiers. FIG.
Among the circuits shown in FIG. 2, a circuit composed of operational amplifiers 200 and 300 and elements 5 to 19 is a bridge circuit and a control circuit thereof. And operational amplifiers 400 and 500
And a circuit composed of 20 to 25 elements, a voltage VM detected as a voltage correlated with the flow rate from the bridge circuit.
Is an output circuit for amplifying and outputting.

Next, the operation of each part of the circuit according to the above configuration will be described. When power is supplied from the battery 1 to the device 1,
The reference voltage source 104 outputs a reference voltage Vref. This reference voltage is made by a band gap type circuit,
This is a high-precision reference voltage having excellent temperature characteristics of V. This V
With reference to the ref voltage, a constant voltage VCC is formed by a constant voltage power supply circuit composed of 105 to 113 and supplied as power for other circuits. In this constant voltage circuit, VCC is given by Equation 1 below.

[0018]

(Equation 1)

In the equation 1, R111, R111
Reference numeral 112 denotes resistance values of the resistors 111 and 112, respectively.
F denotes diodes 107 to 110 and 113 connected in series.
Shows the forward voltage drop of This forward voltage drop VF is-
There is a temperature dependence of about 2 mV / ° C., and therefore, as can be seen from the above equation, VCC is given a temperature dependence. Next, the operation of the bridge circuit will be described. First, activation of the bridge circuit when power is supplied to the device will be described. When power is supplied to the device, a constant voltage Vcc is supplied. In this embodiment, the transistors 322 through 322 are connected to the output of the operational amplifier 300 that drives the power transistor 5.
26 is provided, and the transistor 3
The voltage of the emitter 26 is increased. Therefore, the voltage at the emitter terminal of the transistor 326 is
The voltage is within a predetermined range regardless of the voltage applied to the bases of the 0 input transistors 303 and 304. Therefore, immediately after power is supplied to the device, the voltage of the emitter terminal of the transistor 326 becomes a voltage in a predetermined range regardless of the input of the operational amplifier 300.

The minimum value VE (MIN) and the maximum value VE (MA) of the voltage VE at the emitter terminal of the transistor 326 are shown.
X) is expressed by the following Expressions 2 and 3, where VBE1 is the forward voltage drop between the emitter and the base of each of the transistors 322 to 326.

[0021]

## EQU2 ## VE (MIN) ≒ 5 × VBE1

[0022]

## EQU3 ## VE (MAX) に 5 × VBE1 + Vcc When a voltage is generated at the emitter terminal of the transistor 326 as described above, the transistor 5 forming the emitter follower circuit conducts, and the emitter voltage of the transistor 5 is applied to the input of the operational amplifier 300. Regardless of the voltage, the voltage is within a predetermined range.

The emitter voltage VBR of this transistor 5
The minimum value VBR (MIN) and the maximum value VBR (MAX) of are expressed by the following Expressions 4 and 5.

[0024]

VBR (MIN) = 5 × VBE1-2 × VBE2

[0025]

VBR (MAX) = 5 × VBE1 + VCC-2
× VBE2 In Equation 4, the forward voltage drop between the base and the emitter of each transistor of the two-stage Darlington-connected transistor 5 is defined as VBE2,
Is 2 × VBE2
There is.

Here, VBE1 and VBE2 at normal temperature
Is approximately 0.7 (V), VBR is 2.1 to VC
The voltage is in the range of C + 2.1 (V). By the above operation, a voltage in a predetermined range is applied to the bridge circuit including the temperature-sensitive resistors 7 and 8 regardless of the input state of the operational amplifier 300, and the voltages VM and VK to be input to the operational amplifier 300 are generated. At this time, since the power is supplied immediately, the first temperature-sensitive resistor 7 has not yet generated heat, so that VM> VK. Therefore, the output of the operational amplifier 300 (the emitter voltage of the transistor 326) is VCC + 5 × VBE1, and the emitter voltage of the transistor 5 is 5 × VBE1 + V
The maximum voltage is CC-2 × VBE2. As a result, the maximum current flows through the temperature-sensitive resistor 7, the temperature-sensitive resistor 7 generates heat, the resistance value increases, and the voltage VM decreases. And
The operational amplifier 300 controls the transistor 5 so that VM = VK.

As described above, in this embodiment, since the maximum value of the voltage at the emitter terminal of the transistor 5 is limited, the voltage VBR at the emitter terminal becomes excessive even if the power supply voltage VB of the battery fluctuates. Never rise. For example, even if the battery voltage becomes much higher than the voltage in normal use, or VM or VK takes an abnormal value due to some disturbance noise,
The voltage VBR does not rise excessively, and the temperature sensitive resistor is protected.

The forward voltage drop VBE between the base and the emitter of the transistor is -2 (mV / °
It has a temperature characteristic of about C). Therefore, the voltage VB
R is approximately 3 as shown in Equations 4 and 5 above.
× VBE temperature characteristics. Therefore, in this embodiment, the diodes 107 to 110, 11
3, the voltage VCC is +6 (mV / ° C).
A degree of temperature characteristic is given to cancel a change in the maximum value of the voltage VBR due to the temperature characteristic of 3 × VBE.
As a result, the maximum value of voltage VBR can be maintained substantially constant regardless of the temperature and power supply voltage VB.

The maximum value of the voltage VBR is set to a value which is somewhat higher than the voltage required at the time of the maximum flow rate of the detection target and which protects a bridge circuit such as a temperature-sensitive resistor. Next, the operation after the activation of the bridge circuit will be described.
The main components that make up the bridge circuit are the operational amplifier 20
0, 300, transistor 5, temperature-sensitive resistor 7, 8, fixed resistor 9-11, 17, 18; Hereinafter, the resistance value of the first temperature-sensitive resistor 7 is RH, the resistance value of the second temperature-sensitive resistor 8 is RK, the resistance value of the first fixed resistor 9 is R9, and the first fixed resistor 17 is The resistance value of R17 and the third fixed resistor 18
Is R18, and the resistance values of the fixed resistors 10, 11 are R
Expressed as 10, R11. Also, VBR, V in the figure
M and VK represent the voltages of the portions marked with the symbols.

When the bridge circuit is energized by the above-described operation at the time of starting, input voltages of the operational amplifiers 200 and 300 are generated, and the input voltage of the operational amplifier 300 satisfies the following equation (6). On the other hand, the input voltage of the operational amplifier 200 satisfies the condition of Expression 7 below.

[0031]

[Formula 6] VM + VOS3 = VK

[0032]

(Equation 7)

By substituting equation (7) into equation (6), the following equation (8) is derived, and the voltage VBR expressed by the following equation (9) is obtained.
Substituting the above and rearranging leads to Equation 10 below. Equation 10 is an equilibrium condition equation of the bridge circuit. Note that R
H≪R10 + R11.

[0034]

(Equation 8)

[0035]

(Equation 9)

[0036]

(Equation 10)

In the above equation, VOS3 is the operational amplifier 300
Offset voltage. The voltage VOS3 is obtained by changing the voltage Vref output from the reference voltage source 104 to the resistors 306 and 30.
5 and an internal offset voltage (referred to as VOS3D) generated in the internal circuit of the operational amplifier 300 other than the ΔV in the internal circuit of the operational amplifier 300, and is represented by the following equation (11).

[0038]

VOS3 = ΔV + VOS3D The voltage VOS3D is mainly determined by the transistors 303, 304, and 3
The internal offset voltage is caused by the mismatch between the pairings 08 and 309, the pairs 310 and 311 and the mismatch between the current sources 312 and 313. The voltage VOS2 is an internal offset voltage of the operational amplifier 200, and
Occurs for the same reason as the internal offset voltage VOS3D.

In the following description, VOS2 and VOS3 are assumed to be 0 (V) to simplify the operation. In the above equation 10, VOS2 and VOS3 are set to 0.
When the equations are arranged as (V), the bridge equilibrium condition equation becomes the following equation 12.

[0040]

(Equation 12)

The right side of the above equation is the resistance value RK of the second temperature sensitive resistor 8 installed in the air flow path and the resistance value R1 of the fixed resistor.
7, R18, R10 and R11. Here, the temperature-sensitive resistors 7 and 8 installed in the air passage have substantially the same resistance, and
Although the temperature sensing resistors have the same heat capacity, the voltage applied thereto is adjusted so that the second temperature sensing resistor 8 does not generate heat. In this embodiment, the voltage applied to both ends of the second temperature-sensitive resistor 8 is changed by the voltage dividing circuit of the resistors 10 and 11 and the operational amplifier 200 to the voltage (VBR-VM) applied to both ends of the first temperature-sensitive resistor 7. ) Is set to be about 1/10 to 1/20. Therefore, the right side of the above equation 12 is the second
Is substantially equal to the temperature of the air flowing through the air flow path, so that the value is determined by this air temperature.

On the other hand, RH / R9 on the left side of Expression 12 is R
Since 9 is the first fixed resistor, the value is determined by the resistance value RH of the first temperature-sensitive resistor 7. The first temperature-sensitive resistor 7 is provided in the air flow path, and the current flowing therethrough is expressed by I
Then, the power of I2 RH is consumed and heat is generated. Since the generated power I2 RH is radiated to the air flowing through the air flow path, the amount of heat taken by the air changes depending on whether the flow rate of the air flowing through the flow path is large or small. Therefore, the temperature changes in accordance with the air amount, and the resistance value RH also tends to change. However, the operational amplifier 300 controls the amount of current supplied from the transistor 5 so that the resistance value RH of the first temperature-sensitive resistor 7 does not change. . That is, by changing I according to the air flow rate, I2 RH is changed so that RH is always controlled to a predetermined resistance value.

More specifically, I is changed by changing VBR in the circuit diagram. Therefore, the current I is a value having a correlation with the air flow rate. In this embodiment, the voltage drop of the first fixed resistor 9 (I × R
By detecting 9), a voltage correlated with the air flow rate is detected. This voltage (VM) is amplified by an output circuit described later, and is output as a flow signal voltage to a fuel injection amount control device (not shown).

Next, the role of the second temperature sensitive resistor 8 will be described in detail. As described above, the second temperature-sensitive resistor 8 is
It is premised that self-heating is not substantially generated (self-heating is preferably about 1 ° C. or less), and the temperature-sensitive resistor 8 measures the temperature of air flowing through the air flow path. The resistance values of the temperature sensitive resistors 7 and 8 have a temperature dependency of 3800 ppm / ° C. In this embodiment, the temperature-sensitive resistor 8 compensates for a change in the temperature of the air flowing through the flow path, and obtains a voltage (VM) having a correlation with the flow rate regardless of the temperature of the flowing air. ing.

More specifically, the temperature of the first temperature-sensitive resistor 7 may be always higher than the temperature of the second temperature-sensitive resistor 8 by a certain temperature. This is because R17, R18
Can be easily realized if R17 and R18 are set by setting and solving simultaneous equations for two points of the specific temperature, using as an unknown number. Also, the temperature difference can be given to the temperature difference between the first and second temperature-sensitive resistors according to the temperature of the air flowing through the air flow path, and the temperature dependency can be easily given by the above-described resistance values R17 and R18. Can also.

Next, the operation of the circuit including VOS2 and VOS3 will be described. When the voltage VM having a correlation with the flow rate is obtained by modifying the bridge equilibrium condition expression shown in Expression 10 above, the following Expression 13 is obtained, and the above-mentioned VOS3 = ΔV + V
Formula 14 is obtained by substituting OS3D.

[0047]

(Equation 13)

[0048]

[Equation 14]

The following Expression 15 is obtained by rearranging Expression 14 with respect to the offset voltage.

[0050]

(Equation 15)

The expression in parentheses on the second page on the right side of Expression 15 is the total offset voltage VOS in the circuit configuration of this embodiment.
It is. As can be seen from Equation 15, the operational amplifier 300
And the internal offset voltage of the operational amplifier 200 act to cancel each other. That is, VOS3D and VOS2 are ideally 0 (V), but in reality, an offset voltage always occurs due to a mismatch between the pair elements in the operational amplifier. Therefore, in this embodiment, VOS3D and VOS2 are offset. For this reason, in this embodiment, in the monolithic IC 100,
The operational amplifiers 200 and 300 are formed on the chip, and VOS2 ≒ VOS3D is set by arranging the operational amplifiers 200 and 300 on the chip and optimizing the element size. As a result, the total offset voltage VOS is set to the ideal VOS = Δ
V can be approached.

The temperature drift of the internal offset of the operational amplifier thus produced is also reduced by VOS2 and VOS3D.
Is almost the same. Here, R on the second page on the right side of Expression 15
18 / RK + R17 + R18 is substantially determined by the voltage division ratio of the resistors 10 and 11, but is 0.7 to 0 when the self-heating of the second temperature-sensitive resistor 8 is suppressed to almost negligible level (self-heating 1 ° C. or less). .9.

The total offset voltage VOS is necessary for stable operation of the bridge circuit, and must be a positive voltage. That is, when VOS is negative, positive feedback is applied in view of the relationship between the input and output of the operational amplifier 300, and the operation becomes unstable, and an oscillation phenomenon occurs. For the above reasons, VOS needs to be a positive voltage, but the magnitude of this positive VOS affects the responsiveness of the bridge circuit. Specifically, the response of the voltage VM, which is the flow signal when the amount of air flowing through the air flow path changes abruptly, changes according to the value of the total offset voltage VOS. If the total offset voltage VOS is large, the negative feedback of the bridge circuit becomes strong, and the response becomes slow. Conversely, the smaller the VOS, the weaker the negative feedback and the faster the response. If the response is made faster, ringing or the like occurs during a transition, which is not preferable. If the response is slow, a problem arises because the flow rate during the transition cannot be measured. Therefore, it is necessary to optimize the total offset voltage VOS, and this VOS is usually set to several mV. For the above reasons, it is desirable that VOS be a predetermined voltage value with as little variation as possible.
In this embodiment, the offset voltage VO of the operational amplifier 200 which is an error element of the total offset voltage shown in Expression 15
S2 and the internal offset voltage VOS3 of the operational amplifier 300
D acts to cancel each other, so that the total offset voltage V
OS errors are minimized.

In this embodiment, the current flowing through the first temperature-sensitive resistor 7 and the resistors 9, 10, and 11 is obtained from the power supply voltage VB via the transistor 5. Thereby, the minimum operating voltage can be reduced as compared with the case where the current flowing through the first temperature-sensitive resistor 7 is obtained through the resistor 3. Further, in this embodiment, the second temperature-sensitive resistor 8 and the resistors 17, 1
8 is also supplied from the emitter of the transistor 5. That is, the collector of the output transistor 223 of the operational amplifier 200 is connected to the emitter of the transistor 5 via the resistor 15, and
The emitter 23 is connected to the second temperature-sensitive resistor 8.

Here, the current supplied to the second temperature-sensitive resistor 8 is several mA to several tens mA, and this current is supplied to the constant voltage source V
When the current is supplied from the CC, the current flowing through the resistor 3 increases, and the voltage drop at the resistor 3 becomes large, thereby causing a problem that the minimum operating voltage at which the circuit can operate becomes high. If the resistance of the resistor 3 is reduced to avoid this, the capacity of the Zener diodes 101, 102, and 103 for overvoltage protection needs to be increased, and the chip size of the monolithic IC 100 becomes large, the cost increases, and the like. Problems arise.

However, according to the above configuration of this embodiment,
Since the current of the second temperature-sensitive resistor 8 is obtained from the power supply voltage VB via the transistor 5, the above-described problem does not occur. The resistor 15 is connected to the transistor 22
3 is a resistor for reducing power consumption, and may be removed to form a circuit. As can be seen from the above description of operation, the power consumption of the first temperature-sensitive resistor 7 is
18, a feedback control is performed by a control circuit including the operational amplifiers 200 and 300, the transistor 5, and the like. The voltage control circuit described below is interposed in the feedback control loop.

In the operational amplifier 200, the transistor 20
5, 206, 208, and 209, which amplifies the difference voltage between the base terminal voltages of the transistors 203 and 204, and applies the amplified voltage to the collector terminal of the transistor 209. A voltage is output and usually has an amplification degree of about 60 (dB). Also,
The transistor 221 performs voltage inversion amplification. These 2211 transistors typically have a voltage amplification of about 60 (dB). The power supply for these two amplifier circuits is derived from the constant voltage VCC of the constant voltage power supply to the current sources 210 to 21.
1 and 217. Operational amplifier 200
Other transistors 203, 204, 213, 2 in
14, 222 and 223 all constitute an emitter follower circuit and have no voltage amplification effect (the voltage amplification degree is 1
Times).

Next, the operational amplifier 300 is
Transistors 308, 309, 310, 311
Constitute a differential amplifier circuit. In addition, the transistor 321
Constitute a voltage inversion amplifier circuit. These power supplies are connected to the constant voltage V
Supplied by CC. Other transistors 30
3, 304, 315, 316, 322 to 326 constitute an emitter follower circuit, and do not have a voltage amplification effect.

The power transistor 5 forms an emitter follower circuit, and has no voltage amplification function. Therefore, since the power supply of the voltage amplifier circuit interposed in the feedback loop is all supplied from the constant voltage VCC of the constant voltage power supply, even if the voltage VB of the power supply of the device fluctuates, the voltage VBR controlled according to the constant voltage VCC changes. It does not fluctuate. Next, battery 1
Will be described. That is, the minimum (V) of the voltage VB of the battery 1 with respect to the VBR voltage required for measuring the flow rate is described. Resistor 3
, The voltage drop voltage at the resistor 6 is V2, the forward voltage drop between the base and the emitter of the subsequent transistor of the Darlington power transistor 5 is VBE1, and the forward voltage drop between the base and the emitter of the previous transistor is Assuming that the voltage is VBE2 and the minimum value of the voltage drop voltage of the current source 331 is VCE (MIN), the minimum voltage value required for the flow rate measurement is represented by the following equation (16).

[0060]

VB (MIN) = VBR + VBE2 + VBE
1 + V2 + VCE (MIN) + V1 Note that VCE (MIN) indicates the minimum value of the voltage drop of the current source 331, and indicates the lowest voltage that can supply a base current necessary for normally operating the transistor 5.

Here, a detailed circuit of the constant current source is shown in FIG. PNP transistors 327 to 331 in FIG. 3 correspond to the current sources 327 to 331 in FIGS. The current source circuit shown in FIG. 3 is a general current mirror circuit, and the basic current T1 is determined by the following equation (17).

[0062]

[Equation 17]

The VEB 350 is a transistor 350
VEB 351 is the emitter-base voltage drop of the transistor 351;
R352 is the resistance value of the resistor 352. In FIG.
The current I1 flows as the collector current of the transistor 350, and the transistors 327 to 331 respectively supply the base current I
1 is mirrored and output. Transistors 327 to 331
When the same element is used for the transistor 350 and the transistors 327 to 332 when a circuit is formed on the same chip of the bipolar IC, the output current of each transistor becomes substantially the same as the basic current I1. The current sources 327 to 330 in the present embodiment are current sources having the same current value, and the current source of the current source 331 is another current source 32 for supplying the base current of the power transistor 5.
The current value is several tens times the current value of 7 to 330.
This can be easily achieved by increasing the element area of the transistor 331 with respect to the element area of the transistor 350, or by connecting several elements identical to the transistor 350 in parallel.

Next, the output circuit will be described. The output circuit includes operational amplifiers 400 and 500, resistors 21 to 25, and a capacitor 20. This output circuit functions to input the voltage VM from the bridge circuit as a voltage signal having a correlation with the flow rate to the non-inverting input terminal of the operational amplifier 500, and to output the output voltage Vout represented by the following equation (18). Note that the resistance values of the resistors 21 to 25 are R21 to R25.
And the voltage VR is represented by the following equation (19).

[0065]

(Equation 18)

[0066]

[Equation 19]

As shown in the above equation (18), the output circuit has a voltage amplifying function. In addition, by adjusting VR and R22 / R23 in Expression 18, it is possible to match the difference in the VM characteristic value unique to each product to the same characteristic value. This makes it possible to output an output voltage that matches the A / D convertible range of the A / D converter for A / D converting the output voltage Vout, and wastes the A / D convertible range of the A / D converter. Bit error during A / D conversion can be reduced. In addition, the S / N between the signal voltage of the output voltage Vout and the noise voltage superimposed from the outside can be increased, and the noise can be increased.

By supplying the base current of the power transistor 5 from the constant current source 331 as described above, the lowest voltage value of the battery voltage VB at which the flow rate can be measured is the voltage represented by the above equation (16). By the way, in the conventional device, the base current of the power transistor is usually given from the output transistor of the operational amplifier.
A Darlington-connected transistor is generally used as an output transistor of a conventional operational amplifier. For this reason, the minimum voltage at which the flow rate can be measured needs to include the voltage drop of the output transistor of the operational amplifier. Equation 20 below shows the difference between the lowest voltage according to the present embodiment and the lowest voltage of the conventional device. In this equation 20, the voltage drop between the base and the emitter of all transistors is regarded as equal, and is set as VBE.

[0069]

(Equation 20)

However, V1, V2, VCE (MIN)
Is assumed to be the same in the conventional apparatus and the apparatus of the present embodiment. In Equation 20, the first term on the left side is equal to Equation 16 representing the lowest voltage of the present embodiment, and the second term on the left side represents the lowest voltage of the prior art for supplying the base current from the Darlington-connected transistor to the power transistor. Then, the voltage drop of the Darlington-connected transistor is represented by VBE3, VBE4
And Therefore, the apparatus of this embodiment is 2 times smaller than the conventional apparatus.
VBE (approximately 1.4V) can measure flow rates down to low battery voltage.

FIG. 4 is a circuit diagram of the second embodiment. FIG. 4 shows a circuit configuration of the constant current source. In the above-described first embodiment, the constant current source circuit of FIG. 3 is used, but the constant current source circuit of FIG. 4 may be used. In the circuit shown in FIG. 3, the basic current I1 changes depending on the voltage VB1 of the power supply, whereas the basic current i2 in FIG. 4 is a circuit which is not affected by the voltage of the power supply VB1. In the circuit of FIG. 4, the resistor 353, the diode 347,
The activation is constituted by the transistor 350. Then, the transistors 342, 343, 349, 351, 3
52, a diode 346, and resistors 345 and 348 compensate for changes in the power supply voltage VB1 to generate a basic current i2. The basic current i2 produced by the circuit of FIG. 4 does not depend on the voltage of VB1, and the resistances 348, 3
By selecting a resistance value of 45, it is possible to obtain a basic current which hardly depends on temperature.

FIG. 5 is a circuit diagram showing blocks of the circuits of the embodiment shown in FIGS. 1 and 2. In FIG. 5, the same components as those in FIGS. 1 and 2 are denoted by the same reference numerals. The constant voltage circuit 120 in FIG. 5 is a circuit including the elements denoted by reference numerals 105 to 113 in FIGS. The operational amplifier 23 shown in FIG.
Reference numeral 0 denotes a circuit including elements 201 to 219 in FIGS. 1 and 2 and includes a differential amplifier circuit and its output circuit. The operational amplifier 360 in FIG. 5 is a circuit including the elements 301 to 321 in FIGS. 1 and 2 and includes a differential amplifier circuit, its output circuit, and a resistor for setting an offset voltage.

Next, the operation and effect of the embodiment will be described again by showing a comparative example. FIG. 6 is a circuit diagram showing a circuit of a comparative example. In this comparative example, power is supplied to the temperature-sensitive resistor that generates heat from the power supply, and power to the temperature-compensating temperature-sensitive resistor that does not generate heat is supplied from the power source via the current limiting resistor and the overvoltage protection diode. Supplying. In FIG. 6, a hot-wire flowmeter that measures the amount of intake air of an internal combustion engine of a vehicle operates using a battery 400 of the vehicle as a power supply. Power is supplied from a power supply to a series circuit of the heat-sensitive temperature-sensitive resistor 402 and the resistor 404 through a power transistor 406.
The voltage between both ends of the temperature-sensitive resistor 402 is the resistor 408 and the resistor 4
The voltage is divided by a series circuit with the reference numeral 10. Voltage V of resistor 404
M is input to the inverting input terminal of the operational amplifier 414 via the resistor 412. A constant voltage VR is applied to the inverting input terminal of the operational amplifier 414 via the resistor 416, and a voltage drop in the resistor 412 is input as an offset voltage of the operational amplifier 414. A resistor 418 is connected to the non-inverting input terminal of the operational amplifier 414, and the operational amplifier 41
A series circuit of a temperature-sensitive resistor 420 and a resistor 422 that does not generate heat is connected between the output terminal of No. 4 and the non-inverting input terminal. The voltage VA obtained by dividing the voltage between both ends of the temperature-sensitive resistor 402 by the resistor 408 and the resistor 410 is input to the inverting input terminal of the operational amplifier 424. The output of the operational amplifier 414 is input to a non-inverting input terminal of the operational amplifier 424. Then, the output of the operational amplifier 424 is input to the power transistor 406. The power supply is provided with a current limiting resistor 426 in series, and further provided with zener diodes 428, 430, and 432 for overvoltage protection in series. These elements constitute a power supply circuit 434, and power is supplied to the operational amplifiers 414 and 424.

According to the circuit of FIG. 6, the amount of current supplied to the temperature-sensitive resistor 402 is feedback-controlled so that the temperature of the temperature-sensitive resistor 402 becomes higher than the temperature of the temperature-sensitive resistor 420 by a predetermined value. It is output as a signal corresponding to the flow rate. According to the circuit of FIG. 6 described above, since all the current to the temperature sensitive resistor 420 is supplied from the operational amplifier 414, it is necessary to increase the current flowing from the power supply circuit 434 to the operational amplifier 414. However, as this current value increases, the voltage drop across resistor 426 increases,
The problem arises that the minimum supply voltage required for the operation of the circuit powered via 26 is high. Also, in order to increase the current value and reduce the voltage drop, it is necessary to set the resistance value of the resistor 426 small. However, if the resistance value of the resistor 426 is set small, the currents of the Zener diodes 428, 430 and 432 become An element having a large capacity must be used. Therefore, the power supply circuit 434
There is a problem that the size of the device is increased.

Further, in the circuit of FIG.
4 and 424, the effect of reducing the effect of the internal offset voltage increases the current to the temperature-sensitive resistor 420, and further increases the current flowing through the power supply circuit 434. Hereinafter, a configuration for reducing the influence of the internal offset voltage will be described. In the circuit of FIG. 6, the output voltage VM is represented by the following Expression 21. The resistance value of the temperature-sensitive resistor 402 is RH, and the temperature-sensitive resistor 42 is
The resistance value of 0 is represented by RK, and the resistance value of each resistor is represented by using the last two digits of the sign of each resistor. For example, the resistance value of the resistor 404 is represented by R04. VOS1 is the operational amplifier 4
14, the internal offset voltage VOS2 is the operational amplifier 42
4 is the internal offset voltage. Note that RH << R08
+ R10.

[0076]

(Equation 21)

In the above equation, the third term on the right side and the fourth term on the right side
The term represents the effect of the internal offset voltage of the operational amplifier. When the ratio R08 / 10 between the resistance value R08 and the resistance value R10 is changed as can be seen from Expression 21, the values of the third term on the right side and the fourth term on the right side also change. That is, when R08 / R10 is reduced, the value of the third term on the right side and the value of the fourth term on the right side also decrease, and the influence of the internal offset voltage of the operational amplifier can be reduced.

On the other hand, the voltage VB shown in FIG. 6 is represented by the following expression 22, and the current i1 is represented by the following expression 23. As can be seen from Expression 23, the current i1 increases as R08 / R10 decreases.

[0079]

(Equation 22)

[0080]

(Equation 23)

Therefore, in order to obtain a high-precision operation as a hot wire type flow meter, the influence of the internal offset of the operational amplifier should be small. However, if R08 / R10 is reduced to reduce the effect of the internal offset, the current i1 flowing through the temperature-sensitive resistor 420 increases, and the operational amplifier 41
A new problem arises in that the current supplied to 4 increases. Note that the second term on the right side related to ΔV in Expression 21 represents the feedback amount of the bridge circuit.
When 8 / R10 is changed, the value of the second term on the right side also changes, the feedback amount of the bridge circuit changes, and the responsiveness changes. Therefore, ΔV is set by appropriately changing the resistance values of the resistors 412 and 416 so that the bridge circuit operates stably and an appropriate responsiveness is obtained. That is, in order to obtain a predetermined feedback amount, if R08 / R10 is large, ΔV
, And if R08 / R10 is small, ΔV is increased.

As described above, when the current supplied to the operational amplifier 414 increases to reduce the effect of the internal offset voltage of the operational amplifier, the voltage drop in the resistor 426 increases, and the operational amplifier 424 causes the power transistor 406 to conduct. A problem arises in that the minimum power supply voltage required for measuring the flow rate increases, and the flow rate measurement cannot be performed when the power supply voltage is low. Also, the resistor 42
If the resistance value is reduced in order to reduce the voltage drop of 6, the element having a large current capacity must be used as the Zener diodes 428, 430, and 432. Therefore, there is a problem that the power supply circuit 434 is increased in size. In particular, when a circuit such as an operational amplifier is formed of a single-chip monolithic IC in order to form a high-precision, high-quality circuit at a low cost, a Zener diode having a large current capacity has a large element area, so the chip size becomes large.
The cost is high.

Further, in the circuit of FIG. 6, since the internal offset voltages VOS1 and VOS2 of the two operational amplifiers both appear as positive voltages in VM, the output voltage VM changes under the influence of both the internal offset voltages VOS1 and VOS2. I do. For this reason, the output voltage VM fluctuates under the influence of the temperature drift of both the internal offset voltages VOS1 and VOS2. Therefore, the total offset voltage of the entire control circuit for feedback-controlling the temperature of the temperature-sensitive resistor changes under the influence of both the internal offset voltages VOS1 and VOS2,
This affects the responsiveness of the feedback control circuit.

However, according to the first and second embodiments, the current flowing through the temperature-sensitive resistor 8 is supplied from the emitter of the power transistor 5. For this reason, the ratio R10 / R11 of the resistance values of the resistor 10 and the resistor 11 is determined by the increase in the voltage drop in the resistor 3 and the Zener diodes 101 and 10.
It can be set small so as to reduce the influence of the internal offset voltage of the operational amplifiers 200 and 300 without considering the increase in the capacitance of 2 and 103. Further, according to the first and second embodiments, since the current flowing through the temperature-sensitive resistor 8 is supplied from the emitter of the power transistor 5, the resistance value of the resistor 3 can be set larger than that of the circuit of FIG. . Therefore, the capacity of the Zener diode can be reduced without lowering the minimum operating power supply voltage of the device.
In particular, when the circuit is formed into an IC, the element area of the Zener diode can be reduced, the chip size can be reduced, and a low-cost hot-wire flowmeter can be provided.

According to the first and second embodiments,
Since the internal offset voltages VOS1 and VOS2 of the two operational amplifiers 200 and 300 appear as voltages of opposite polarities in the output voltage, they cancel each other. Therefore, the internal offset voltage VOS to the output voltage
1. The effect of VOS2 is reduced. In particular, in the above embodiment, since the two operational amplifiers 200 and 300 are both configured on a one-chip IC, their internal offset voltages and their temperature drifts are substantially equal. Therefore, as a result of these internal offset voltages being canceled out, the total offset voltage can be made substantially equal to the reference voltage and the offset voltage ΔV by the resistors 305 and 306.

In the above embodiment, the offset voltage ΔV is applied to one of the pair of transistors 308 and 309 provided inside the operational amplifier 300. It may be applied to the terminal or the non-inverting input terminal. Further, the offset voltage may be applied from inside or outside the operational amplifier 200.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a part of a circuit according to a first embodiment to which the present invention is applied.

FIG. 2 is a circuit diagram showing a part of a circuit according to a first embodiment to which the present invention is applied.

FIG. 3 is a circuit diagram showing a constant current circuit.

FIG. 4 is a circuit diagram showing a constant current circuit according to a second embodiment.

FIG. 5 is a circuit diagram showing a circuit of a first embodiment to which the present invention is applied.

FIG. 6 is a circuit diagram showing a circuit of a comparative example to which the present invention is not applied.

[Explanation of symbols]

 REFERENCE SIGNS LIST 1 battery 5 power transistor 7 first temperature-sensitive resistor 8 second temperature-sensitive resistor 100 range of monolithic IC 200 operational amplifier (op-amp) 300 op-amp 400 op-amp 500 op-amp 230 op-amp 360 op-amp

Continuation of the front page (56) References JP-A-2-35315 (JP, A) JP-A-64-88218 (JP, A) JP-A-2-77619 (JP, A) JP-A-2-124429 (JP) , A) JP-A-55-43447 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) G01F 1/68

Claims (2)

(57) [Claims] A first temperature-sensitive resistor provided in a fluid flow path; a first power adjusting circuit provided in series with the first temperature-sensitive resistor for adjusting power supplied from a power source; A second temperature-sensitive resistor installed in the flow path, a second power adjustment circuit provided in series with the second temperature-sensitive resistor, and adjusting power supplied to the second temperature-sensitive resistor; An operational amplifier to which a detection signal of the first temperature-sensitive resistor and the detection signal of the second temperature-sensitive resistor are input is included, and a temperature difference between the first temperature-sensitive resistor and the second temperature-sensitive resistor is maintained at a predetermined value. A first control circuit that controls the first power adjustment circuit so as to adjust the power supplied to the first temperature-sensitive resistor; and that the power supplied to the second temperature-sensitive resistor is the first temperature-sensitive resistor. Controlling the second power adjustment circuit so as to reduce the power supplied to the resistor by a predetermined ratio; A second control circuit that adjusts the voltage applied to both ends of the second temperature-sensitive resistor by the voltage (V) applied to both ends of the first temperature-sensitive resistor.
(BR-VM) is set to about 1/10 to 1/20. 2. The voltage dividing circuit (10, 11) for detecting a voltage (VBR-VM) applied to both ends of the first temperature sensing resistor, and a detection signal of the voltage dividing circuit is provided by the second control circuit. And an operational amplifier to be input, wherein the voltage dividing circuit changes the voltage applied to both ends of the second temperature-sensitive resistor from 1/10 of the voltage (VBR-VM) applied to both ends of the first temperature-sensitive resistor. 2. The thermal flow meter according to claim 1, wherein the thermal flow meter is set to about 1/20.