DE2829538A1 - CHANNEL FILTER ARRANGEMENT AND EQUIPPED CHANNEL SELECTOR FOR A TELEVISION RECEIVER - Google Patents

Description

Patent attorneys

DIpL-lna. DIpl.-Chem. DfpL-kig.

E. Prince - Dr. G. Hauser - G. Leiser

Ernsbergerstrasse 19

8 Munich 60

Our sign; T 3110 July 3, 1978

TEXAS INSTRUMENTS INCORPORATED 13500 North Central Expressway Dallas, Texas 75222, V.St.A.

Channel filter arrangement and channel selector equipped therewith for a television receiver

The invention relates generally to an electronic arrangement for receiving a plurality of emitted electromagnetic signals, for filtering a selectable frequency channel from the received signals and for demodulating the signals of the selected filtered channel. In particular, the invention relates on a television receiver.

Known television receivers contain a radio frequency section (RF section) and an intermediate frequency section (IF section). The HF section Contains RF filters that are tuned to allow coarse filtering of a band of channels, the center of which lies at a manually selected channel.

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The output signal of the RF filter is fed to the input of a RF amplifier supplied. Typically the total gain is in the HF section at least 20 to 30 dB. This amplification increases the amplitude of the signals within the selected channel, and it renders the system noise figure of subsequent elements in the Receiver essentially independent. The output signal of the RF amplifier is fed to an input of a mixer, which receives mixed signals with a selectable frequency at a second input. The selectable Frequency is generated so that the frequency of the selected Channel is shifted to about 45 MHz. The output of the mixer is fed to a channel selection filter which has a relatively high impedance for frequencies outside the selected channel and a relatively high impedance for signals inside of the selected channel has a relatively low impedance. The signals at the output of the channel selection filter exist therefore primarily from frequencies within the selected channel.

Each television channel contains sound information, picture information and image synchronization information. The output of the channel selection filter is sent to an audio frequency demodulator which separates the audio information in the selected channel; the output signal of the audio frequency demodulator is fed to a loudspeaker which produces audible tones. The output signal of the channel selection filter becomes also fed to a video processing device, which receives the image and image synchronization information from the detaches selected channel; the output signal of the video processing unit is fed to a picture tube, converts the image and image synchronization information into images.

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As obeli has been described, a basic function of all television receivers is the frequency of the selected Channel to move with the help of a mixer from an RF frequency to a predetermined IF frequency. This mixing process has hitherto been carried out with the aid of various non-linear components. These components can be vacuum tubes, diodes, MOS field effect transistors or be bipolar transistors. However, the mixers used up to now also generate output signal components', proportional to the third power or higher odd powers of their input signals are. In a television receiver, these signal components of odd-numbered powers can interfere with signals create the desired channel. For example, such interference signals are generated when signals are in channels are present at a distance of one and two channels from the selected channel on one side of this selected channel. This phenomenon is known as intermodulation distortion. The components the third power or higher odd powers generate interference signals in the desired channel even if in one of the undesired channels, a carrier with amplitude modulation is present. This phenomenon is called cross modulation distortion known. The frequencies that occur in a desired channel as a result of inter! Modulation distortion or from cross modulation distortion, can be caused by the information signals in this channel not be separated. When the interfering frequencies increase, noticeable picture and sound distortion occurs on.

A major advantage of the television receiver described here is that its cross-modulation and intermodulation

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distortions are greatly reduced. This becomes the for the most part achieved by means of a special HF mixer. This mixer contains an MES field effect transistor (Metal-semiconductor field-effect transistor), which has an almost completely square current-voltage characteristic. Since the MESFET mixer has an almost completely square characteristic, it only produces extremely low distortions into the system. In particular, the MESFET mixer processes an interference signal level greater than +6 dBm at its output with less than 1% cross modulation distortion products and -40 dB intermodulation distortion products. *

The quadratic current-voltage characteristic of an MES field effect transistor and their use to reduce intermodulation and cross-modulation distortions in a television receiver is not apparent from the prior art. So far, MES field effect components have been used primarily used to achieve high working speeds. For example, they were used in digital switchgear assemblies used with switching times in the picosecond range. For example, refer to the publication referenced by Cahen and others entitled "A Subnanosecond Switching Circuit", published on IEEE International Solid State Circuits Conference on Feb. 14 1974 in Orsay, France. Also see the article "X and KU Band Amplifiers with GaAs Schottky Barrier Field Effect Transistors by Weiner Baechtold in the IEEE Journal of Solid State Circuits, Volume Sc 8, No. Referenced February 1973. The high-speed digital switching is one of high-speed working pulse amplitude modulation arrangement with a

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rapid linear amplification, which is described in the publication "Performance of Dual Gate GaAs MESFETS as Gain Controlled Low Noise Amplifiers and High Speed Modulators "by Liechti in IEEE Transactions on Microwave Theory and Techniques, Volume MIT-23, No. 6, June 1975. Use all of these circuit arrangements in the cited publications the MES field effect transistors to achieve high working speeds.

The television receiver to be described here also contains a special combination of the technology of Surface acoustic wave components with the technology of MESr field effect components in the HF-IF section. A MESFET mixer gives a frequency shift, while the channel selectivity by means of a single low-loss surface wave band filter for surface acoustic waves is achieved. This filter has a steep transition from the pass band to the stop band. Examples are signals that are around 1.5 MHz above the sound carrier located in the passband are attenuated by more than 65 dB. On the other hand, the filter points has a low insertion loss in the passband. The preferred embodiment is one in one Directional effective filter in which the insertion loss within the passband is less than Is 3.5 dB.

The surface acoustic wave filter (hereinafter referred to as SWD filter also has a high center frequency and a pass band that occupies a small percentage of the center

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frequency is. The SWD filter can therefore easily be implemented with piezoelectric material which has a relatively low coupling coefficient. Quartz has this property that it is used as a preferred substrate material. In comparison, SWD filters were only used in known television receivers at a much lower intermediate frequency of 45 MHz. Its pass band is therefore a relatively large percentage of its center frequency, so that the filter component required a piezoelectric material with relatively large coupling coefficients such as lithium niobate. Lithium niobate, however, has piezoelectric properties which show great sensitivity to temperature changes in the range from 0 ° to 70 °. For this reason, temperature compensation circuits are required. In comparison, the piezoelectric properties of quartz relatively unempfind Lich for temperature changes in the range of 0 ° to 70 ⁰ C; Temperature compensation circuits are therefore not required.

Known television receivers contain tuned RF filters upstream of the mixer so that it is ensured that the image frequency sequences of the selected signal at the mixer input are sufficient attenuated so that no interfering mixer output signals are generated. To achieve this result The bandwidth of the matched RF filters is only a few channels, and the center frequency is like that set to be in line with the selected channel. RF filter covering all channels of the VHF band and pass the UHF band at the same time cannot be used, as image frequencies prevent reception in the

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would destroy the selected channel.

In order to confuse the above, consider the following: VHF television channels are in the ranges of 55.25 MHz to 71.75 MHz and from 177.25 MHz to 215.75 MHz, while UHF television channels are between 471.25 MHz and 889.75 MHz. It is also known that the image frequencies of a selected channel are twice the value of the intermediate frequency above the frequency of the selected channel. Channel 6, located at 87.75 MHz, has one in a conventional television receiver, for example. Image frequency, which is at (87.75 + 90) MHz. This results in the frequency 177.75 MHz, which lies within channel 7. The image carrier of channel 14 is at 471 MHz; its image frequency therefore has that lying within the channel 29 'Value of 561 MHz. Television receivers that have been customary up to now therefore need to filter out the image frequencies upstream of the mixer an RF tuner.

So far, the HF tuning has been carried out with the help of capacitance period filters, mechanically variable capacitors and the like performed. However, these components are expensive and difficult to match. In comparison, the television receiver to be described here does not contain any Tuner in the HF section. A preferred embodiment contains a fixed band filter that passes the entire lower VHF band, a second filter that passes the entire upper VHF band passes, as well as a third filter that passes the entire UHF band. These solid filters can can be easily set up, and alignment problems no longer occur. The invention also includes a mixer, its intermediate output frequency in the range between 300 MHz

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and 400 MHz. As a result, the image frequencies of the selected channels are at least 600 MHz above the selected channel where they are in a simple manner can be suppressed by the fixed HP band filters. The output of the mixer to be described here Receiver is fed to a channel filter, which is formed by a surface wave component. In so far well-known television receivers were also already channel selection filters applied from surface acoustic wave components. The SWD filter to be described here is, however, an improved one Execution because it is formed on a relatively small substrate area. The one used by an SWD filter Area is proportional to its center frequency. With conventional receivers an intermediate frequency of 45 MHz is used used so that the SWD filters used require considerably more space.

Another feature of the arrangement described here is that the IF section has two mixers on Contains output of the SWD filter, which shift the frequency of the selected channel back into the baseband. The first IF mixer effects a frequency shift of the selected channel to about 45 MHz. The biggest Part of the amplification is then carried out on the signals of the selected channel. The second IF mixer is used to demodulate the signals synchronously in the selected channel. This structure allows it to be achieved the great advantages of the IF system without RF feedback problems appear.

As already mentioned, in known television receivers there is an amplification of at least in the HF section

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IN THE

up to 30 dB, so that a low system noise figure is achieved will. The system noise figure results from

NF ₉ -I NF ₉ -I NF ₉ -I NF ₁ + - = «+« - = * + α α μ

where NF ₁ and G ^ are the noise figure and the gain for the ith function block of the system, respectively. The introduction of a large gain in the first functional blocks (ie in the RF section) lowers the system noise figure by making it independent of the noise figures of the subsequent circuit.

The arrangement according to the invention has a special structure which at the same time results in a low noise figure, small intermodulation distortions and small cross modulation distortions. In the preferred embodiment, the arrangement according to the invention in the channel selection section has a maximum gain which is less than 10 dB; circuit elements are used which individually have such a noise figure that a low system noise figure results, the low gain allowing the RF and IF circuit elements to operate within their dynamic range without generating higher odd order signal components. * ■

A special feature of the arrangement according to the invention is that it has two IF frequencies around which the System gain is selectively distributed. In a preferred embodiment, a first mixer moves the selected channel to a frequency of about 330 MHz.

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The output signal of this mixer is fed to a channel selection filter; the output of the channel selection filter leads to-a .second mixer that adjusts the frequency of the filtered selected channel to a second ZP frequency of about 45 MHz. Most of the reinforcement of the Systems (about 60 dB) is performed at the lower IF frequency after the channel selection filter. The reinforcement at the high intermediate frequency is low, as stated above. As a result, there are several advantages of a system obtained with a high intermediate frequency, for example a simplified filtering of image frequencies while a reaction which is a property of a high-gain IF section "is avoided.

The arrangement to be described is also simpler and less expensive as a known television receiver, since the low-gain RF section is an integration of the main section of the receiver on a single semiconductor chip. Two differently structured HF-IF sections are described, which are suitable for integration on a semiconductor chip.

In accordance with one feature of the invention, a television receiver includes a MESFET mixer along with a surface acoustic wave filter for surface acoustic waves. The MESFET mixer receives several channels at the same time and mixed signals with selectable frequency, and it shifts the frequency of selected ones in dependence thereon Channels at a predetermined center frequency. "The MESFET mixer has an almost completely square characteristic, so that it is only extremely small in the system

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2k

Distortion leads. In one embodiment, the MESFET mixer handles noise levels greater than +6 dBm with less than 3/4% cross-modulation distortion products and -40 dB intermodulation distortion products. The surface acoustic wave filter has an input connected to the output of the MESFET mixer. The SWD filter essentially provides the entire channel selectivity of the receiver. In a preferred embodiment, the SWD filter is a low-loss, unidirectional filter that works with three clock phases and whose impulse response is determined by removing finger-like electrodes. This SWD filter has an insertion loss of ^ 3.5 dB in the pass band, and its blocking loss at 1.5 MHz above and below the pass band is> 65 dB.

In accordance with another feature of the invention, an untuned RF channel selector created. The channel selector consists of an antenna connection, unmatched filters and a high frequency mixer. The antenna connection receives radiated electromagnetic signals, for example contain multiple television channels. The mismatched filters have inputs connected to the antenna connector and they filter out fixed spectra from television channels from the received signals. A preferred embodiment contains a fixed filter that allows the entire lower VHF spectrum to pass, another fixed filter, which lets through the entire upper VHF spectrum, as well as another fixed filter that lets through the entire UHF spectrum. The mixer simultaneously receives one of the fixed channel spectra and mixed signals with a high input at the inputs selectable frequency. The high selectable frequency shifts selectable channels of the spectrum to a high one

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Intermediate frequency that is much higher than 45 MHz. The high intermediate frequency is in the preferred embodiment between 300 and 400 MHz. The preferred embodiment also includes a surface acoustic wave filter for surface acoustic waves. This SYJDi filter is connected to the mixer output connected and filters out the selectable channel at the high intermediate frequency. The substrate of the SWD filter has a low coupling coefficient; in the preferred embodiment it consists of quartz.

According to a further feature of the invention, a low gain RF channel selector created, which has an antenna connector, a spectrum filter, a mixer and a channel selection filter contains. The antenna connection receives radiated electromagnetic signals. The spectrum filter is with • connected to the antenna connection and filters a signal spectrum, for example television channels, from several. Inputs of the mixer are connected to the spectrum filter, and it there is a frequency shift of selected channels of the spectrum to a predetermined intermediate frequency. In a Embodiment, a low-gain MESFET mixer is used. The channel selection filter has one with the Mixer connected input, and it filters out the channel at the predetermined intermediate frequency. The overall gain from the antenna connection via the channel selection filter is not greater than the gain required to "achieve a desired system noise figure is required. In one embodiment, the overall gain is less than 10 dB. An amplifier device is connected to the output of the channel selection filter. The amplifier device causes a gain that is significantly higher than the overall gain of the system before the channel selection filter. A preferred one Embodiment includes a second mixer that is connected to the

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Output of the amplifier is connected and the frequency of the selected channel to a lower intermediate frequency moves where additional reinforcement is needed.

The invention will now be explained by way of example with reference to the drawing. Show it:

1 shows a block diagram of a channel selector according to the Invention,

2 several frequency diagrams to illustrate Signals that occur at selected points of the channel selector from Fig. 1,

3 shows a time diagram to explain the operation of a phase-locked loop in the channel selector from FIG.

Fig. 4 several diagrams to illustrate the amplitude of signals lying inside and outside the frequency band at different points of the channel selector of Fig. 1,

Fig. 5 several precise circuit diagrams of HF filters in the channel selector of Fig. 1,

6 shows a detailed circuit diagram of a 2-to-1 switch in Channel selector from Fig. 1,

7 shows a detailed circuit diagram of an RF amplifier in the channel selector of Fig. 1,

Fig. 8 shows a detailed circuit diagram of a MESFET filter in the channel selector of Fig. 1,

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9a to 9e representations to illustrate the structure and operating characteristics of an MES »-Eeldeffßkttransistor for use in the MESFET mixer of Fig. 8,

Fig.10 is a circuit diagram of a second 2-to-1 switch in Channel selector from Fig. 1,

11a to 11h diagrams to illustrate structural and details of operations of a surface acoustic wave filter in the channel selector of Fig. 1,

12a and 12b are detailed circuit diagrams of a linear amplifier in the channel selector of Fig. 1,

Fig. 13 is a detailed circuit diagram of a mixer in the channel selector of Fig. 1,

Flg.14a and 14b are detailed circuit diagrams of a surface acoustic wave component formed oscillator in the channel selector of Fig. 1,

Fig.15 is a detailed circuit diagram of a high gain linear Amplifier in the channel selector of Fig. 1,

Fig. 16 is a detailed circuit diagram of a synchronous detector in Channel selector from Fig. 1,

Fig. 17 shows a detailed circuit diagram of an LC oscillator in the channel selector of Fig. 1,

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Fig.18 is a detailed circuit diagram of a phase detector, a Sögezahngenerator and a loop filter in the channel selector of Fig. 1,

Fig. 19 is a detailed circuit diagram of a voltage controlled Oscillator in the channel selector of Fig. 1,

Fig.20 is a block diagram of a television receiver which contains the channel selector of Figure 1, and

Fig. 21 is a block diagram of another embodiment of the channel selector from FAg.1.

In Figure 1, the channel selection section of a television receiver according to the invention is shown in the form of a block diagram. The channel selector contains a VHF antenna 10, the output of which is connected via a line 11 to a fixed band filter and a further fixed band filter 13. The signals appearing on line 11 are then referred to as S ₁ Cf). The band filter 12 has a center frequency of 69 MHz and a 3 dB bandwidth of 34.6 MHz so that the lower VHF television channels can be passed. The center frequency of the band filter 13 is 193 MHz and its 3 dB bandwidth is 44.5 MHz in order to filter out the upper VHF television channels. The output of the band filter 12 is connected to one of the inputs of a 2 × 1 switch 15 via a line 14, while the output of the band filter 13 is connected to a second input of the switch 15 via a line 16. The switch 15 effects the selection of the output signal of the band filter 12

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or the output signal of the band filter 13. The combination from the band filters and the switch results in the passband a signal attenuation of about 1 dB.

The output of the switch 15 is connected to an RF amplifier 17 via a line 18. The amplifier 17 has a variable gain factor which is controlled by the signal on an AGC line 19. The maximum gain of amplifier 17 is about 4 dB. The amplifier's noise figure is around 3 dB.

The output of the amplifier 17 "is via a line 21 with one input of a MESFET mixer 20 (mixer with metal-semiconductor field effect transistor) tied together. The mixer has a fixed gain of around 4 dB and a noise figure Λτοη about 8 dB. A second input of the mixer 20 is via a line 23 to a voltage controlled VHF oscillator 22 connected. The oscillator 22 generates local oscillator signals (LO signals) on line 23 with a selectable frequency in the range from 385 to 541 MHz. Depending on these signals shifts the mixer 20 changes the frequency of the RF signals on the line to a new intermediate frequency range. The frequency of the LO signals on line 23 is selected so that the frequency of the channel to be received is in a predetermined high Intermediate frequency range is shifted between 300 MHz to 400 MHz. In one embodiment, this is high intermediate frequency about 330 MHz. The frequency shifted signals are generated on line 24; she are denoted by S2 i-f).

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Line 24 has one input of a 2x1 switch connected, whose second input receives via a line 26 frequency-shifted UHF television channels. The desk 25 has the same structure as switch 15. The circuit arrangement for shifting the frequency of the UHF television channels, which is designated with UHF-HF section is structured in the same way as the VHF-HF section; she is explained in more detail below.

The output of the switch 25 is via a line 27 with one formed by a surface acoustic wave component Filter 28 (SWD filter) connected. The pass band of filter 28 is shaped to only include one of the television channels on the line 27 lets through, in particular the filter 28 lets the television channel with the predetermined high intermediate frequency. The fixed RF filters, together with the SWD filter, essentially form the whole Filter arrangement of the system. In a preferred embodiment, filter 28 is a three-phase clock unidirectional filter that has low insertion loss. Typically the attenuation of the filter 28 in the passband is below 3 »5 dB. Those lying outside the passband Signals, on the other hand, are strongly attenuated.

The output of the filter 28 is via a line 30 with an IF amplifier 29 is connected. The signals on line 30 are labeled S, (f). The amplifier 29 has a variable gain which is controlled by AGC signals on a line 31. The maximal The gain of the amplifier 39 is about 30 dB. This

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Amplifier 29 is therefore the first high-gain component in the system. Up to this point the reinforcement of the System characterized in that it is not greater than the gain necessary to achieve the desired System noise figure is necessary. The noise figure of the amplifier 29 is about 4 dB.

The output of the amplifier 29 is connected to a mixer 32 via a line 33. A second entrance to the Mixer 32 is connected to the output of an oscillator 35 via a line 34. The oscillator 35 contains as Frequency control a γοη a surface acoustic wave component formed resonator 36. In the preferred embodiment, the resonator 36 and the oscillator operate 35 in such a way that on the line 34 a mixed signal with the frequency 285MHz is generated. The 285 MHz signal on line 34 is matched with the signal on line 33 mixed so that 32 signals S ^ (f) are generated at the output of the mixer. The signal S ^ (f) is similar to the signal S, (f) except that its frequency is shifted down by 285 MHz. It therefore contains an image carrier of about 45 MHz.

An additional amplification of the signal of the selected channel occurs after its shift to the lower one Intermediate frequency of 45 MHz. The mixer 32 gives a fixed gain of +10 dB. The output of the mixer is connected to an IF amplifier 38 via a line 37, whose maximum gain is +50 dB. The gain of the amplifier 38 is made with the aid of an AGC signal changed on line 39. By having most of the gain at the lower intermediate frequency,

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the system stability is improved, since the effect of interference capacitances, radiation and the like is significantly less at 1 + 5 MTf than at 330 MHz.

The output of the amplifier 38 is connected via a line to an oscillating circuit 41, to a synchronous demodulator and to a phase detector 43. The center frequency of the resonant circuit 41 is about 45 MHz. The synchronous demodulator 42 has a second input which is connected to an oscillator 45 via a line 44. The oscillator 45 produces on a line 44 clock signals having a fixed frequency of 45 MHz ₄ The signals on line 44 are connected to the 45 MHz picture carrier at the line 40 in phase. The synchronous demodulator 42 mixes the signals on the lines 40 and 44 to generate output _{signals S c} (f) on the line 46. The signal S = (f) contains the desired television channel with the video carrier at the frequency 0 Hz and the sound carrier the frequency 4.5 MHz. The line is then connected to a conventional television circuit for separating the audio signal from the video signal and for conventionally generating the audio or video from these signals.

The synchronous clock signal on the line 44 is held with the image carrier on the line 40 by means of the phase detector in phase, working in cooperation with the oscillator 45th The phase detector 43 generates on the line

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Phase fixation signals PD2, the phase difference between the picture carrier on line 40 and the oscillator signal on line 47. Hold signals PD2 a phase difference of 90 ° is maintained between the signals on line 40 and on line 47. These The phase difference is compensated by the oscillator 45, which is between the oscillator signals on the lines 44 and 47 maintains a compensation phase difference of 90 °.

To complete the phase-locked loop, the line 48 is connected to a loop filter 49, its output via a line 51 with a summing element 50 is connected. The summing element 50 has a second input, which via a line 53 with a Control member 52 is connected; a third input of the summing element is via a line 55 with a sawtooth oscillator 54 connected. The control member 52 and the oscillator 54 allow the output of a coarse Voltage value to select a channel. Channel selection switches 56 are connected to the control member 52 via lines 57 tied together. The channel selection switches 56 generate digital signals on lines 57 which represent the selected Show channel. The control element 52 contains a digital-to-analog converter, the line 53 is a coarse selection voltage generated for channel selection depending on the digital signals. The signals on lines 53 are with the coarse tuning signals on lines 51 and 55 are summed to form a loop which is phase-locked with respect to the image carrier of the selected channel. The output of the summer is via a line 58 connected to the voltage controlled VHF oscillator 22. The oscillator 22 generates LO signals on the line 23

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bb

at a frequency in the range of 385 to 541 MHz in Depending on the phase detector signals on lines 58, thereby closing the loop.

The UHF-HF section of the channel selector from Fig. 1 begins in the case of the UHF antenna 59. The antenna 59 is connected to a high-pass filter 60 via a line 61. That Filter 60 has a fixed 3 dB cutoff frequency of approximately 380 MHz. The output of filter 60 is over a line 63 connected to the Einarig of an RF amplifier. The amplifier 62 also has an AGC input receiving the AVR signals on line 19. The maximum gain of the amplifier 62 is 4 dB; its noise figure is about 4 dB. With the output of the amplifier 62 is the input of a low-pass filter tied together. The low-pass filter 65 has a fixed 3 dB cutoff frequency of about 936 MHz. Filter 65 and filter 60 together therefore leave the entire UHF band while other frequencies are suppressed.

A mixer 67 receives over a line 66 those in the UHF band lying signals from the. Filter 65. At the same time, the mixer 67 receives mixed signals with selectable Frequency from a voltage controlled UHF oscillator These selectable frequencies are in a range from approximately 801 MHz to 1215 MHz. The at a certain time point just present frequency is generated depending on the channel selection switches 56 so that a frequency shift of the selected channel takes place at the predetermined high intermediate frequency.

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3>

The circuit arrangement described in connection with Fig. 1 works as follows:

The antennas 10 and 59 receive radiated electromagnetic Signals that contain the VHF and UHF frequency spectrum. Those selected by the antenna 10 Signals are fed to the untuned filters 12 and 13 covering the entire lower VHF frequency spectrum or allow the entire upper VHF frequency spectrum to pass. In the same way, the signals from the antenna 59 received signals fed to the mismatched filters 60 and 65, which pass the entire UHF frequency spectrum. The switches 50 and 25 select one of these three from the channel selection switches 56 in dependence on digital signals Frequency spectra.

The MESFET mixers 20 or 67 then cause a frequency shift of the selected frequency band in a predetermined intermediate frequency range from 300 to 400 MHz « The mixer 20 shifts the frequency of the VHF signals, while the mixer 67 shifts the frequency of the UHF signals shifts. The two mixers have a large dynamic range in which their output signal is almost perfect Is the product of their input signals. As a result, it results better behavior of the circuit arrangement. For example, the M4j3Cher 20 and 67 can have noise levels of more than +6 dBm at their output with less than 1 # Handle cross modulation distortion.

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The switch 25 then connects one of the mixers the input of the SWD filter 28, the filter 28 causes a strong attenuation of the signals lying outside the selected channel. The blocking band of the filter 28 has a notch, so that the lower adjoining sound carrier and the upper adjoining picture carrier by more than 65 dB attenuated. All other signals outside the pass band are attenuated by at least 55 dB. The insertion loss of the filter 28 within the passband, on the other hand, is only 3.5 dB.

The output signals of the filter 28 are transmitted to the amplifier 29, the first high-gain The unit of the system is. The maximum gain of the amplifier 29 is 30 dB. Compared to that the total gain of the system before amplifier 29 is less than 10 dB. As a result of the low HF-IF amplification, the intermodulation distortion and the cross-modulation distortion are reduced. Besides that the system stability is improved, since there is an HF reaction due to interference capacitances, from radiation and the like is avoided. At the same time, a low system noise figure is achieved through circuit elements, which individually have a low noise figure in combination with a sufficiently high gain.

The low gain RF IF section also enables an integration of the entire channel selector, with the exception of the fixed filters, on a single semiconductor chip. In FIG. 1, an outline of such a semiconductor chip is indicated by the dashed line 69. That Process for manufacturing the semiconductor chip includes a combination of process steps that are currently available for the production of MESFET components and of bipolar components are known. A second chip

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is used for the construction of the SWD filter 28 and the SWD resonator 36 used. The fixed filters 12, 13, 60 and 65 are turned off constructed from discrete components.

Signals in the selected channel at the output of the amplifier are frequency shifted by mixer 32 to 45 MHz subjected. They are then further amplified by the amplifier 38 to -10 dBm. The video signals of the selected Channel are then returned to the baseband by the synchronous demodulator 42, while the sound carrier of the selected channel is shifted to 4.5 MHz. A 4.5 MHz trap 70 removes the sound from the video signals, and a low pass filter 71 removes all signals except the video signal of the selected channel. The video signals at the output of the filter 71 are transmitted over the line 72 to the video processing circuit, while the audio signals be transmitted at the frequency of 4.5 MHz on line 46 to the sound processing circuit. These Sound and video processing circuits will be explained in connection with FIG.

In the figures 2a to 2e frequency _{diagrams of the signals «S ^ to S 5} (f) are shown. Fig. 2a shows the lower VHF band 75a, the upper VHF band 75b and the UHF band 75c. Each of these bands contains several channels; each channel has a frequency spectrum, as shown in more detail at 76. The frequency allocation and the type of modulation of the signals within each. Channels correspond to a well-known established standard. FIG. 2b shows the example of a frequency diagram for the signal S ₂ (f). In the example shown, the signal S ₂ (f) contains channels in the lower VHF spectrum. The selected channel is close to 330 MHz. The frequency of the local oscillator signal on line 23 minus the frequency of the video carrier from the selected channel

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has the value 330 MHz. Since the mixing frequency is higher than the video carrier, the frequency spectrum is at the output of the Mixer 20 with respect to the input frequency spectrum shown at 77 is reversed.

The frequency spectrum of the signal S * (f) is shown in FIG. The signal S ^ (f) is the output signal of the channel selection filter 28. It therefore basically only contains frequencies within the selected channel. The signal S, (f) is then amplified and its frequency is shifted down 285 MHz. The result is that in Fig.2d signal S shown

The signal S ^ (f) is further amplified and then demodulated by the synchronous demodulator 42. These processes generate the signal S ₅ (f), which is shown in Figure 2e. It should be noted that the mixing action of the synchronous demodulator U2 causes another inversion of the frequency spectrum of the selected signal. Therefore, the selected channel's picture carrier is 0 Hz and the selected channel's audio carrier is 4.5 MHz, as shown at 78.

FIG. 3 shows a timing diagram that shows the mode of operation of the phase-locked loop of the channel selector of Fig.1 shows. As can be seen from the illustration, the signal PD4 on line 58 consists of the Components PD1 to PD3. The signal PD3 forms a coarse selection voltage for the channel selection, which is determined by the control element 52 is produced. The signals PD2 and PD1 provide the fine tuning for the phase locked loop. Yes example of Figure 3 is during a first time interval Δ T1 a certain channel is selected, while another channel is selected in the course of a time interval Δ T2. The signal PD3 gives the coarse selection voltage for the

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Channel selection. Signal PD1 compensates for a DC offset between signal PD3 and the desired one Voltage value. The signal PD2 results in a dynamic correction voltage for the concentration of momentarily occurring Phase or frequency differences between the signals on lines 40 and 44.

The size of the signals of Figure 2 is shown in Figures 4a and 4b. FIG. 4a shows the size of the signals in the desired channel, while FIG. 4b shows the size of the interference signals in a channel which is two channels away. The curves 81 to 84 indicate the signal size in the desired channel at different points of the System on when the input signal strength is 0 dBm, -35 dBm, - 55 dBm or - 85 dBm. As these curves show, there is a first amplification by the amplifier 38, then a further amplification by the amplifier 29 and finally an amplification by the amplifier 17, when the input signal strength decreases in the selected channel. In particular, the RF section has no gain when the input signal strength is selected Channel is not less than -55 dBm. The RF section reaches its maximum gain of +3 dBm when when the input signal strength of the selected channel is between -55 dBm and -85dBm.

As has already been explained above, the low-gain RF section together with the MESFET RF amplifiers and the MESFET mixer results in an arrangement with superior channel resolution. This is shown in Fig. 4b by the relative signal strength of the selected and of the unselected channels. The curve 85 in FIG. 4b indicates, for example, the case

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where the input signal strength of the selected channel is - 55 dBm, while the input signal strength two channels further is + 1 dBm. The curve 86 indicates the case in which the input signal strength in the selected channel is -35 dBM, while the input signal strength two channels further is + 2 dBm. The strictest demands on the receiver occur when the first RF amplifier has to amplify (ie when the signal strength of the desired channel is below -55 dBm). This is because the added gain increases the non-linearities in the RF section and exacerbates the cross-modulation and intermodulation distortion. The curve 85 therefore indicates the most severe conditions for the system. Under the conditions of curve 85, the RF amplifier 17 and the mixer must have cross-modulation and intermodulation distortion below Λ% . This requirement is met because the HESFET components described here have cross-modulation and intermodulation distortion of less than 1 # when their output signal level is below +6 dBa.

After the selected channel has been shifted to 330 MHz, those are relative to the selected channel Signals that are two channels away are strongly attenuated. The mixer 20 has a matched output, which attenuates the two channels further lying signals by -4 dB. The SVD filter 28 attenuates all outside of the Passband signals by at least -53 dB. At the same time, the mixer 20 effects an amplification of the desired signal by 4 dB. The filter 28 only has the effect of attenuating the signals of the selected channel by 3.5 dB.

The details of the various circuit blocks of Figure 1 will now be in connection with Figures 5 to

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described. Fig. 5 shows a circuit diagram of the band filter 12. The band filter 12 basically consists of two LC series resonance circuits 91 and 92 and one LC parallel resonance circuit 93. The filter 13 is constructed in the same way.

A main function of the RF filters 12 and 13 is to pass a band of frequency channels and simultaneously Reject image frequencies to such an extent that perceptible image disturbances are eliminated. Since an intermediate frequency of 330 MHz is used in this system, all image frequencies are around 660 MHz above the frequency of the desired channel. The image frequencies of the signals in the lower VHF range become therefore attenuated by more than 80 dB by the three-pole bandpass filter of FIG. In the system of Fig. 1 are just stepping Image interference is still perceptible when the image frequency signals are less than 36 dB below the Level of the useful signal at the filter output. Thus, for example, the filter of FIG. 5 gives an adequate one Image frequency suppression if the desired image signal level is - 55 dBm and the image frequency level lie at - 11 dBm. The image frequencies for the lower VHF spectrum and the upper VHF spectrum are within the UHF spectrum, and the level of the UHF signal at the input of the VHF antenna can normally with a level below - 11 dBm can be expected.

Figures 5b and 5c show circuit diagrams of a high-pass filter and a low-pass filter for use in the UHF-HF section of Fig. 1. All UHF image frequencies are outside the television band. The first image frequency is around 1130 MHz, and the final image frequency is around 15 ^ 5 MHz. These frequencies are assigned to flight navigation, whereby the navigation

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system TACAN has the highest output power. However, the TACAN system has an output power of only 5 kW with a duty cycle of 1.8 χ 10 to The TACAN signal is therefore 50 dB below the signal, broadcast by a 1 MW television station. The filters of Figures 5b and 5c therefore provide an adequate one Image signal suppression for the UHF band.

Fig. 6 shows a detailed circuit diagram of the damage from One section consists of a diode 101 connecting lines 14 and 18 and an RLC bias voltage element 102 with a control input 103. A DC voltage control signal is applied to line 103 SELLOVHF applied, which in a selected manner turns the diode 101 on or off so that the signals of the lower VHF range present on line 14 selected or not selected. Another section that is just like the one just described Section is constructed, connects the line to the line 16; it serves to open the channels in the upper Select or not select VHF range.

The circuit diagram of the RF amplifier 17 is shown in FIG and FIG. 8 shows the circuit diagram of the mixer 20. The amplifier 17 contains an MES field effect transistor 111 with two gate electrodes, at its source electrode a bias resistor and a coupling capacitor 112 and an LC bandpass filter 113 connected to its drain electrode are. The gain factor of the field effect transistor is determined with the aid of a gain control signal AVRHF changed by the values given in Fig. 4a. The gain control signal AVRHF is connected to a gate electrode of the field effect transistor 111 via a voltage

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divider 114 created. The mixer 20 contains an MES field effect transistor 121 with a gate electrode its source has a bias resistor and a coupling capacitor 122 and its drain an LC ribbon blender circuit 123 are connected. the Band filter circuit 123 has a center frequency of 330 MHz. The mixed signals from the voltage controlled VHF oscillator become the gate electrode of the field effect transistor fed via an RC element 124.

It should be emphasized that the gate structure of the MES field effect transistors, which has a Schottky barrier layer of the amplifier 17 and the mixer 20 have considerable improvements in properties over known ones Circuits result. MOS field effect transistors of the Narrow gate depletion types work at high frequencies, but they only have one very narrow range of the gate preload a characteristic curve approximated to a quadratic curve «As a deviation from an operation on a quadratic curve Characteristic curve leads to cross modulation and grid modulation distortion, such components are on one limited dynamic range. In comparison, the MES field effect transistors 111, 112 have almost ideal quadratic characteristics over the entire working range indicated in Fig. 4a. Junction field effect transistors also have a good square-law characteristic, but their behavior is high Frequencies as a result of interfering capacities and difficulties in production that can be exploited Limit geometric configurations, severely impaired.

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A photograph of a complete MES field effect transistor which is suitable for use as a mixer 20 is shown in FIG. The MES field effect transistor of FIG. 9a has a closed gate electrode. The length of the gate electrode 131 is approximately 2 mm (80 mils). The gate metal is approximately 7.5 yum (0.3 mils) wide while the Schottky barrier gate electrode, which is in contact with the gate metal Contact is approximately 3.75 µm (0.15 mils) wide. The source electrode consists of five fingers 132 to 136 that are 0.82 mm (32.7 mils) long, 102.5 / m (4.1 mils), 102.5 µm (4.1 mils), 192.5yum (7.7 mils) and 302.5yum (12.1 mils), respectively. The width of these fingers is approximately 7.5 Aim (0.3 mils). The drain electrode consists of fingers 13-140.

FIG. 9b shows a greatly enlarged section along the line AA from FIG. 9a. The MES field effect transistor is built on a silicon substrate 140 with a P line. The resistivity of the substrate 140 is about 50 ohm · cm. Each of the source electrodes 132 to 136 is connected to an N ⁺ -doped region 142 to 146. The specific resistance of these doped zones is about 0.005 ohm »approx. The doped zones protrude about 7.5 / µm above their respective electrodes and are spaced about 3.75 µm from the spaced gate electrodes. In the same way, the metal electrodes 137 to 140, which form the dl · drain sleeves, are connected to the N ⁺ -doped zones to 150 underneath. The specific resistance and the geometry of the doped drain ions are the same as those of the doped source zones.

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Pig.9c shows a greatly enlarged section through a Source-drain electrode pair in an MES field effect transistor with two gate electrodes for use in amplifier 17. The entire component is constructed in the same way like the component shown in Fig.9a, but a modification is that between the source and drain electrodes are inserted into two gate electrodes. In Fig.9c the source electrodes correspond 132a and the drain electrode 137a to the corresponding electrodes 132 and 137 of FIG. 9a. The qate electrodes 131a and 131b take the place of electrode 131 of Figure 9a.

FIG. 9d shows the current-voltage characteristics of the component from FIG. 9a, while FIG. 9e the counteractive conductance of this component as a function from the gate voltage shows. In the case of a component with an ideal quadratic characteristic, the is Drain current proportional to the square of the gate voltage. Since the counteractive conductance is equal to the partial derivative of the drain current is after the gate voltage, it is the gate voltage for ideal quadratic operation directly proportional. FIG. 9e shows such a linear relationship between the negative conductance and the gate voltage for the MES field effect transistor of Figure 9a.

Fig. 10 shows the circuit diagram of switch 25. Switch 25 is the same as that previously in connection Switch 15 described with Figure 6 constructed. The signals on lines 24 and 26 are using of VHF dial-up signals (SELVHF) and of UHF-Wfchlsignalen (SELUHF), which are DC voltage control signals, are coupled to output line 27.

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Details of the SWD filter 28 are now illustrated with reference to FIGS. 11a to 11h. The filter 28 consists from a surface acoustic wave component 160 with an LC input circuit 170 and an LC output circuit for impedance matching and phase shifting of the component supplied to and from the component 160 given signals. Lines 172-174 and 192-194 connect circuits 170 and 190 to the Surface acoustic wave component 160, as shown in Figure 11a is.

The surface acoustic wave component 16O has the one shown in FIG. 11b specified frequency response. The component 160 attenuates signals that are 1.5 MHz above the desired image carrier Channel are at least 65 dB. The sound carrier of the desired one is at this frequency Channel adjacent channel. It is important to attenuate this sound carrier heavily, as it is caused by the synchronous demodulator 42 is converted into the video signals of the selected channel. That means -that the sound carrier is at 46.5 MHz is converted by the synchronous demodulator 42 to 1.5 MHz. In the receiver of Fig.1 there are barely noticeable Image interference occurs when the sound carrier at 46.5 MHz is allowed to pass through the SWD filter 28 with an amplitude that is within 36 dB of the 45 MHz video carrier. The SWD filter 28, the adjacent channel suppression of which is greater than 65 dB, enables the receiver of FIG Good image reception, even if the signals at the filter input in the adjacent channel are significantly greater than the signals are in the desired channel. For example, if the signal level of the desired channel is -55 dBm at the input of the filter 28, then the level of the sound carrier

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with the frequency of 46.5 MHz at the input of the filter can be up to -26 dBm, and the receiver of Fig. 1 complies the requirements of -36 dB spacing between those in-band and out-of-band Signals at the output of the filter 28, in comparison, known television receivers have perceptible picture interference when the level of the sound carrier of the adjacent channel is -40 dBm.

DLe FIGS. 11c and 11d show the spatial, geometric structure of an embodiment of the surface wave component 160. The component 16O includes a piezoelectric substrate 161, which in a preferred Embodiment consists of quartz. Quartz has the desired property that the speed of the surface waves transmitted by the quartz are practically independent of temperature. The dependence of the speed the temperature is extremely important as the speed is the center frequency of the filter influences, as will be described below.

On the substrate 161 are three electrically independent ones Conductors 162-164 attached. Each conductor has corresponding finger electrodes 162a to 164a, which are arranged on the substrate 161 in a comb-like manner. The finger electrodes 162a to 164a are in the same direction Distances from each other. The distance between two consecutive fingers on the same conductor is the same a wavelength of the center frequency of the filter. The speed of a surface wave in quartz is about 3300 m / s, and the center frequency of the device is about 330 MHz. The distance between two consecutive Fingers 162a is therefore, for example, approximately

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10x10 " ⁶ m.

Conductors 162 through 164 are connected to input circuit 170 of Figure 11a via lines 172 through 174 tied together. The input circuit 170 generates voltages at the electrodes 162 to 164 which are phase shifted by 120 ° are against each other. The phase relationship creates one on substrate 161 in one direction running surface wave. This means that waves add up in the forward direction building up, while waves in the opposite direction add up destructively. As a result, the component 160 has for Signals in the pass band have a low insertion loss. The attenuation by the filter component I60 is not more than 3.5 dB. US Pat. No. 3,686,518 provides further details of the construction of such a device Surface acoustic wave filter 160 indicated.

11d shows one possibility for shaping the impulse response of the surface acoustic wave filter. This method is known as the electrode finger removal procedure. In this process are selected from Sections of the substrate 161 removed groups of electrode fingers. The amplitude of the impulse response will be in these areas from which the electrode fingers are removed have been reduced below the value that would result if the electrode fingers were not removed. With This method can therefore be used to control the relative amplitude of the impulse response along the substrate 161. The desired impulse response is obtained by taking the inverse Fourier transform of the frequency response of Fig.11b is taken; the electrode fingers 162a to 164a are then selectively removed according to the desired impulse response. More details of this

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Procedures are described in U.S. Patent 3,946,342.

FIGS. 11e to 11h illustrate a method with the aid of which the values of the coils 175 to 177 and the capacitor 178 in the input circuit 170 can be determined. As already described, one of the tasks of the input circuit 170 is the generation of voltages on the lines 172 to 174, the are out of phase with each other by 120 °. Fig.11e shows that this phase shift by 120 ° by a Circuit 180 which shifts the phase between the voltage on line 172 and the voltage on line 173 of 60 ° generated, can be achieved in combination with the application of the line 74 to ground. this is further illustrated with the aid of the phase diagram of FIG. 11f.

ELne circuit for generating the phase shift of 60 ° is shown in Fig.11g. This circuit consists of an RLC circuit in the form of a π-element with a Capacitor 181, a coil 182, a capacitor 183 and a resistor 184. In Fig.11g are two equations specified, which relates to the angle of the phase adjustment relate between the voltages on lines 172 and 173 expressed by components 181-184. In the present case this angle is 60 ° and the resistor 184 is the resistance between the electrodes 162 and 163 after dividing by 2. Using of the equations given in FIG. 11g, the values for I X, I and | X «(to be calculated.

The impedance 183 is practically realized in that the Coil 175 is connected to lines 173 and 174; the impedance 182 is practically realized in that

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Sir

the coil 176 is connected to lines 172 and 173, as shown in FIG. 11h. The parallel connection of the coil 175 and the capacitance between lines 173 and 174 due to the electrodes of the surface acoustic wave device chosen so that it is equal to the impedance 183. The parallel connection of the coil 176 is in the same way and the capacitance between lines 172 and 173 due to the electrodes of the surface acoustic wave device chosen so that it is equal to the impedance 182. The inductance of coil 175 can typically be 30 nH while the inductance of coil 182 can typically be 35 nH.

The coil 177 is then placed between the leads 172 and 174 inserted and the capacitor 178 is inserted between the Lines 172 and 29 inserted so that the impedance between lines 172 and 174 is matched. the The inductance of coil 177 is typically about 25 nH and the capacitance of capacitor 174 is 10 pF.

The explanation of FIGS. 11a to 11h mainly related to the structure of the input circuit 170 and of the input transducer of the surface acoustic wave component 160. However, it can be seen that the surface acoustic wave component 16O also has an output transducer on substrate 161, which like the input transducer is constructed. The output circuit 190 of FIG. 11a is also constructed in the same way as the input circuit 170.

The circuit diagram of the RF amplifier 29 is shown in FIGS. 12a and 12b. This RF amplifier 29 consists from stages 200 and 201. Stage 200 contains a bipolar transistor 202 as an amplifying component,

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Sb

while the stage 201 as a gain element has several DC-coupled bipolar transistors in one integrated circuit 203. The circuit 203 is shown in Figure 12a as a single switching element; Fig.12b shows their details.

The amplifier 29 amplifies the signals on its input line by a maximum of 30 dB. A small part of this gain is achieved by stage 200, which has a relatively good noise figure; the rest of the amplification is effected with the aid of stage 201. The gain control signal AVRZF is applied to stages 200 and 201 via bias circuits 216 and 217, as shown in Figure 12a. The size of this gain control signal ÄVRZF changes so that the output level of the amplifier 29 is kept at about -26.5 dBm. The change in the gain of the amplifier 29 as a function of the input signal strength was previously indicated in FIG.

The output of the amplifier 29 is connected to the mixer 32, the circuit diagram of which is shown in FIG. The mixer switching element used in this mixer is a bipolar transistor 220. The emitter 221 of this transistor 220 is connected to the output of the amplifier 29. The base 223 of the transistor 220 is connected to the surface acoustic wave oscillator 35. The oscillator 35 generates mixed signals with the frequency 285 MHz at the base 223 of the transistor 220. As a result, sum and difference frequencies are generated at the collector 225 of the transistor 220. An LC resonant circuit 226 is connected to the collector 225, the resonance frequency of which is around 45 MHz. The output signal of the resonant circuit 226 is coupled via an inert tapped transformer 227 to the line 37, on which the signals S ^ (f) are generated.

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S ¥

In Figure 14a, the circuit diagram of the oscillator 35 is shown; 14b shows a schematic diagram of a resonator 36 for surface acoustic waves. The oscillator 35 contains a bipolar transistor 231 as an amplifying component. The base 232 of this transistor 231 is connected to ground, its emitter is connected to the input of the resonator 36 via a line 233, and its collector is connected to the output of the resonator 36 via a capacitor tied together. An LC circuit 236 supplies a DC supply voltage V _DD to the collector of the transistor; in addition, this LC circuit emits an output signal on line 34.

The surface acoustic wave resonator 36 has a relatively high resonance frequency of 285 MHz. Ee therefore has relative small dimensions. The component size decreases as the resonance frequency increases. Typically has the surface acoustic wave resonator of Fig. 14b only a ' Length of about 2.5 mm. The surface acoustic wave resonator also has good long-term frequency stability. this is the case because it has a large Q factor. Typically, the quality factor Q is greater than 15000. The Q factor is the ratio of the energy stored in the component per cycle to that in the Component energy consumed per cycle.

The surface acoustic wave resonator 36 includes a piezoelectric substrate 240, which in the preferred Embodiment consists of quartz. At the two ends of the substrate 240 are reflective grating structures

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and 242 attached. These grating structures form discontinuities in the surface of the substrate 240 which reflect surface waves. The grid structures 241 and 242 can consist of grooves or alternating gold or copper strips, for example. The strips are at a distance from one another which is equal to half a wavelength of the resonance frequency. Typically, each of the grid structures 241 and 242 Ire contains a total of 250 to 400 strips. The quality factor Q of the resonator increases with the number of strips. As already stated above, the speed of the surface waves in quartz is relatively insensitive to temperature changes. The resonance frequency of the resonator 36 therefore has a low temperature drift. Typically, the resonance frequency changes in a temperature range from 0 to 70 ⁰ C by less than 20 kHz.

An input transducer 243 and a are located on the substrate 240 in the space between the grating structures 241 and 242 Output transducer 244 attached. Line 233 is connected to the input transducer and to the output transducer Line 235 is connected to 244. The transducers 243 and 244 consist of several nested one inside the other Fingers, which are arranged at the tips of the standing wave present in the case of resonance, the arises due to the reflective grating structures 241 and 242. ,,

Typically there are sixty fingers on each transducer. Further details of the surface acoustic wave resonator 36 are described in U.S. Patent 3,886,504.

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The circuit diagram of the amplifier 38 is shown in FIG. The amplifier 38 consists in principle of a circuit 250, as shown in detail in FIG. 12b has been. The circuit points A to G of the circuit 250, which are shown in Figure 15, correspond the circuit points A to. G of Figure 12b. The signals from mixer 32 pass through a capacitor 251 to the input of circuit 250. The gain of circuit 250 is automatic with the help a gain control signal AVRZF set. The gain control signal AVRZF is the node C. supplied via an RLC circuit 252. The outcome of the Circuit 250 is connected to an LC resonant circuit 253 coupled to line 40.

FIGS. 16 to 19 show detailed circuit diagrams of the remaining circuit sections of the receiver from FIG Fig.1 shown. These circuits make use of conventional components; they are for that A person skilled in the art generally understandable without further explanation. The synchronous demodulator 42 is shown in FIG shown. This synchronous demodulator 42 consists essentially of a commercially available one integrated circuit of the type MC1596. The integrated The MC1596 circuit is on pages 8-404 through 8-414 of the catalog "Linear Integrated Circuits" of the "Motorola's Semiconductor Data Library? Volume 8, 1975 edition. Circuit details are given on page 8-411 in Fig. 23 of this catalog. The MC1596 integrated circuit receives its inputs and outputs via RLC circuits up to 269 suitable pretensioning as indicated in Fig. 16

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--56 -

is _o The bias voltages required for this integrated circuit are also described in the catalog mentioned

The oscillator 45 is shown in FIG. This oscillator 45 contains an MES field effect transistor 270 as an amplifying component, and it contains frequency-determining LC feedback circuits 271 and 272. The oscillator emits two separate output signals. One output signal is generated on line 47 and the other output signal is generated on line 44. The lines 44 and 47 are separated from one another by means of a RIC phase shifter element 273, which generates a phase difference of approximately 90 ° between the two signals. The phase shifter 273 ensures that the picture carrier on line 40 is in phase with the oscillator signal on the line. The phase detector circuit 43, which is shown in FIG. 18, generates the phase detector signals PD1 and PD2, which lock the phase of the oscillator signal on line 47 with a shift of 90 relative to the picture carrier on line 40. The phase shifter element i
shift back on.

The phase shifter element 273 leads this 90 ° phase

In the phase detector 43 of Fig. 18 is also of the MC1596 integrated circuit made use of which has already been used in the synchronous demodulator 42. To achieve the phase detector function, the MC1596 integrated circuit with bias circuits 281 to 287 wired. At the outputs of the integrated Circuit MC1596 are the sawtooth generator according to Fig. 18

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and the loop filter 49 connected. This circuit arrangement generates the phase detector signals PD1 and PD2 on lines 51.

In Fig.19 is the circuit diagram of the voltage controlled VHF oscillator 22 is shown. The oscillator 22 uses the phase detection signals PD1 and PD2 together with a third signal pd3 in addition, on line 23 Generate local oscillator signals with a selectable frequency. The signal PD3 is an analog signal with several values that determine a rough voltage value of a specific channel to be selected for each channel Voltage value. The signal PD3 is carried out by the controller 52 as a function of a manually Actuation of the channel selection switch 56 is generated, as was explained above in connection with FIG. The signals PD1 to PD3 are fed to the summing element 50 via a capacitance diode 290. The capacitance diode 290, together with capacitors 291 and a microstrip component 292, forms the frequency-determining device Circuit of the frequency-controlled VHF oscillator 22. The signal amplification in the oscillator 22 is with the help a bipolar transistor 293 causes. The transistor 293 is selectable by means of a control signal ENVHF switched on or off for VHF release. The ENVHF signal is passed through a resistor 294 to the base of the transistor 293 and via a coil 295 to the collector of this transistor 293. It is switched on in a selectable manner depending on the position of the channel selection switch 56.

A complete television receiver which contains the channel selector from FIG. 1 is shown in FIG. 20 in the form of a block diagram shown. The channel selector is with

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-52-

Input lines 11 and 61 provided on which he or the VHF. Receives UHF television signals as above has been described. The channel selector also has an input line 53 on which it receives analog coarse tuning voltages to display the selected channel. The RF section of the channel selector shifts the frequency of the selected channel to about 330 MHz, while the IF section of the channel selector sets the channel to 330 MHz filters out and then shifts the filtered channel into the baseband.

The line 46 forms the audio output of the channel selector. As already described, the sound carrier of the selected channel on line 46 at the frequency 4.5 MHz. The line 46 is connected to the input of a tone demodulator 301 connected. The tone demodulator 301 generates signals on a line 302 with an amplitude that the frequency of the frequency modulated signals on line 46 is proportional. This frequency demodulation can be performed with various circuits known in the art. The demodulated signals on line 302 are fed to the input of a loudspeaker 303, in which they are electromechanical can be converted into audible tones.

Line 72 forms the channel selector output for the image blanking synchronous signal. That means the signals on line 72 includes frame sync information and video information of the selected channel. the Line 72 is to the input of a video signal processing unit 304 connected. The video signal processing unit 304 separates the image signals from the image synchronizing signals. The image signals are sent to a

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QO

Line 305 is generated connected to the input of the electron gun a picture tube 306 is connected. The frame sync signals are sent to a Line 307 is generated, which is connected to the input of a control circuit 308. The control circuit generates horizontal and vertical sync signals on line 309 connected to the electron beam deflection circuit 310 of the picture tube 306 is connected. The driver circuit 308 also generates horizontal synchronization signals on a line 311 which is connected to an input of the high-voltage generator 312 of the picture tube. The driver circuit also generates synchronization signals on line 370 which is connected to the AGC circuit 74 of the channel selector is.

The structural units 301 to 312 of the television receiver have already been described in detail in many publications. For example, reference is made here to the book "Fundamentals of Display System Design" by Sol Sherr, which at Wiley-Interscience was published in 1970. One on the Bibliography given on pages 445 to 469 of this book also contains numerous references.

■ 4,

InFig.21 a second embodiment of one according to of the invention constructed channel selector shown in the form of a block diagram. An essential part This second embodiment is similar in structure to the channel selector in the embodiment of FIG. the the same sections are identified by the same reference symbols.

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A difference in the construction of the second embodiment is that there is only one VHF filter is. This means that the signals appearing at the VHF antenna 10 are fed to a local oscillator mixer be fed through a fixed filter 321 which passes all frequencies of the VHF band ^ A second difference is that the embodiment of Figure 21 does not include an RF amplifier. As a consequence of this the system has improved intermodulation and cross-modulation distortion, but is its noise figure greater. Another difference is that the embodiment of Figure 21 only has an RF MESFET mixer contains. A 2- X -1 switch 322 has a first input that feeds the VHF signals to a Line 323 receives and a second input that receives the UHF signals on line 324. The output of switch 322 is connected to MESFET mixer 320 via a line 325.

The embodiment of Figure 21 also includes another Circuit arrangement for generating the coarse selection voltages for the channel selection on the lines 53. As shown in FIG. 21 As can be seen, the controller 52 consists of a phase lock loop. This phase-locked loop receives reference signal from a reference signal circuit 331

■ 4 »

fixed frequency signals; at the same time it receives feedback signals from the voltage controlled local oscillator. The feedback signals from the oscillator are fed through a variable counter 332 transfer. The counter 332 effects a frequency division by a number, which is transmitted via digital signals on the lines 333 can be selected. The signals on the lines are dependent on a digital circuit 334

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- Ol -

from digital signals that this digital circuit receives from the channel select switches 56. A phase detector 335 compares the output signals of the counter 332 with the reference signals and generates Phase detector signals for the voltage controlled local oscillator. That way becomes a relatively high local oscillator frequency generated when the inputs of counter 332 are so chosen are that the counter divides by a relatively large number and vice versa.

There are now various embodiments of the invention has been described. Within the scope of the invention, however, modifications and changes to the specified Details to be made. For example, the intermediate frequency of the mixer output signal is not at the Limited to 330 MHz. Other intermediate frequencies in the range from 300 to 400 MHz can also be used. For example, it is also possible to use an HF amplifier with bipolar circuit elements for the MESFET HF amplifier. This is possible because the mixer has a much larger proportion of cross-modulation and intermodulation distortion in the Introduces the receiver as the linear RF amplifier · The Using a MESFET mixer results in a receiver that greatly improves the third order distortion are, even if the RF amplifier are bipolar circuit elements It is also possible to use the In contrast, generating the 285 MHz mixed signals used surface acoustic wave resonator with only one transducer to be built into a double converter version. At a

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Resonator with a transducer is the line 233 with a Electrode group of the transducer connected, while the lead 235 is connected to the other electrode group is. The individual converter is constructed in the same way as the converter 243. The channel selector according to FIG. 1 or according to FIG. 21 can be readily adapted for use in systems other than television receivers. Of the Channel selector is used wherever a frequency channel is made up of several non-overlapping frequency channels should be selected. The information contained in the channels need not be television signals,

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Claims (1)

Patent attorneys T 3110 3 Dipl.-Ing.
G. Quieter Dipl.-Ing.
E. Prince Dipl.-Chem.
Dr. G. Hauser Ernsbergerstrasse 19 8 Munich 60 July 1978 Our sign: TEXAS INSTRUMENTS INCORPORATED 13500 North Central Expressway Dallas, Texas 75222, V.St.A. Claims 1. Arrangement for filtering out a selectable frequency channel from several, non-overlapping frequency channels, with a mixer that has inputs that are simultaneous the frequency channels and mixed signals with selectable frequency to shift the frequency of the selectable Channel received at a predetermined center frequency, and a filter device that connects to the mixer having connected input and only lets through the channel with the predetermined center frequency, characterized in that the mixer is a MESFET mixer (mixer with metal-semiconductor field effect transistor) - and that the filter device is a filter for acoustic upper -_.- surface waves is. 2. Arrangement according to claim 1, characterized in that the MESFET mixer consists of MES field effect transistors a silicon substrate. - χ Schw / Ba 809884/0867 ORIGINAL INSPECTED 3. Arrangement according to claim 1, characterized in that the MESFET mixer consists of a single MES field effect transistor, the one for receiving the mixed signals connected gate electrode, one for the Reception of the source electrode connected to several frequency channels and a drain electrode connected to the input of the surface acoustic wave filter. 4. Arrangement according to claim 1, characterized in that the MESFET mixer is a mixer with a low gain factor is. 5. Arrangement according to claim 1, characterized in that the MESFET mixer the frequency of the selectable channel shifts to a range between 300 and 400 MHz. 6. An arrangement according to claim 1, characterized in that the surface acoustic wave filter is formed on a substrate, the heat-resistant piezoelectric surface waves in the temperature range of 0 ° to 70 ⁰ C transmits. 7. Arrangement according to claim 1, characterized in that the surface acoustic wave filter is formed on a quartz substrate is. 8. Arrangement according to claim 1, characterized in that the surface acoustic wave filter is an ararbeitendes with three clock phases, is a low-loss, unidirectional filter. 809884/0867 9. Arrangement according to claim 1, characterized in that the surface acoustic wave filter is a finger-like by removing Has defined impulse response electrodes. 10. Arrangement according to claim 1, characterized in that the surface acoustic wave filter in the pass band has an insertion loss of ^ 3.5 dB and in the stop band at 1.5 MHz and has an insertion loss of ^ 65 dB below the passband. 11. TV receiver with an antenna for receiving TV signals from several TV channels, a channel selector with first inputs for reception the television signals are connected to the antenna, as well as with a second input for the reception of Channel selection signals indicating the channels selected from the plurality of channels, one with the second Input connected control device, which at this input depending on the input by a Operator generates the channel selection signals, the channel selector having a first output for delivery modulated audio frequency signals of the selected channel and a second output for outputting the image blanking synchronous signals in the baseband of the selected channel, a Tonfr connected to the first output equenzdemodulator for demodulating the audio frequency signals, a loudspeaker device that is connected to the Audio frequency demodulator is connected and the demodulated audio frequency signals into audible tones converts, a video processing device which is connected to the second output and which Separates image blanking synchronous signals into image signals and image synchronization signals and a re- 8Q9884 / 0867 pre-delivery to the video processing facility is connected and the image signals and the image synchronization signals converts into television pictures, characterized in that the channel selector has a MESFET mixer (Mixer with metal-semiconductor field-effect transistor), which has inputs that simultaneously the multiple channels and mixed signals received with selectable frequency, thus the frequency of the selected Channel is shifted to a predetermined center frequency, and that the channel selector is a surface acoustic wave filter for surface acoustic waves, which has an input connected to the mixer and only the channel passes with the predetermined center frequency. 12. Television receiver according to claim 11, characterized in that the MESFET mixer consists of MES field effect transistors a silicon substrate. 13. Television receiver according to claim 11, characterized in that the MESFET mixer consists of a single MES field effect transistor consists, the one connected for the reception of the mixed signals gate electrode, one for the Reception of the several frequency channels connected to the source electrode and one to the input of the surface acoustic wave filter having connected drain electrode. t4. Television receiver according to claim 11, characterized in that that the MESFET mixer is a low gain mixer. 15. Television receiver according to claim 11, characterized in that the MESFET .Mischer the frequency of the selectable channel shifts in a range between 300 and 400 MHz. 809884/0867 16. Television receiver according to claim 11, characterized in that the surface acoustic wave filter is formed on a substrate which transmits the heat-insensitive piezoelectric surface waves in the temperature range from ⁰ to 70 °. 17. Television receiver according to claim 11, characterized in that that the surface acoustic wave filter is formed on a quartz substrate. 18. Television receiver according to claim 11, characterized in that that the surface acoustic wave filter is a working with three clock phases, low loss, effective in one direction filter is. 19. Television receiver according to claim 11, characterized in that the surface acoustic wave filter has a fingex by removing »- like electrodes has defined impulse response. 20. Television receiver according to claim 11, characterized in that the surface acoustic wave filter has an insertion loss in the passband 43f5 dB and in the stop band at 1.5 MHz above and below the pass band an insertion loss of > 65 dB. . 21.Untuned channel selector in a television receiver, characterized by an antenna connection for receiving radiated electromagnetic signals, the several TV channels contain an untuned filter with an input connected to the antenna connection for filtering out a fixed television channel spectrum from the received signals and a first mixer with inputs that simultaneously hold the fixed television channel spectrum and mixed signals received with a selectable frequency, so 809884/0867 Selectable channels from the spectrum on a high intermediate frequency which is considerably larger than 45 MHz. 22. Channel selector according to claim 21, characterized in that one of twice the value of the high intermediate frequency
plus the lowest frequency of the spectrum is outside the passband of the filter. 23. Channel selector according to claim 21, characterized in that the high intermediate frequency between 300 MHz and 400 MHz
lies. 24. Channel selector according to claim 21, characterized in that the fixed spectrum is approximately the range from 55 to 87 MHz
includes. 25. Channel selector according to claim 21, characterized in that the fixed spectrum is approximately the range from 175 to 215 MHz
includes? 26. Channel selector according to claim 21, characterized in that the fixed spectrum is approximately the range from 470 to 890 MHz
includes. 27. Channel selector according to claim 21, characterized in that the mixer to achieve the high intermediate frequency is an MES-FEfc mixer (Mixer with metal-semiconductor field-effect transistor) is. 28. Channel selector according to claim 27, characterized in that the MESFET mixer is a silicon MESFET mixer. 809884/0867 29ν channel selector according to claim 21, characterized by a Surface acoustic wave filter for surface acoustic waves with one connected to the output of the first mixer Input for filtering out the selectable channel at the high intermediate frequency. 30. Channel selector according to claim 29, characterized in that the surface wave filter is a piezoelectric substrate material whose coupling coefficient is considerably lower than the coupling coefficient of lithium niobate. 31. Channel selector according to claim 29, characterized in that the surface acoustic wave filter contains a quartz substrate. 32. Channel selector according to claim 29, characterized in that the surface acoustic wave filter is a relatively small piezoelectric Contains substrate on which a plurality of electrodes are attached at a distance from one another, the high intermediate frequency is inversely proportional. 33. Channel selector according to claim 29, characterized by a second Mixer with inputs that simultaneously contain the filtered channel with the high intermediate frequency and a second mixed signal with a predetermined fixed frequency received so that the frequency of the filtered out channel with the high Intermediate frequency to a much lower second intermediate frequency is moved. 34. Television receiver with an antenna for receiving TV signals from several TV channels, a channel selector with first inputs for receiving the TV signals are connected to the antenna, and with a second input for receiving channel selection signals, the display the channels selected from the multiple channels, 8 09884/0867 a control device connected to the second input, which at this input as a function of the input by an operator generates the channel selection signals, the channel selector having a first output for outputting modulated Audio frequency signals of the selected channel and a second output for outputting the image blanking synchronous signals in the baseband of the selected channel, an audio frequency demodulator connected to the first output. to the Demodulating the audio frequency signals, a loudspeaker device, which is connected to the audio frequency demodulator and the demodulated audio frequency signals into audible Tones converts, a video processing device, which is connected to the second output, and the image blanking synchronous signals separates into image signals and image synchronization signals and a playback device connected to the Video processing device is connected and the image signals and the image synchronization signals in television images converts, characterized in that the channel selector contains an untuned filter device that is one with the Antenna-connected input contains, and filters out a fixed television channel spectrum from the received signals, and that the channel selector contains a first mixer which has inputs which simultaneously the fixed television signal spectrum and receiving mixed signals at a selectable frequency, thus the frequency of the selectable channels, of the spectrum be shifted to a high intermediate frequency, which is much greater than 45 MHz. 35. Television receiver according to claim 34, characterized in that one of twice the value of the high intermediate frequency ■ the quantity formed plus the lowest frequency of the spectrum lies outside the passband of the filter «, 809884/0867 36. Television receiver according to claim 34, characterized in that the high intermediate frequency between 300 and 400 MHz lies. 37. Television receiver according to claim 34, characterized in that the fixed spectrum, approximately the range from 55 to 87 MHz includes. 38. Television receiver according to claim 34, characterized in that the fixed spectrum is approximately the range from 175 to 215 MHz includes. 39. television receiver according to claim 34, characterized in that that the fixed spectrum covers approximately the range from 470 to 890 MHz. 40. Television receiver according to claim 34 _f, characterized in that the mixer to achieve the high intermediate frequency is a MESFET mixer (mixer with metal-semiconductor field-effect transistor). 41. Television receiver according to claim 34, characterized in that the MESFET mixer is a silicon MESFET mixer. 42. Television receiver according to claim 34, characterized by · .. a surface acoustic wave filter for surface acoustic waves with one connected to the output of the first mixer Input for filtering out the selectable channel at the high intermediate frequency. 809884/0867 43. Television receiver according to claim 42, characterized in that the surface acoustic wave filter is a piezoelectric substrate material contains, the coupling coefficient of which is considerably lower than the coupling coefficient of lithium niobate is. 44. Television receiver according to claim 42, characterized in that that the surface acoustic wave filter contains a quartz substrate. 45. television receiver according to claim 42, characterized in that that the surface acoustic wave filter contains a relatively small piezoelectric substrate on which several electrodes are mounted at a distance from one another which is inversely proportional to the high intermediate frequency. 46. Television receiver according to claim 42, characterized by a second mixer with inputs that simultaneously filtered channel with the high intermediate frequency and a second mixed signal with a predetermined fixed Frequency received so that the frequency of the filtered out channel with the high intermediate frequency becomes a substantial lower second intermediate frequency is shifted. 47. Channel selector, characterized by an antenna connection a spectrum filter for the reception of radiated electromagnetic signals that contain several television channels, with an input connected to the antenna connection for filtering out a channel spectrum from several, a mixer with inputs connected to the spectrum filter for shifting the frequency of selected channels of the spectrum to a predetermined intermediate frequency and a channel selection filter with an input connected to the mixer to filter out the channel at the predetermined intermediate 809884/0867 frequency, whereby the channel selector has an overall gain from the antenna connection via the channel selection filter, which is not greater than the gain necessary to achieve a desired noise figure. 48ο channel selector according to claim 47, characterized in that the total gain is less than 10 dB. 49 *. Channel selector according to Claim 47, characterized in that the mixer is a low gain mixer, its cross modulation distortion and intermodulation distortion for input signals with 0 dBm are less than 1%. 50o channel selector according to claim 47, characterized in that the mixer is a low gain MESFET mixer. 51. channel selector according to claim 47, characterized in that the channel selection filter is a surface acoustic wave filter Surface waves is. 52. Channel selector according to claim 51 »characterized in that the surface acoustic wave filter is a low-loss, with three Is clock phase working, effective in one direction filter. 53. Channel selector according to claim 47, characterized in that the predetermined intermediate frequency is in the range between 300 and 400 MHz. 54. Channel selector according to Claim 53, characterized in that the spectrum filter is a tuned fixed band-pass filter ^is * 809884/0867 55. Channel selector according to claim 47, characterized by a second mixer with one connected to the channel selection filter Input for shifting the frequency of the channel with the predetermined intermediate frequency to one second intermediate frequency and an amplifier device with an input connected to the second mixer to amplify the signals with the second intermediate frequency far beyond the total amplification. 56. Channel selector according to claim 55, characterized in that the predetermined intermediate frequency is substantially greater than is the second intermediate frequency and that the second intermediate frequency is about 45 MHz. 57. television receiver with an antenna for receiving television signals from several television channels, one Channel selector with first inputs that are connected to the antenna for receiving television signals, and with a second input for receiving channel selection signals from the multiple channels display selected channels, a control device connected to the second input, which is connected to this The input generates the channel selection signals as a function of the input by an operator, the channel selector having a first output for outputting modulated audio frequency signals of the selected channel and a second output for outputting the image blanking synchronous signals in the baseband of the selected channel, a sound connected to the first output t frequency demodulator for demodulating the audio frequency signals, a loudspeaker device which is connected to the Tonf re quenzdemodula tor and the 809884/0867 demodulated audio frequency signals into audible tones converts, a video processing device which is connected to the second output and which Separates picture blanking synchronous signals into picture signals and picture synchronizing signals and a reproducing device, which is connected to the video processing device and the image signals and the image synchronization signals converts into television pictures, characterized in that the channel selector has a spectrum filter with a connected to the antenna Input for filtering out a channel spectrum from several contains that the channel selector contains a mixer which has inputs connected to the spectrum filter and the frequency of selected channels of the spectrum shifts to a predetermined intermediate frequency, and that the channel selector is a channel selection filter with an input connected to the mixer for filtering out the channel with the predetermined intermediate frequency does not contain the overall gain of the channel selector from the antenna via the channel selection filter is greater than the gain required to achieve a desired noise figure. 58. Television receiver according to claim 57, characterized in that that the total gain is less than 10 dB Ü. • s, 59. Television receiver according to claim 57, characterized in that the mixer is a mixer with low gain, its cross modulation distortion and intermodulation distortion be less than 1% for input signals with 0 dBm. 809884/08 6 7 Atf 60. Television receiver according to claim 57 »characterized in that that the mixer is a low gain MESFET mixer. 61. Television receiver according to claim 57, characterized in that the channel selection filter is a surface acoustic wave filter for surface acoustic waves. 62. Television receiver according to claim 61, characterized in that the surface acoustic wave filter is a low-loss, is a unidirectional filter working with three clock phases. 63. Television receiver according to claim 57> characterized in that the predetermined intermediate frequency is in the range between 300 and 400 MHz. 64. Television receiver according to claim 63, characterized in that that the spectrum filter is a tuned fixed band-pass filter. 65. Television receiver according to claim 57, characterized by a second mixer with an input connected to the channel selection filter for shifting the frequency of the channel at the predetermined intermediate frequency to a second intermediate frequency and a Amplifier device with an input connected to the second mixer for amplifying the signals with the second intermediate frequency far beyond the total gain. 66. Television receiver according to claim 65, characterized in that the predetermined intermediate frequency is essential 809884/0867 is greater than the second intermediate frequency and that the second intermediate frequency is about 45 MHz amounts to. 809884/0867