CN110365611B - Doppler frequency shift estimation method and device - Google Patents

Doppler frequency shift estimation method and device Download PDF

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CN110365611B
CN110365611B CN201910458387.8A CN201910458387A CN110365611B CN 110365611 B CN110365611 B CN 110365611B CN 201910458387 A CN201910458387 A CN 201910458387A CN 110365611 B CN110365611 B CN 110365611B
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cumulant
received signal
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frequency shift
angle
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CN110365611A (en
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许文俊
郑文卿
高晖
冯志勇
徐雅倩
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Beijing University of Posts and Telecommunications
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
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    • H04L25/024—Channel estimation channel estimation algorithms
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
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Abstract

The invention discloses a Doppler frequency shift estimation method and a device, which calculate the cumulant angle estimation value of a received signal, calculate the cumulant frequency shift estimation value of the received signal, and correct the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value. The invention can accurately estimate the Doppler frequency shift value under the condition of multiple antennas, improve the estimation accuracy and improve the accuracy of signal recovery.

Description

Doppler frequency shift estimation method and device
Technical Field
The present invention relates to the field of wireless communication technologies, and in particular, to a doppler frequency shift estimation method and apparatus.
Background
Doppler frequency shift estimation has great significance for the recovery of distorted signals at a receiving end in wireless communication. The wireless communication channel is mostly a time-varying fading channel, the amplitude and the phase decay speed of the received signal depend on the magnitude of the doppler frequency shift, and the larger the value is, the faster the fading speed of the wireless communication channel is. In a wireless communication system, real-time estimation of Doppler frequency offset is of great significance to obtain optimal receiving performance. The Doppler frequency shift estimation technology is widely applied to the fields of communication system parameter selection, channel estimation and self-adaptive methods.
The estimation method based on the cyclostationarity is a Doppler frequency shift estimation method, and adopts a method of calculating a signal cyclostationarity power spectrum to extract Doppler characteristics. The existing cyclostationary estimation method has the problems of poor noise immunity and inaccurate estimation, generally assumes that a single antenna is configured at a sending end and a receiving end, cannot estimate the arrival angle of a wave beam and cannot obtain the Doppler frequency shift directly generated by the moving speed from the Doppler frequency shift.
Disclosure of Invention
In view of this, the present invention provides a doppler frequency shift estimation method and device based on fourth-order cyclostationary cumulant, which can improve the anti-noise performance and improve the estimation accuracy.
Based on the above object, the present invention provides a doppler shift estimation method, including:
carrying out cumulant angle estimation on the received signal to obtain a cumulant angle estimation value;
carrying out cumulant frequency shift estimation on the received signal to obtain a cumulant frequency shift estimation value;
and correcting the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value.
Optionally, the antenna array for receiving the received signal includes two uniform linear arrays, each array includes 2M +1 isotropic antennas, and a distance between two antennas in the same array is dxThe spacing between two columns being dy。
Optionally, let L be the total number of paths existing under the multipath condition, and the three-dimensional arrival angle of the ith received signal is:
Figure GDA0002600677230000021
Figure GDA0002600677230000022
Figure GDA0002600677230000023
wherein,
Figure GDA0002600677230000024
is represented by betaiIs determined by the estimated value of (c),
Figure GDA0002600677230000025
denotes thetaiIs determined by the estimated value of (c),
Figure GDA0002600677230000026
to represent
Figure GDA0002600677230000027
An estimated value of (d); λ is signal wavelength, ξiThe characteristic value of the cumulant matrix of the received signal corresponding to the ith path;
Figure GDA0002600677230000028
wherein h isj(M) represents the mth element of the jth eigenvector of eigenvalues, M being smaller than M.
Optionally, the fourth-order cyclic spectrum of the received signal is:
Figure GDA0002600677230000029
where f is the frequency offset range of the signal, T is the sampling period of the signal, A is a constant determined by the Rice factor of the channel, σwIs the power of the transmitted signal.
Optionally, the estimator based on the transmit-end pilot frequency is:
Figure GDA00026006772300000210
the estimator without pilot frequency at the transmitting end is as follows:
Figure GDA00026006772300000211
optionally, the calculation method of the doppler shift estimate value is as follows:
Figure GDA0002600677230000031
where θ is the estimated angle shown in equation (18).
Optionally, the cumulant matrix is:
Figure GDA0002600677230000032
Figure GDA0002600677230000033
note C1Is a pseudo-inverse matrix of
Figure GDA0002600677230000034
Then
Figure GDA0002600677230000035
Opens up a signal subspace.
Optionally, for one path of the received signal, the received signal is processed to obtain a fourth-order cumulative amount of the received signal, the fourth-order cumulative amount of the received signal is multiplied by a plurality of sine wave signals within a certain frequency range to obtain a plurality of corresponding demodulated signals, a frequency with the maximum correlation is selected from the plurality of demodulated signals to be used as a cyclic frequency for extracting doppler frequency shift, after the sine wave signal with the cyclic frequency is multiplied by the received signal, an expectation is obtained for the obtained product signal, and then fast fourier transform is performed to obtain a fourth-order cyclic spectrum of the received signal shown in formula (40).
Optionally, the calculation of the cumulative angle estimation value and the calculation of the cumulative frequency shift estimation value are performed on all received signals of each antenna of the antenna array, and then the doppler shift estimation value is calculated.
An embodiment of the present invention further provides a doppler shift estimation device, including:
the first calculation module is used for carrying out cumulant angle estimation on the received signal to obtain a cumulant angle estimation value;
the second calculation module is used for carrying out cumulant frequency shift estimation on the received signal to obtain a cumulant frequency shift estimation value;
and the third calculation module is used for correcting the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value.
As can be seen from the above description, the doppler shift estimation method and apparatus provided by the present invention calculate the cumulant frequency shift estimation value of the received signal by calculating the cumulant angle estimation value of the received signal, and correct the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain the corrected doppler shift estimation value. The invention can accurately estimate the Doppler frequency shift value under the condition of multiple antennas, improve the estimation accuracy and improve the accuracy of signal recovery.
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In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings used in the description of the embodiments or the prior art will be briefly described below, it is obvious that the drawings in the following description are only some embodiments of the present invention, and for those skilled in the art, other drawings can be obtained according to the drawings without creative efforts.
FIG. 1 is a schematic flow chart of a method according to an embodiment of the present invention;
FIG. 2 is a schematic flow chart of a cumulant angle estimation method according to an embodiment of the invention;
fig. 3 is a schematic diagram of an antenna array structure according to an embodiment of the present invention;
FIG. 4 is a block diagram of an accumulated amount frequency shift estimation module according to an embodiment of the present invention;
FIG. 5 is a block diagram of an apparatus according to an embodiment of the present invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to specific embodiments and the accompanying drawings.
It should be noted that all expressions using "first" and "second" in the embodiments of the present invention are used for distinguishing two entities with the same name but different names or different parameters, and it should be noted that "first" and "second" are merely for convenience of description and should not be construed as limitations of the embodiments of the present invention, and they are not described in any more detail in the following embodiments.
FIG. 1 is a schematic flow chart of a method according to an embodiment of the present invention. As shown in the figure, the doppler shift estimation method provided in the embodiment of the present invention includes:
s10: carrying out cumulant angle estimation on the received signals;
fig. 2 is a schematic flow chart of an cumulant angle estimation method according to an embodiment of the present invention. As shown in the figure, the calculation of the accumulated angle estimation for the received signal specifically includes the following steps:
s101: determining an accumulated amount of received signals;
fig. 3 is a schematic diagram of an antenna array structure according to an embodiment of the present invention. The antenna array is arranged at the receiving end and consists of Xa、YaTwo groups of Uniform Linear Arrays (ULA), each column of ULA comprising 2M +1 isotropic antennas, with a distance d between each two antennasxThe antenna array direction is defined as y-axis, and the antenna array XaThe central antenna of (2) is defined as a coordinate origin O, and the two antenna arrays are increased from-M to M along the positive direction of the y-axis with the central antenna as zero. Two columns of ULAs are aligned in parallel, the interval between two columns of ULAs is dyThe origin of coordinates O points to the antenna array YaThe direction of the midpoint antenna is defined as the x-axis, and the z-axis is defined by the x-axis and the y-axis according to the rule of the right-hand coordinate system. Where θ represents the angle between the arriving signal and the y-axis, β represents the angle between the arriving signal and the x-axis, and φ represents the angle between the arriving signal and the z-axis. When there are L directions of arrival, the angles are each θi,，βi，φi，i＝1,2,...,L。
In the antenna array, at any time t, the received signal of the antenna m is marked as xm(t)、ym(t), defining the cumulative amount:
Figure GDA0002600677230000051
Figure GDA0002600677230000052
wherein x is0(t) being central antennaThe signal is received and the received signal is transmitted,
Figure GDA0002600677230000053
is x0(t) conjugate signal.
The fourth order cumulant performed by the channel including the doppler shift can be expressed as:
cum[a,b,c,d]＝E[abc*d*]-E[ab*]E[cd*]-E[ac*]E[bd*]-E[ad*]E[bc*] (3)
wherein a, b, c and d are four complex random variables of cumulant to be calculated (.)*Representing the conjugate of a variable, i.e. the quadrature component of the signal is inverted and the parallel component is kept constant.
S102: calculating an accumulation matrix of the received signals;
the cumulant matrix is:
Figure GDA0002600677230000054
Figure GDA0002600677230000055
note C1Is a pseudo-inverse matrix of
Figure GDA0002600677230000056
Exactly span the signal subspace,
Figure GDA0002600677230000057
and the guide vector of the non-zero characteristic vector and matrix
Figure GDA0002600677230000058
Where M ═ M, -M + 1.., M, λ denote the signal wavelength. Beta is aiIs the direction of arrival angle to be estimated. Thus, pair
Figure GDA0002600677230000059
Decomposing the eigenvalues to obtain eigenvectorshiThen, a three-dimensional angle of arrival value can be obtained.
Wherein,
Figure GDA0002600677230000061
the proof that the eigenvectors exactly span the signal subspace is as follows:
the antenna element accumulation is defined as:
Figure GDA0002600677230000062
Figure GDA0002600677230000063
there are P directions of incoming wave, A ═ aT(θ1),aT(θ2),...,aT(θP)]TRepresents a set of drive vectors, wherein
Figure GDA0002600677230000064
D＝diag(d1,d2,...,dP) Wherein d isiIs a constant. Binding of ATAnd D, obtaining:
C1＝ADA
C2＝AVDAH (8)
to obtain:
DAg H＝(Ag HAg)-1Ag HC1 (9)
substitution of formula (8) gives:
C2＝AgVDAg H＝AgV(Ag HAg)-1Ag HC1 (10)
right ride on two sides
Figure GDA0002600677230000071
Obtaining:
Figure GDA0002600677230000072
assume that the eigenvalues of C1 and its pseudo-inverse decompose as follows:
Figure GDA0002600677230000073
Figure GDA0002600677230000074
substituting formulae (12) and (13) for formula (11) yields:
Figure GDA0002600677230000075
the equivalence is as follows:
Figure GDA0002600677230000076
the drive vectors a (θ) are based on equation (15)i) The spanned subspace and the signal subspace are identical.
S103: carrying out eigenvalue decomposition on the cumulant matrix;
the characteristic value decomposition process comprises the following steps:
Figure GDA0002600677230000077
wherein L is the total number of paths, ξ, existing under multipath conditionsiIs the eigenvalue corresponding to the ith path, hiAs a characteristic value xiiThe corresponding feature vector is used as a basis for determining the feature vector,
Figure GDA0002600677230000078
is hiThe conjugate transpose of (c).
S104: a three-dimensional angle of arrival of the received signal is calculated.
Calculating the three-dimensional arrival angle of the ith path of received signal:
Figure GDA0002600677230000081
Figure GDA0002600677230000082
Figure GDA0002600677230000083
wherein,
Figure GDA0002600677230000084
is represented by betaiIs determined by the estimated value of (c),
Figure GDA0002600677230000085
denotes thetaiIs determined by the estimated value of (c),
Figure GDA0002600677230000086
to represent
Figure GDA0002600677230000087
An estimate of (d).
Figure GDA0002600677230000088
Wherein h isj(M) denotes the mth element of the jth feature vector, M being smaller than M.
S11: carrying out cumulant frequency shift estimation on a received signal;
for a single-antenna system, if a baseband envelope of a transmitted signal is s (t), channel fading is h (t, τ), and a noise signal is v (t), a received signal is:
Figure GDA0002600677230000089
calculating fourth-order cumulant of the channel containing the Doppler frequency shift to obtain:
c4h＝E[h(t)h(t+τ1)h*(t+τ2)h*(t+τ3)]-E[h(t)h(t+τ1)]E[h*(t+τ2)h*(t+τ3)]-E[h(t)h*(t+τ2)]E[h(t+τ1)h*(t+τ3)]-E[h(t)h*(t+τ3)]E[h(t+τ1)h*(t+τ2)] (22)
the fourth order cumulant of the resulting channel is expressed as follows:
Figure GDA0002600677230000091
wherein f ismRepresenting the Doppler frequency offset, J0Representing a zero-order Bessel function, omegap、Ω4pConstants determined for the second and fourth order statistics of the channel, respectively.
To simplify the calculation, the following convention is adopted:
τ1＝0,τ2＝τ3＝τ (24)
simplifying to obtain:
Figure GDA0002600677230000092
and (3) solving Fourier transform of the formula (25) to obtain an accumulated quantity spectrum of the channel under the Rayleigh fading environment:
Figure GDA0002600677230000093
when a direct path exists (i.e., in the rice channel environment), the channel fading envelope becomes: h (t) ═ hRay(t)+hLOS(t) wherein hLOS(t) is a Rayleigh fading, and the cross-correlation of the Rayleigh fading is 0,
Figure GDA0002600677230000094
center of spectral line is
Figure GDA0002600677230000095
Width of 4fmIn the presence of direct radiation, two impacts exist in the spectral line
Figure GDA0002600677230000101
And
Figure GDA0002600677230000102
location.
The fourth order cumulant of the direct path component is:
Figure GDA0002600677230000103
the fourth order cumulative spectrum of the fading envelope in the presence of the direct path is obtained as:
Figure GDA0002600677230000104
it can be seen that the spectral width of the cumulant is four times the Doppler frequency when the direct path is not present, and that the cumulant has an impulse line at the cosine product between the Doppler frequency and the angle of arrival when the direct path is present, i.e., f + -2 f in equation (28)mcosθ0。
Calculating fourth-order cumulant of the received signal containing Doppler frequency shift to obtain:
c4z＝E[z(t)z(t+τ1)z*(t+τ2)z*(t+τ3)]
-E[z(t)z(t+τ1)]E[z*(t+τ2)z*(t+τ3)]
-E[z(t)z*(t+τ2)]E[z(t+τ1)z*(t+τ3)]
-E[z(t)z*(t+τ3)]E[z(t+τ1)z*(t+τ2)]
(29)
according to the convention shown in equation (24), simplifying equation (29) yields:
c4z＝E[z2(t)z2(t+τ)]-E[z2(t)]E[z2(t+τ)]-2E2[z(t)z(t+τ)] (30)
fig. 4 is a block diagram of a structure of an accumulated amount frequency shift estimation module according to an embodiment of the present invention. As shown in the figure, each path of received signals of the antenna array is respectively input into the cumulant frequency shift estimation module, and a fourth-order cyclic spectrum of the path of received signals is obtained after processing. Specifically, one path of a received signal z (t) is delayed by a time delay unit tau to obtain z (t + tau), the other path of the received signal z (t) and the delayed signal z (t + tau) are subjected to a multiplier to obtain an expectation, and then the expectation is obtained through a squarer to obtain E2[z(t)z(t+τ)]Multiplying by 2 to obtain 2E2[z(t)z(t+τ)]After taking the inverse, inputting the inverse to an adder; the received signal z (t) passes through a squarer to obtain z2(t) for z2(t) obtaining the desired result E [ z ]2(t)](ii) a The time-delayed signal z (t + tau) is processed by a squarer to obtain z2(t+τ)，z2(t) and z2After (t + tau) passes through the multiplier, the desired E [ z ] is obtained2(t)z2(t+τ)]Inputting the signals into an adder; to z2(t + T) to obtain the desired value E [ z ]2(t+τ)]Is a reaction of E [ z ]2(t)]And E [ z ]2(t+τ)]Input multiplier to obtain E z2(t)]E[z2(t+τ)]After the inversion, the input signal is input to an adder, and the adder calculates the input three terms to obtain the fourth-order cumulative quantity of the noise suppression of the received signal shown in the formula (30).
As shown in fig. 4, the fourth-order cumulative quantity c of the received signal output from the adder4zWith a sine wave signal ej2πkt/TMultiplying, demodulating and extracting the sending signal. Wherein k is the frequency of the sine wave signal, T is the time of the sine wave signal, T is the signal sampling period, and the fourth-order cumulant c of the received signal4zSinusoidal signals e corresponding to frequency values within a certain frequency range tauj2πkt/TThe demodulated signals are multiplied to obtain respective demodulated signals, and the frequency having the maximum correlation is selected from the demodulated signals as a cyclic frequency for extracting the doppler shift.
Multiplying the extracted sine wave signal with the cyclic frequency with the received signal, obtaining the expectation of the obtained product signal, and obtaining the expectation value according to the distribution shown in the formula (22) and the specification shown in the formula (24) for the specific time delay tau to obtain C4z(k, τ), see formula (38).
Then, the desired received signal C is obtained4zPerforming Fast Fourier Transform (FFT) calculation on tau of (k, tau) to obtain a cyclic spectrum omega of a fourth-order cyclic quantity4z(k,f)∣k＝2The cyclic frequency point k-2 is selected based on experimental results, and waveforms obtained by correlating the k-2 frequency points have higher signal-to-noise ratio than other k values under the fourth-order cyclic spectrum.
In the embodiment of the present invention, the relationship between the accumulated amount of the received signal and the accumulated amount of the transmitted signal is:
when noise is not considered, the received signal is
Figure GDA0002600677230000111
The fourth order cumulant at z (t) is thus calculated:
c4z＝c4h(τ1,τ2,τ3)c4s(t,τ1,τ2,τ3)-rh(τ1)rh(τ2-τ3)rs(t,τ1)rs(t+τ2,τ3-τ2)-rh(τ2)rh(τ3-τ1)rs(t,τ2)rs(t+τ3,τ1-τ3)-rh(τ3)rh(τ3-τ1)rs(t,τ3)rs(t+τ1,τ2-τ1) (31)
when pilot frequency exists, the second-order cyclic spectrum of the pilot frequency is as follows:
Figure GDA0002600677230000112
wherein σwIs the power of the transmitted signal, i.e. s when the transmitted signal is BPSK modulatedn＝(-1)nσw。
Using the convention shown in equation (24), the fourth order cumulant of the pilot can be calculated as:
c4s(t,τ)＝E[s(t)s*(t+τ)s(t)s*(t+τ)]-0-2E2[s(t)s*(t+τ)]＝-[rs(t,τ)]2 (33)
wherein r iss(t, τ) is the second order circulation quantity, also ΩsTime domain form of (1, f). Fourier transform equation (33) yields:
Figure GDA0002600677230000121
equation (31) can be simplified as:
Figure GDA0002600677230000122
and (3) solving a cycle by multiplying the above expression by a sine wave with a cycle frequency of alpha to obtain:
Figure GDA0002600677230000123
wherein,
Figure GDA0002600677230000124
obtaining:
C4z(k,τ)＝c4h(τ)C4s(k,τ)-Aej2πατ (38)
where a is a constant determined by the channel rice factor.
And (3) performing Fourier transform on the tau in the step (38) to obtain a cyclic spectrum of fourth-order cumulant of the receiving end, wherein the cyclic spectrum is the convolution of the spectrum of the signal at the transmitting end and the channel spectrum. Where k is the frequency order of the cycle frequency, first order timeIndicating that the cyclic frequency is equal to the symbol rate of the transmitted signal, and the k-th order indicates that the cyclic frequency is k times the symbol rate of the transmitted signal. Assuming that the transmitted signal is a specific pilot, the pilot pattern is sn＝(-1)nThe cyclic spectrum of the transmitted signal at this time is:
Figure GDA0002600677230000131
substituting the formula (39) into the formula (38) to obtain a fourth-order cyclic spectrum expression of the received signal, wherein the fourth-order cyclic spectrum expression is as follows:
Figure GDA0002600677230000132
wherein
Figure GDA0002600677230000133
The symbol rate of the signal, it follows that the fourth order cyclic spectrum of the received signal shifts to the center of the transmission symbol rate for the channel fading spectrum. Because the width of the channel fading frequency spectrum and the spectral line have the information of Doppler frequency shift, the Doppler frequency shift can be obtained by measuring the spectral line width and the peak position.
The estimator based on the transmitting end pilot frequency is obtained as follows:
Figure GDA0002600677230000134
the estimator without pilot frequency at the transmitting end is as follows:
Figure GDA0002600677230000135
as shown in fig. 4, the FFT computation is performed for different values of the delay τ, assuming that the doppler shift resolution (i.e. the estimation error range or the search network size to be obtained) is:
Figure GDA0002600677230000136
where T is the symbol period of the transmitted signal, fDT is normalized Doppler frequency, P is any integer, and reference interval is P epsilon [10 ∈1,103]. In order to achieve sufficient precision, the number of fast fourier transform FFT points used by the receiving end should be N-20P, and the maximum value of the delay is τmax10PT, the value range of instant delay is
Figure GDA0002600677230000137
In order to obtain different time delay values tau, the time delay device can be buffered for N times and calculated once, or the time delay device can be calculated once
Figure GDA0002600677230000138
Secondary cache, each cache calculating continuously
Figure GDA0002600677230000139
Next, the process is carried out. Note that the latter has less spatial complexity but greater estimated delay. Wherein, is performed N times or
Figure GDA00026006772300001310
The secondary buffering is to obtain the N-point data required for FFT computation.
The method for reading the cyclic spectrum width comprises the steps of converting a time domain signal into a fourth-order cumulant cyclic frequency domain by fast Fourier transform, and according to estimators (41) and (42) at a transmitting end:
the center of the circulation spectrum is positioned
Figure GDA0002600677230000141
Due to the periodicity of the phase, when the absolute value of the maximum Doppler shift exceeds
Figure GDA0002600677230000142
Is detected as
Figure GDA0002600677230000143
To
Figure GDA0002600677230000144
Other values within, thus selecting
Figure GDA0002600677230000145
Is half the spectral width maximum (actual system doppler shift rarely arrives)
Figure GDA0002600677230000146
). Thus, the position of the occurrence of the maximum of the cyclic spectrum is at the result of the fast Fourier transform
Figure GDA0002600677230000147
And
Figure GDA0002600677230000148
in the method, the point number of the fast Fourier transform is respectively in the interval of 50M < n < 10M and 10M < n < 15M, two maximum values of amplitude values (the cyclic spectrum can be complex, and the maximum value of the absolute value of the complex can be taken) are respectively searched in the two intervals, then the corresponding frequency values are subtracted, the cyclic spectrum width is obtained by normalizing to the frequency value, and the cyclic spectrum width is the cumulant Doppler frequency shift estimation quantity and represents the projection value of the Doppler frequency shift in the direction of arrival.
S12: and correcting the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value.
Inputting each output of the cumulant Doppler frequency shift estimation module of each antenna of the antenna array into a Doppler frequency estimator, and performing correction processing to correct the corrected Doppler frequency into a radial accurate value (namely, compensating the projection deviation caused by the fact that the arrival direction of the beam is not vertical to the antenna array):
the Doppler frequency estimator of the embodiment of the invention comprises the following steps:
Figure GDA0002600677230000149
wherein f is the frequency offset range of FFT output, and T is the sampling period of the signal. θ is the estimated angle shown in equation (18).
FIG. 5 is a block diagram of an apparatus according to an embodiment of the present invention. As shown in the figure, an embodiment of the present invention further provides a doppler shift estimation device, including:
the first calculation module is used for carrying out cumulant angle estimation on the received signal to obtain a cumulant angle estimation value;
the second calculation module is used for carrying out cumulant frequency shift estimation on the received signal to obtain a cumulant frequency shift estimation value;
and the third calculation module is used for correcting the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value.
According to the Doppler frequency shift estimation method and device provided by the embodiment of the invention, the cumulant frequency shift estimation value of the received signal is calculated by calculating the cumulant angle estimation value of the received signal, and the cumulant angle estimation value and the cumulant frequency shift estimation value are corrected to obtain the corrected Doppler frequency shift estimation value. The invention can accurately estimate the Doppler frequency shift value under the condition of multiple antennas, improve the estimation accuracy and improve the accuracy of signal recovery.
The apparatus of the foregoing embodiment is used to implement the corresponding method in the foregoing embodiment, and has the beneficial effects of the corresponding method embodiment, which are not described herein again.
Those of ordinary skill in the art will understand that: the discussion of any embodiment above is meant to be exemplary only, and is not intended to intimate that the scope of the disclosure, including the claims, is limited to these examples; within the idea of the invention, also features in the above embodiments or in different embodiments may be combined, steps may be implemented in any order, and there are many other variations of the different aspects of the invention as described above, which are not provided in detail for the sake of brevity.
In addition, well known power/ground connections to Integrated Circuit (IC) chips and other components may or may not be shown within the provided figures for simplicity of illustration and discussion, and so as not to obscure the invention. Furthermore, devices may be shown in block diagram form in order to avoid obscuring the invention, and also in view of the fact that specifics with respect to implementation of such block diagram devices are highly dependent upon the platform within which the present invention is to be implemented (i.e., specifics should be well within purview of one skilled in the art). Where specific details (e.g., circuits) are set forth in order to describe example embodiments of the invention, it should be apparent to one skilled in the art that the invention can be practiced without, or with variation of, these specific details. Accordingly, the description is to be regarded as illustrative instead of restrictive.
While the present invention has been described in conjunction with specific embodiments thereof, many alternatives, modifications, and variations of these embodiments will be apparent to those of ordinary skill in the art in light of the foregoing description. For example, other memory architectures (e.g., dynamic ram (dram)) may use the discussed embodiments.
The embodiments of the invention are intended to embrace all such alternatives, modifications and variances that fall within the broad scope of the appended claims. Therefore, any omissions, modifications, substitutions, improvements and the like that may be made without departing from the spirit and principles of the invention are intended to be included within the scope of the invention.

Claims (6)

1. A Doppler frequency shift estimation method, characterized in that, an antenna array for receiving a received signal comprises Xa、YaTwo lines of uniform linear arrays, each line containing 2M +1 isotropic antennas, the distance between two antennas in the same line being dxThe spacing between two columns being dyEstablishing an XYZ coordinate system based on the antenna array; the method comprises the following steps:
carrying out cumulant angle estimation on a received signal to obtain a cumulant angle estimation value, wherein the cumulant angle estimation value comprises the following steps:
determining the cumulative amount of received signals: the cumulative amount is defined as:
Figure FDA0002625458910000011
Figure FDA0002625458910000012
xm(t)、ym(t) is the received signal of antenna m, x, at any time tO(t) is the received signal of the central antenna,
Figure FDA0002625458910000013
is x0(ii) the conjugate signal of (t),
Figure FDA0002625458910000014
is xm(t) conjugate signal;
calculating an accumulation quantity matrix of the received signals according to the accumulation quantity:
carrying out eigenvalue decomposition on the cumulant matrix;
Figure FDA0002625458910000015
wherein, C2In the form of a matrix of accumulated quantities,
Figure FDA0002625458910000016
is a cumulant matrix C1L is the total number of paths existing under multipath conditions, xiiIs the eigenvalue corresponding to the ith path, hiAs a characteristic value xiiThe corresponding feature vector is used as a basis for determining the feature vector,
Figure FDA0002625458910000017
is hiThe conjugate transpose of (1); cumulant matrix C1、C2Respectively expressed as:
Figure FDA0002625458910000018
Figure FDA0002625458910000019
wherein M ═ M, -M +1,. times, M,
Figure FDA00026254589100000110
the eigenvectors of (a) span a signal subspace;
calculating a three-dimensional angle of arrival of the received signal;
Figure FDA0002625458910000021
Figure FDA0002625458910000022
Figure FDA0002625458910000023
wherein, the lambda is the signal wavelength,
Figure FDA0002625458910000024
angle beta between the arriving signal representing the ith direction of arrival and the x-axisiIs determined by the estimated value of (c),
Figure FDA0002625458910000025
angle theta between the arrival signal representing the ith direction of arrival and the y-axisiIs determined by the estimated value of (c),
Figure FDA0002625458910000026
angle between arrival signal representing ith direction of arrival and z-axis
Figure FDA0002625458910000027
Is estimated byA value;
Figure FDA0002625458910000028
wherein h isj(M) denotes the mth element of the jth feature vector, M being less than M;
performing cumulant frequency shift estimation on the received signal to obtain a cumulant frequency shift estimation value, including: calculating the fourth-order cumulant of the received signal containing the Doppler shift as:
c4z＝E[z2(t)z2(t+τ)]-E[z2(t)]E[z2(t+τ)]-2E2[z(t)z(t+τ)] (30)
wherein z (t) is a received signal, and τ is a time delay;
correcting the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value, wherein the method comprises the following steps: utilizing a Doppler frequency estimator to correct the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value; the Doppler frequency estimator is as follows:
Figure FDA0002625458910000029
wherein f is the frequency offset range of FFT output, T is the sampling period of the signal, θ is the estimation angle shown in formula (18), and Ω4zAnd (2, f) is a cyclic spectrum of the fourth-order cumulant of the received signal.
2. The method of claim 1, wherein the antenna array direction is defined as y-axis, and the antenna array X is defined as X-axisaThe central antenna is defined as a coordinate origin O, the two groups of antenna arrays are increased from-M to M in the positive direction of the Y axis by taking the central antenna as zero, the two rows of uniform linear arrays are aligned in parallel, and the coordinate origin O points to the antenna array YaThe direction of the central antenna is defined as x-axis, and the x-axis and the y-axis are defined according to the rule of right-hand coordinate systemThe z-axis.
3. The method of claim 1, further comprising: reading Doppler frequency shift estimation quantity according to a sending end estimator;
the estimator based on the pilot frequency of the sending end is as follows:
Figure FDA0002625458910000031
the estimator without pilot frequency at the transmitting end is as follows:
Figure FDA0002625458910000032
4. the method according to claim 1, wherein the received signal is processed for one path of the received signal to obtain a fourth order cumulative amount of the received signal, the fourth order cumulative amount of the received signal is multiplied by a plurality of sine wave signals within a certain frequency range to obtain a plurality of corresponding demodulated signals, a frequency with the maximum correlation is selected from the plurality of demodulated signals as a cyclic frequency for extracting doppler shift, the sine wave signal with the cyclic frequency is multiplied by the received signal, an expectation is obtained for the obtained product signal, and then fast fourier transform is performed to obtain a cyclic spectrum of the fourth order cumulative amount of the received signal.
5. The method of claim 1, wherein the calculation of cumulative magnitude angle estimate and the calculation of cumulative magnitude frequency shift estimate are performed for all received signals for each antenna of the antenna array, and then the doppler shift estimate is calculated.
6. A Doppler frequency shift estimation device, characterized in that, an antenna array for receiving a received signal comprises Xa、YaTwo uniform linear arrays of two columns, each column containing a numberThe quantity is 2M +1 isotropic antennas, and the distance between every two antennas in the same column is dxThe spacing between two columns being dyEstablishing an XYZ coordinate system based on the antenna array; the device comprises:
the first calculation module is used for carrying out cumulant angle estimation on the received signals to obtain cumulant angle estimation values, and comprises:
determining the cumulative amount of received signals: the cumulative amount is defined as:
Figure FDA0002625458910000033
Figure FDA0002625458910000034
xm(t)、ym(t) is the received signal of antenna m, x, at any time t0(t) is the received signal of the central antenna,
Figure FDA0002625458910000041
is x0(ii) the conjugate signal of (t),
Figure FDA0002625458910000042
is xm(t) conjugate signal;
calculating an accumulation quantity matrix of the received signals according to the accumulation quantity:
carrying out eigenvalue decomposition on the cumulant matrix;
Figure FDA0002625458910000043
wherein, C2In the form of a matrix of accumulated quantities,
Figure FDA0002625458910000044
is a cumulant matrix C1L is the sum of paths existing under multipath conditionsNumber xiiIs the eigenvalue corresponding to the ith path, hiAs a characteristic value xiiThe corresponding feature vector is used as a basis for determining the feature vector,
Figure FDA0002625458910000045
is hiThe conjugate transpose of (1); cumulant matrix C1、C2Respectively expressed as:
Figure FDA0002625458910000046
Figure FDA0002625458910000047
wherein M ═ M, -M +1,. times, M,
Figure FDA0002625458910000048
the eigenvectors of (a) span a signal subspace;
calculating a three-dimensional angle of arrival of the received signal;
Figure FDA0002625458910000049
Figure FDA00026254589100000410
Figure FDA00026254589100000411
wherein, the lambda is the signal wavelength,
Figure FDA00026254589100000412
angle beta between the arriving signal representing the ith direction of arrival and the x-axisiIs determined by the estimated value of (c),
Figure FDA00026254589100000413
angle theta between the arrival signal representing the ith direction of arrival and the y-axisiIs determined by the estimated value of (c),
Figure FDA00026254589100000414
angle between arrival signal representing ith direction of arrival and z-axis
Figure FDA00026254589100000415
An estimated value of (d);
Figure FDA0002625458910000051
wherein h isj(M) denotes the mth element of the jth feature vector, M being less than M;
a second calculating module, configured to perform cumulative frequency shift estimation on the received signal to obtain a cumulative frequency shift estimated value, where the second calculating module includes: calculating the fourth-order cumulant of the received signal containing the Doppler shift as:
c4z＝E[z2(t)z2(t+τ)]-E[z2(t)]E[z2(t+τ)]-2E2[z(t)z(t+τ)] (30)
wherein z (t) is a received signal, and τ is a time delay;
a third calculating module, configured to correct the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected doppler frequency shift estimation value, where the third calculating module includes: utilizing a Doppler frequency estimator to correct the cumulant angle estimation value and the cumulant frequency shift estimation value to obtain a corrected Doppler frequency shift estimation value; the Doppler frequency estimator is as follows:
Figure FDA0002625458910000052
where f is the frequency offset range of FFT output, T is the sampling period of the signal, and θ isEstimate angle, Ω, shown in equation (18)4zAnd (2, f) is a cyclic spectrum of the fourth-order cumulant of the received signal.
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Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5327893A (en) * 1992-10-19 1994-07-12 Rensselaer Polytechnic Institute Detection of cholesterol deposits in arteries
CN103926573A (en) * 2014-04-17 2014-07-16 哈尔滨工程大学 Mono-static MIMO radar distribution type target angle estimation method based on fourth-order cumulant
CN104330783A (en) * 2014-11-05 2015-02-04 河海大学 High-order cumulant based bistatic MIMO (Multiple Input Multiple Output) radar parameter estimation method
CN106597407A (en) * 2016-12-06 2017-04-26 西安电子科技大学 Combined cooperative multi-satellite weak echo signal time delay and doppler frequency shift estimation method

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP6415288B2 (en) * 2014-03-03 2018-10-31 三菱電機株式会社 Radar equipment
US20180106889A1 (en) * 2016-10-14 2018-04-19 Lockheed Martin Corporation System and method for radar based threat determination and classification
CN108709552A (en) * 2018-04-13 2018-10-26 哈尔滨工业大学 A kind of IMU and GPS tight integration air navigation aids based on MEMS

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5327893A (en) * 1992-10-19 1994-07-12 Rensselaer Polytechnic Institute Detection of cholesterol deposits in arteries
CN103926573A (en) * 2014-04-17 2014-07-16 哈尔滨工程大学 Mono-static MIMO radar distribution type target angle estimation method based on fourth-order cumulant
CN104330783A (en) * 2014-11-05 2015-02-04 河海大学 High-order cumulant based bistatic MIMO (Multiple Input Multiple Output) radar parameter estimation method
CN106597407A (en) * 2016-12-06 2017-04-26 西安电子科技大学 Combined cooperative multi-satellite weak echo signal time delay and doppler frequency shift estimation method

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
基于四阶循环平稳的STBC-OFDM信号盲识别;黄波 等;《信号处理》;20170930;全文 *

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