CN105763119B - A kind of control system and control method of CMG frames permanent magnet synchronous motor - Google Patents
A kind of control system and control method of CMG frames permanent magnet synchronous motor Download PDFInfo
- Publication number
- CN105763119B CN105763119B CN201610248632.9A CN201610248632A CN105763119B CN 105763119 B CN105763119 B CN 105763119B CN 201610248632 A CN201610248632 A CN 201610248632A CN 105763119 B CN105763119 B CN 105763119B
- Authority
- CN
- China
- Prior art keywords
- module
- motor
- rotating speed
- permanent magnet
- phase
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/06—Rotor flux based control involving the use of rotor position or rotor speed sensors
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
Abstract
The present invention relates to the high performance control systems and control method of a kind of CMG (control-moment gyro) frame permanent magnet synchronous motor, more particularly to a kind of CMG frames permanent magnet synchronous motor rotary speed precision and the control system and control method of bandwidth, suitable for all kinds of high performance servo systems based on permanent magnet synchronous motor including control-moment gyro frame system.Sliding formwork control is combined by the method for the present invention with rotor flux-orientation vector control method can improve the rejection ability of the non-linear low frequency disturbance torque that system introduces the factors such as gyroscopic couple, rotor static unbalance, the robustness of lift frame rotating speed control performance and system.
Description
Technical Field
The invention relates to a high-performance control system and a control method of a CMG (control moment gyro) frame permanent magnet synchronous motor, in particular to a control system and a control method of the rotation speed precision and the bandwidth of the CMG frame permanent magnet synchronous motor, which are suitable for various high-performance servo systems based on the permanent magnet synchronous motor, including a control moment gyro frame system.
Background
The Control Moment Gyroscope (CMG) is one of inertia executing components of a spacecraft, is mainly applied to attitude control of large-scale spacecrafts and quick maneuvering of small and medium-scale spacecrafts, and has wide development prospect. With the development of aerospace technology, a new satellite platform puts extremely high requirements on the control precision, the control stability, the control bandwidth and other properties of a control moment gyroscope frame. As a key technical bottleneck of a novel high-precision satellite platform, the research on the high-performance control method for controlling the moment gyro frame is of great significance and urgent need.
The control performance of the frame rotating speed of the control moment gyroscope directly influences the output moment performance of the whole control moment gyroscope, and is one of the most key indexes of the control moment gyroscope. In the working process of the frame servo system, the components of the disturbance torque received by the frame servo system are complex, the disturbance amplitude is large, the low-frequency disturbance torque introduced by the factors such as gyro torque, rotor static unbalance and friction and the high-frequency disturbance torque introduced by high-speed rotor unbalance are included, and the control algorithm is required to have strong disturbance inhibition capability. In particular, the high-frequency disturbance generated by the high-speed rotor, the frequency of which is far greater than the bandwidth of the controller, needs to design a special compensation algorithm for the high-frequency disturbance of the rotor. In addition, when the frame servo system changes in conditions such as mechanics and heat, system parameters change, and a control algorithm is required to have strong parameter adaptability. The high-performance control method is a main means for improving the rotating speed control performance of the frame servo system. The existing method improves the performance of a frame servo system, but the adaptability to parameter change and the suppression effect on disturbance torque fluctuation still need to be improved, and the suppression capability on high-frequency disturbance torque fluctuation is obviously insufficient. Therefore, intensive research needs to be carried out on a control method of a control moment gyro frame servo system so as to meet the performance requirement of a control moment gyro.
Disclosure of Invention
The technical problem to be solved by the invention is as follows: the control method comprises a rotating speed closed-loop control method combining sliding mode control and rotor magnetic field directional vector control, a high-speed rotor disturbance self-adaptive feedforward compensation method and position and rotating speed feedforward. By combining the sliding mode control with the rotor magnetic field orientation vector control method, the suppression capability of the system on the nonlinear low-frequency interference torque introduced by the gyro torque, the rotor static imbalance and other factors can be improved, and the frame rotating speed control performance and the system robustness are improved; the self-adaptive feedforward compensation of the disturbance vibration of the high-speed rotor is added, so that the high-frequency disturbance torque can be effectively inhibited, and the frame control precision is improved. And the frame control bandwidth can be further improved by adding a position and rotating speed feedforward design.
The technical solution of the invention is as follows:
a control system of a CMG frame permanent magnet synchronous motor comprises a full-bridge drive circuit module, a current acquisition module, a three-phase/two-phase coordinate transformation module, a two-phase rotating coordinate transformation module, a first difference module, an exciting current proportional-integral adjustment module, a two-phase rotating coordinate inverse transformation module, a space vector pulse width modulation generation module, a rotating speed calculation module, a second difference module, a position loop proportion differential adjustment module, a speed feedforward module, a third difference module, a rotating speed loop sliding mode control module, a position feedforward module, a high-speed rotor disturbance self-adaptive feedforward compensation module, an accumulation module, a permanent magnet synchronous motor and an angle measurement module;
the angle measurement module measures to obtain the actual mechanical angle theta of the permanent magnet synchronous motor m And electrical angle theta e And calculating the actual mechanical angle theta m Output to the second difference module and the rotating speed calculation module to obtain the electrical angle theta e The two-phase rotation coordinate transformation module and the two-phase rotation coordinate inverse transformation module are output;
the current acquisition module acquires A-phase current i of the permanent magnet synchronous motor A And phase B current i B Then, the current acquisition module acquires the current i A And i B Outputting the current to a three-phase/two-phase coordinate transformation module which transforms the received current i A And i B After three-phase/two-phase coordinate transformation, current i under a two-phase static coordinate system (alpha-beta coordinate system) is output α And i β For two-phase rotation coordinate transformation module, two-phase rotation coordinate transformationModule is traded and PMSM electrical angle theta output by angle measuring module is received simultaneously e (ii) a Two-phase rotating coordinate transformation module for received current i α And i β And an electrical angle theta e After two-phase rotating coordinate transformation, the exciting current i under the d-q coordinate system is output d For the first difference module, the first difference module receives the current set value at the same timeAnd is provided withExciting current i under d-q coordinate system d Anddifferentiating in a first difference module (i.e. differentiating)) Then the difference result (namely the exciting current deviation) is output to an exciting current proportional-integral regulating module, and the exciting current proportional-integral regulating module carries out proportional-integral regulation on the received difference result and then outputs a d-axis voltage control signalGiving a two-phase rotating coordinate inverse transformation module;
the second difference module receives the actual mechanical angle theta of the permanent magnet synchronous motor measured by the angle measuring module m And the set mechanical angle value of the permanent magnet synchronous motorAnd to the actual mechanical angle theta m And setting the mechanical angle valueCarrying out difference to obtain a position deviation valueAnd output to the position loop ratioA position loop proportional differential adjustment module for receiving the position offset valueOutputting a rotating speed ring rotating speed setting signal after proportional differential regulationA third difference module;
the rotating speed calculation module obtains the actual mechanical angle theta of the permanent magnet synchronous motor according to the measurement of the angle measurement module m Calculating to obtain the actual rotating speed omega of the permanent magnet synchronous motor m And output to the third difference module;
the third difference module sets the rotating speed of the rotating speed ring receivedAnd the actual rotating speed omega of the permanent magnet synchronous motor m Carrying out difference to obtain a deviation value of the rotating speedThe third difference module is used for deviating the rotating speed valueRespectively output to a rotating speed ring sliding mode control module, a speed feedforward module and a high-speed rotor disturbance self-adaptive feedforward compensation module;
the rotating speed ring sliding mode control module is used for receiving the rotating speed deviation valuePerforming sliding mode control operation to obtain a motor q-axis voltage control signal u q1 Then output to the accumulation module;
received rotating speed deviation value of high-speed rotor disturbance self-adaptive feedforward compensation moduleCarrying out self-adaptive feedforward compensation operation to obtain the q-axis voltage control of the motorSignal u q2 Then output to the accumulation module;
position feedforward module to received position deviation valueCarrying out proportional operation to obtain a motor q-axis voltage control signal u q3 Then output to the accumulation module;
speed feedforward module receives deviation value of rotating speedCarrying out proportional operation to obtain a q-axis voltage control signal u of the motor q4 Then output to the accumulation module;
the accumulation module is used for controlling the received q-axis voltage of the motor to a signal u q1 、u q2 、u q3 、u q4 After accumulation calculation, the q-axis control voltage of the motor is outputGiving a two-phase rotating coordinate inverse transformation module;
two-phase rotating coordinate inverse transformation module is used for receiving the electrical angle theta e D-axis voltage control signalAnd motor q-axis control voltageAfter two-phase rotating coordinate inverse transformation is carried out, control voltage under an alpha-beta coordinate system is outputAndthe space vector pulse width modulation generation module is used for generating the control voltage under the received alpha-beta coordinate systemAndand after generating the space vector pulse width modulation signal, outputting a modulation signal SVPWM to the full-bridge driving circuit module, and controlling the permanent magnet synchronous motor by the full-bridge driving circuit module according to the received modulation signal SVPWM.
A control method of a CMG frame permanent magnet synchronous motor comprises a rotating speed closed-loop control method combining sliding mode control and rotor magnetic field directional vector control, a high-speed rotor disturbance self-adaptive feedforward compensation method and position and rotating speed feedforward.
The rotating speed closed-loop control method comprises a motor d-axis control method and a motor q-axis control method;
the motor d-axis control method comprises the following steps:
(1.1) three-phase/two-phase coordinate transformation module according to current i of permanent magnet synchronous motor A And i B Calculating the current i under a two-phase static coordinate system alpha-beta α And i β According to the electrical angle theta of the motor e Calculating the exciting current i under a two-phase rotating coordinate system d-q d ；
(1.2) setting the exciting currentAnd (3) calculating the excitation current i obtained in the step (1.1) d Differentiating to obtain exciting current deviation, and calculating d-axis voltage control signal of motor by proportional-integral regulator in exciting current controller according to exciting current deviation Is 0;
the method for controlling the q axis of the motor comprises the following steps:
(2.1) according to the set rotation speed omega set Calculating the set mechanical angle of the permanent magnet synchronous motorValue ofAnd calculating the set mechanical angle valueActual mechanical angle theta of the permanent magnet synchronous motor measured by the angle measuring module m Carrying out difference to obtain position deviation, and calculating a rotating speed setting signal of the rotating speed ring by the position ring proportional differential adjusting module according to the position deviation
(2.2) obtaining the actual rotating speed omega of the motor by using an angular acceleration interference observation method m And disturbance torque T f ；
The angular acceleration interference observation method comprises the following specific steps: the Kalman filter carries out filtering processing on the actual position information of the permanent magnet synchronous motor output by the angular position sensor to obtain a rotating speed signal omega of the permanent magnet synchronous motor m Speed signal omega m The angular acceleration alpha of the permanent magnet synchronous motor is obtained after the processing of the nonlinear differentiator, and the disturbance observer obtains the angular acceleration alpha and the torque current i of the permanent magnet synchronous motor q And using the formula T f ＝k e ·i q Calculating the disturbance torque T of the permanent magnet synchronous motor by J & alpha f ；
Wherein, K e Is the moment coefficient of the motor, and J is the rotational inertia of the motor;
(2.3) the rotating speed ring sliding mode control module sets a signal according to the rotating speed output by the position ring proportion differential regulation module in the step (2.1)And the rotation speed signal omega obtained in the step (2.2) m And disturbance torque T f Calculating the q-axis voltage control signal u of the motor q1 Q-axis voltage control signal u of motor q1 The calculation steps are as follows:
(a) Calculating the variable x 1 、x 2 ，
(b) According to variable x 1 、x 2 Obtaining a switching plane s of the sliding mode controller,
(c) Using variable x 1 、x 2 Calculating a motor q-axis voltage control signal u of the drive circuit by a switching plane s of the sliding mode controller q1 ：
Wherein k is v Is the counter potential coefficient of the motor, J is the rotational inertia of the rotor of the motor, r is the two-phase resistance of the motor, k e For the motor moment coefficient, q is a positive odd number, p is a positive odd number, and p&Q, c are regulating coefficients of the switching plane,for setting the signal for the speed, T f As disturbance torque, k s Controlling the proportionality coefficient, k, for sliding modes s &0, delta is sliding mode control switching coefficient, delta>, 0, sign(s) is a sign function of s.
The self-adaptive feedforward compensation method comprises the following steps:
(3.1) calculating an adaptive parameter a according to the negative gradient adaptive law i (k) And b i (k)，
a i (k)＝a i (k-1)+α i e(k)cos(2πf hi kT s +ξ)
b i (k)＝b i (k-1)+α i e(k)sin(2πf hi kT s +ξ)
Wherein alpha is i For adaptive gain, i =1,2,3,. M, m being a natural number; for example, α when i =1 1 For adaptive gain at fundamental frequency, α when i =2 2 Is 2 frequency multiplication adaptive gain;k =1,2,3,. N, n is a natural number; k =1, a i (0)＝0，b i (0) =0; e (k) is the deviation of the rotational speed, f hi Representing the rotor fundamental frequency and the frequency multiplication, e.g. f, when i =1 h1 Represents the rotor fundamental frequency, i =2, f h2 Representing rotor 2 frequency multiplication, T s Is a sampling period, and xi is a phase lead coefficient;
(3.2) calculating the adaptive feedforward compensation control voltage u q2 ，u q2 (k) Is u q2 Is represented by a discretization of the period T s I.e. sampling period, i.e.
Wherein, a i (k) And b i (k) For adapting the parameters, α i For adaptive gain, beta i E (k) is the rotation speed deviation, xi, alpha i And beta i For three important parameters in adaptive feedforward control, xi determines the phase angle margin of the system, alpha i Determining the convergence speed, beta i Affecting the transient response.
The position and rotating speed feedforward method comprises the following steps:
(4.1) the position feedforward module carries out proportional operation on the position deviation obtained in the step (2.1) to generate a motor q-axis voltage control signal u q3 ；
(4.2) the speed feedforward module carries out proportional operation on the rotating speed deviation e (k) to generate a motor q-axis voltage control signal u q4 Wherein the rotating speed deviation e (k) is a rotating speed setting signal output by the position ring proportional-derivative adjusting moduleWith the rotational speed signal omega m A difference of (d);
the q-axis voltage control signal of the motor is obtained through the operation of the three partsComprises the following steps:
v * q ＝u q1 +u q2 +u q3 +u q4 。
motor d-axis voltage control signalAnd motor q-axis voltage control signalObtained by two-phase rotation coordinate inverse transformationAndaccording to obtainingAndSVPWM (space vector pulse width modulation) signals for controlling the permanent magnet synchronous motor can be generated by an SVPWM (space vector pulse width modulation) generation method, and the high-performance control of the CMG (control moment gyro) frame permanent magnet synchronous motor is ensured.
Compared with the prior art, the invention has the advantages that:
(1) The method combines the sliding mode control with the rotor magnetic field orientation vector control method, so that the suppression capability of the system on the nonlinear low-frequency interference torque introduced by the gyro torque, the rotor static imbalance and other factors can be improved, and the frame rotating speed control performance and the robustness of the system are improved;
(2) The method of the invention can effectively inhibit the high-frequency disturbance torque by adding the self-adaptive feedforward compensation of the high-speed rotor disturbance, and improve the frame control precision, xi, alpha i And beta i For three important parameters in adaptive feedforward control, xi determines the phase angle margin of the system, alpha i Determining the convergence speed, beta i Affecting the transient response;
(3) The method can further improve the frame control bandwidth by adding the position and rotating speed feedforward design;
(4) The method has strong inhibition capability on the nonlinear low-frequency interference torque introduced by the gyro moment and other factors and the high-frequency disturbance torque introduced by the unbalanced vibration of the rotor, and can effectively improve the control performance of the control moment gyro frame.
Drawings
FIG. 1 is a schematic diagram of the system of the present invention;
FIG. 2 is a block diagram of the method of the present invention.
Detailed Description
The invention is further described with reference to the following figures and specific embodiments.
A control system of a CMG frame permanent magnet synchronous motor comprises a full-bridge drive circuit module, a current acquisition module, a three-phase/two-phase coordinate transformation module, a two-phase rotating coordinate transformation module, a first difference module, an exciting current proportional-integral adjustment module, a two-phase rotating coordinate inverse transformation module, a space vector pulse width modulation generation module, a rotating speed calculation module, a second difference module, a position loop proportion differential adjustment module, a speed feedforward module, a third difference module, a rotating speed loop sliding mode control module, a position feedforward module, a high-speed rotor disturbance self-adaptive feedforward compensation module, an accumulation module, a permanent magnet synchronous motor and an angle measurement module;
the angle measurement module measures to obtain the actual mechanical angle theta of the permanent magnet synchronous motor m And electrical angle theta e And calculating the actual mechanical angle theta m Output to the second difference module and the rotating speed calculation module to obtain the electrical angle theta e The two-phase rotation coordinate transformation module and the two-phase rotation coordinate inverse transformation module are output;
the current acquisition module acquires phase A current i of the permanent magnet synchronous motor A And phase B current i B Then, the current collecting module collects the current i A And i B Output to a three-phase/two-phase coordinate transformation module for three-phase/two-phase coordinate transformationThe module receives the current i A And i B After three-phase/two-phase coordinate transformation, current i under a two-phase static coordinate system (alpha-beta coordinate system) is output α And i β The two-phase rotating coordinate transformation module is used for simultaneously receiving the electrical angle theta of the permanent magnet synchronous motor output by the angle measuring module e (ii) a Two-phase rotating coordinate transformation module for received current i α And i β And an electrical angle theta e After two-phase rotating coordinate transformation, the exciting current i under the d-q coordinate system is output d For the first difference module, the first difference module receives the current set value at the same timeAnd isExciting current i under d-q coordinate system d Anddifferentiating in a first difference module (i.e. differentiating)) Then the difference result (namely the exciting current deviation) is output to an exciting current proportional-integral regulating module, and the exciting current proportional-integral regulating module carries out proportional-integral regulation on the received difference result and then outputs a d-axis voltage control signalGiving a two-phase rotating coordinate inverse transformation module;
the second difference module receives the actual mechanical angle theta of the permanent magnet synchronous motor measured by the angle measuring module m And the set mechanical angle value of the permanent magnet synchronous motorAnd to the actual mechanical angle theta m And setting the mechanical angle valueCarrying out difference to obtain a position deviation valueAnd the position difference is output to a position ring proportional differential regulation module and a position feedforward module, and the position ring proportional differential regulation module receives the position deviation valueOutputting a rotating speed ring rotating speed setting signal after proportional differential regulationFeeding the third difference module;
the rotating speed calculation module obtains the actual mechanical angle theta of the permanent magnet synchronous motor according to the measurement of the angle measurement module m Calculating to obtain the actual rotating speed omega of the permanent magnet synchronous motor m And output to the third difference module;
the third difference module sets the rotating speed of the rotating speed ring receivedAnd the actual rotating speed omega of the permanent magnet synchronous motor m Carrying out difference to obtain a deviation value of the rotating speedThe third difference module is used for deviating the rotating speed valueRespectively output to a rotating speed ring sliding mode control module, a speed feedforward module and a high-speed rotor disturbance self-adaptive feedforward compensation module;
the rotating speed ring sliding mode control module is used for receiving the rotating speed deviation valuePerforming sliding mode control operation to obtain a motor q-axis voltage control signal u q1 Then output to the accumulation module;
high speed rotor disturbance self-adaptationFeedforward compensation module receives rotational speed deviation valueCarrying out self-adaptive feedforward compensation operation to obtain a motor q-axis voltage control signal u q2 Then output to the accumulation module;
position feedforward module to received position deviation valueCarrying out proportional operation to obtain a q-axis voltage control signal u of the motor q3 Then output to the accumulation module;
speed feedforward module receives deviation value of rotating speedCarrying out proportional operation to obtain a q-axis voltage control signal u of the motor q4 Then output to the accumulation module;
the accumulation module is used for controlling the received q-axis voltage of the motor to a signal u q1 、u q2 、u q3 、u q4 After accumulation calculation, the q-axis control voltage of the motor is outputA two-phase rotating coordinate inverse transformation module is provided;
two-phase rotating coordinate inverse transformation module is used for receiving the electrical angle theta e D-axis voltage control signalAnd motor q-axis control voltageAfter two-phase rotating coordinate inverse transformation is carried out, control voltage under an alpha-beta coordinate system is outputAndthe space vector pulse width modulation generation module is used for generating the control voltage under the received alpha-beta coordinate systemAndand after generating the space vector pulse width modulation signal, outputting a modulation signal SVPWM to the full-bridge driving circuit module, and controlling the permanent magnet synchronous motor by the full-bridge driving circuit module according to the received modulation signal SVPWM.
A control method of a CMG frame permanent magnet synchronous motor comprises a rotating speed closed-loop control method combining sliding mode control and rotor magnetic field directional vector control, a high-speed rotor disturbance self-adaptive feedforward compensation method and position and rotating speed feedforward.
The rotating speed closed-loop control method comprises a motor d-axis control method and a motor q-axis control method;
the motor d-axis control method comprises the following steps:
(1.1) three-phase/two-phase coordinate transformation module according to current i of permanent magnet synchronous motor A And i B Calculating the current i under the two-phase static coordinate system alpha-beta α And i β And then according to the electrical angle theta of the motor e Calculating the exciting current i under a two-phase rotating coordinate system d-q d ；
(1.2) setting the exciting currentAnd (3) calculating the excitation current i obtained in the step (1.1) d Carrying out difference to obtain exciting current deviation, and calculating d-axis voltage control signal of motor by proportional-integral regulator in exciting current controller according to exciting current deviation Is 0;
the transfer function of the excitation current controller is:
G i (s)＝k p_i +k i_i /s＝1+0.001/s
the method for controlling the q axis of the motor comprises the following steps:
(2.1) according to the set rotation speed omega set Calculating the set mechanical angle value of the permanent magnet synchronous motorAnd calculating the set mechanical angle valueActual mechanical angle theta of the permanent magnet synchronous motor measured by the angle measuring module m Carrying out difference to obtain position deviation, and calculating the rotating speed setting signal of the rotating speed ring by the position ring proportional-differential adjusting module according to the position deviation
(2.2) obtaining the actual rotating speed omega of the motor by using an angular acceleration interference observation method m And disturbance torque T f ；
The angular acceleration interference observation method comprises the following specific steps: the Kalman filter carries out filtering processing on the actual position information of the permanent magnet synchronous motor output by the angular position sensor to obtain a rotating speed signal omega of the permanent magnet synchronous motor m Speed signal ω m The angular acceleration alpha of the permanent magnet synchronous motor is obtained after the processing of the nonlinear differentiator, and the disturbance observer obtains the angular acceleration alpha and the moment current i of the permanent magnet synchronous motor q And using the formula T f ＝k e ·i q Calculating the disturbance torque T of the permanent magnet synchronous motor by J & alpha f ；
Wherein, K e Is the moment coefficient of the motor, and J is the rotational inertia of the motor;
(2.3) the rotating speed ring sliding mode control module sets a signal according to the rotating speed output by the position ring proportion differential regulation module in the step (2.1)And the rotating speed signal omega obtained in the step (2.2) m And disturbance torque T f Calculating the q-axis voltage control signal u of the motor q1 Q-axis voltage control signal u of motor q1 The calculation steps are as follows:
(a) Calculating a variable x 1 、x 2 ，
(b) According to the variable x 1 、x 2 Obtaining a switching plane s of the sliding mode controller,
(c) Using variable x 1 、x 2 Calculating a motor q-axis voltage control signal u of the drive circuit by a switching plane s of the sliding mode controller q1 ：
Wherein, ω is m Is a rotation speed signal, J is the rotational inertia of the motor (1 kgm is taken) 2 )，T f Is disturbance torque, k e The motor torque coefficient (2 Nm/A), r is the motor two-phase resistance (r =3.0 omega), k v Is the motor back emf coefficient (taken as 2V/rpm), c is the adjustment coefficient of the switching plane (c = 1.0), p is a positive odd number (p = 1), q is a positive odd number (q = 1), and p is a positive odd number (q = 1)<q，k>0(k＝0.8)，δ>, 0 (δ = 1.0), sign(s) is the sign function of s.
The self-adaptive feedforward compensation method comprises the following steps:
(3.1) calculating an adaptive parameter a according to the negative gradient adaptive law i (k) And b i (k)，
a i (k)＝a i (k-1)+α i e(k)cos(2πf hi kT s +ξ)
b i (k)＝b i (k-1)+α i e(k)sin(2πf hi kT s +ξ)
Wherein, T s To adopt a period (T) s =100 μ s), ξ is the phase lead coefficient (ξ = π/6), f hi Representing the rotor fundamental frequency and the frequency multiplication, i is 1-3.i =1 time f h1 Denotes the rotor fundamental frequency, f when i =2 h2 Representing rotor 2 octave, f when i =3 h2 Representing rotor 3 frequency multiplication, and e (k) is rotation speed deviation; alpha (alpha) ("alpha") i Taking i as frequency multiplication number, and taking i as 1-3; α when i =1 1 For adaptive gain at fundamental frequency, α when i =2 2 Is 2 frequency multiplication adaptive gain; α when i =3 3 3, frequency multiplication adaptive gain; k =1,2,3, \ 8230, n, n is a natural number; k =1, a i (0)＝0，b i (0)＝0，e(0)＝0；
(3.2) calculating the adaptive feedforward compensation control voltage u q2 ，u q2 (k) Is u q2 Is represented by a discretization of the period T s I.e. sampling period, i.e.
Wherein, a i (k) And b i (k) For adapting the parameters, α i For adaptive gain (alpha) i ＝3),β i Is a feed forward coefficient (beta) i = 0.5), e (k) is the rotational speed deviation ξ, α i And beta i For three important parameters in adaptive feedforward control, xi determines the phase angle margin of the system, alpha i Determining the convergence speed, beta i Affecting the transient response;
the position and rotating speed feedforward method comprises the following steps:
(4.1) the position feedforward module carries out proportional operation on the position deviation obtained in the step (2.1) to generate a motor q-axis voltage control signal u q3 (ii) a The scaling factor was taken to be 0.05.
(4.2) the speed feedforward module carries out proportional operation on the rotating speed deviation e (k) to generate a motor q-axis voltage control signal u q4 Wherein the rotating speed deviation e (k) is a position ring proportional differential regulating moduleOutput rotation speed setting signalWith the rotation speed signal omega m A difference value of (a); the scaling factor was taken to be 0.08.
The q-axis voltage control signal of the motor is obtained through the operation of the three partsComprises the following steps:
v * q ＝u q1 +u q2 +u q3 +u q4 。
motor d-axis voltage control signalAnd motor q-axis voltage control signalObtained by two-phase rotation coordinate inverse transformationAndaccording to obtainingAndSVPWM (space vector pulse width modulation) signals for controlling the permanent magnet synchronous motor can be generated by an SVPWM (space vector pulse width modulation) generation method, and the high-performance control of the CMG (control moment gyro) frame permanent magnet synchronous motor is ensured.
The present invention has not been described in detail as is known to those skilled in the art.
Claims (4)
1. The utility model provides a control system of CMG frame PMSM which characterized in that: the control system comprises a full-bridge driving circuit module, a current acquisition module, a three-phase/two-phase coordinate transformation module, a two-phase rotation coordinate transformation module, a first difference module, an exciting current proportional-integral regulation module, a two-phase rotation coordinate inverse transformation module, a space vector pulse width modulation generation module, a rotating speed calculation module, a second difference module, a position loop proportional differential regulation module, a speed feedforward module, a third difference module, a rotating speed loop sliding mode control module, a position feedforward module, a high-speed rotor disturbance self-adaptive feedforward compensation module, an accumulation module, a permanent magnet synchronous motor and an angle measurement module;
the angle measurement module measures to obtain the actual mechanical angle theta of the permanent magnet synchronous motor m And electrical angle theta e And the actual mechanical angle theta m Output to the second difference module and the rotating speed calculation module to obtain the electrical angle theta e The two-phase rotation coordinate transformation module and the two-phase rotation coordinate inverse transformation module are output;
the current acquisition module acquires phase A current i of the permanent magnet synchronous motor A And phase B current i B Then, the current collecting module collects the current i A And i B Outputting the current to a three-phase/two-phase coordinate transformation module which transforms the received current i A And i B Outputting current i under a two-phase static coordinate system after three-phase/two-phase coordinate transformation α And i β The two-phase rotating coordinate transformation module is used for simultaneously receiving the electrical angle theta of the permanent magnet synchronous motor output by the angle measuring module e (ii) a Two-phase rotating coordinate transformation module for received current i α And i β And an electrical angle theta e After two-phase rotating coordinate transformation, the exciting current i under the d-q coordinate system is output d For the first difference module, the first difference module receives the current set value at the same timeAnd isExciting current i under d-q coordinate system d Anddifference is obtained in a first difference moduleThen the difference result is output to an exciting current proportional-integral regulating module, and the exciting current proportional-integral regulating module carries out proportional-integral regulation on the received difference result and outputs a d-axis voltage control signalGiving a two-phase rotating coordinate inverse transformation module;
the second difference module receives the actual mechanical angle theta of the permanent magnet synchronous motor measured by the angle measuring module m And set mechanical angle value of permanent magnet synchronous motorAnd to the actual mechanical angle theta m And setting the mechanical angle valueCarrying out difference to obtain a position deviation valueAnd the position difference is output to a position ring proportional differential regulation module and a position feedforward module, and the position ring proportional differential regulation module receives the position deviation valueOutputting a rotating speed ring rotating speed setting signal after proportional differential regulationFeeding the third difference module;
the rotating speed calculation module obtains the actual mechanical angle theta of the permanent magnet synchronous motor according to the measurement of the angle measurement module m Calculating to obtain the actual rotating speed of the permanent magnet synchronous motorω m And output to the third difference module;
the third difference module sets the rotating speed of the rotating speed ring receivedAnd the actual rotating speed omega of the permanent magnet synchronous motor m Carrying out difference to obtain a deviation value of the rotating speedThe third difference module is used for deviating the rotating speed valueRespectively output to a rotating speed ring sliding mode control module, a speed feedforward module and a high-speed rotor disturbance self-adaptive feedforward compensation module;
the rotating speed ring sliding mode control module is used for receiving a rotating speed deviation valuePerforming sliding mode control operation to obtain a motor q-axis voltage control signal u q1 Then output to the accumulation module;
received rotating speed deviation value of high-speed rotor disturbance self-adaptive feedforward compensation moduleCarrying out self-adaptive feedforward compensation operation to obtain a motor q-axis voltage control signal u q2 Then output to the accumulation module;
position feedforward module to received position deviation valueCarrying out proportional operation to obtain a q-axis voltage control signal u of the motor q3 Then output to the accumulation module;
the speed feedforward module is used for receiving the deviation value of the rotating speedCarrying out proportional operation to obtain a motor q-axis voltage control signal u q4 Then output to the accumulation module;
the accumulation module is used for controlling the received q-axis voltage of the motor to a signal u q1 、u q2 、u q3 、u q4 After accumulation calculation, the q-axis control voltage of the motor is outputA two-phase rotating coordinate inverse transformation module is provided;
two-phase rotating coordinate inverse transformation module is used for receiving the electrical angle theta e D-axis voltage control signalAnd motor q-axis control voltageAfter two-phase rotating coordinate inverse transformation is carried out, control voltage under an alpha-beta coordinate system is outputAndthe space vector pulse width modulation generation module is used for generating the control voltage under the received alpha-beta coordinate systemAndand after generating the space vector pulse width modulation signal, outputting a modulation signal SVPWM to the full-bridge driving circuit module, and controlling the permanent magnet synchronous motor by the full-bridge driving circuit module according to the received modulation signal SVPWM.
2. A method for controlling a CMG frame permanent magnet synchronous motor using the control system of claim 1, characterized in that: the method comprises a rotating speed closed-loop control method, a high-speed rotor disturbance self-adaptive feedforward compensation method and position and rotating speed feedforward three contents, wherein the rotating speed closed-loop control method is combined with rotor magnetic field directional vector control;
the rotating speed closed-loop control method comprises a motor d-axis control method and a motor q-axis control method;
the motor d-axis control method comprises the following steps:
(1.1) three-phase/two-phase coordinate transformation module according to current i of permanent magnet synchronous motor A And i B Calculating the current i under the two-phase static coordinate system alpha-beta α And i β According to the electrical angle theta of the motor e Calculating the exciting current i under a two-phase rotating coordinate system d-q d ；
(1.2) setting the exciting currentAnd (3) calculating the excitation current i obtained in the step (1.1) d Carrying out difference to obtain exciting current deviation, and calculating d-axis voltage control signal of motor by proportional-integral regulator in exciting current controller according to exciting current deviation Is 0;
the q-axis control method of the motor comprises the following steps:
(2.1) according to the set rotation speed omega set Calculating the set mechanical angle value of the permanent magnet synchronous motorAnd calculating the set mechanical angle valuePermanent magnet measured by angle measuring moduleActual mechanical angle θ of magnetic synchronous motor m Carrying out difference to obtain position deviation, and calculating the rotating speed setting signal of the rotating speed ring by the position ring proportional-differential adjusting module according to the position deviation
(2.2) obtaining the actual rotating speed omega of the motor by using an angular acceleration interference observation method m And disturbance torque T f ；
The angular acceleration interference observation method comprises the following specific steps: the Kalman filter carries out filtering processing on the actual position information of the permanent magnet synchronous motor output by the angular position sensor to obtain a rotating speed signal omega of the permanent magnet synchronous motor m Speed signal omega m The angular acceleration alpha of the permanent magnet synchronous motor is obtained after the processing of the nonlinear differentiator, and the disturbance observer obtains the angular acceleration alpha and the moment current i of the permanent magnet synchronous motor q And using the formula T f ＝k e ·i q Calculating the disturbance torque T of the permanent magnet synchronous motor by J & alpha f ；
Wherein, K e Is the moment coefficient of the motor, J is the rotational inertia of the motor;
(2.3) the rotating speed ring sliding mode control module sets a signal according to the rotating speed output by the position ring proportion differential regulation module in the step (2.1)And the rotating speed signal omega obtained in the step (2.2) m And disturbance torque T f Calculating the q-axis voltage control signal u of the motor q1 Q-axis voltage control signal u of motor q1 The calculation steps are as follows:
(a) Calculating the variable x 1 、x 2 ，
(b) According to variable x 1 、x 2 A switching plane s of the sliding mode controller is obtained,
(c) Using variable x 1 、x 2 Calculating a motor q-axis voltage control signal u of the driving circuit with a switching plane s of the sliding mode controller q1 ：
Wherein k is v Is the counter potential coefficient of the motor, J is the rotational inertia of the rotor of the motor, r is the two-phase resistance of the motor, k e For the motor moment coefficient, q is a positive odd number, p is a positive odd number, and p&Q, c are regulating coefficients of the switching plane,for setting the signal for the speed, T f As disturbance torque, k s Controlling the proportionality coefficient, k, for sliding modes s &0, delta is sliding mode control switching coefficient, delta>, 0, sign(s) is a sign function of s.
3. The control method of the CMG frame permanent magnet synchronous motor according to claim 2, characterized in that: the self-adaptive feedforward compensation method comprises the following steps:
(3.1) calculating an adaptive parameter a according to the negative gradient adaptive law i (k) And b i (k)，
a i (k)＝a i (k-1)+α i e(k)cos(2πf hi kT s +ξ)
b i (k)＝b i (k-1)+α i e(k)sin(2πf hi kT s +ξ)
Wherein alpha is i For adaptive gain, i =1,2,3,. M, m being a natural number; α when i =1 1 For adaptive gain at fundamental frequency, α when i =2 2 Is 2 frequency multiplication adaptive gain; k =1,2,3,. N, n being a natural number; k =1, a i (0)＝0，b i (0) =0; e (k) is the deviation of the rotational speed, f hi Representing the rotor fundamental frequency and the frequency multiplication, i =1, f h1 Denotes the rotor fundamental frequency, i =2, f h2 Representing rotor 2 frequency multiplication, T s Is the sampling period, xi is the phase lead coefficient;
(3.2) calculating the adaptive feedforward compensation control voltage u q2 ，u q2 (k) Is u q2 Is represented by a discretization of s I.e. sampling period, i.e.
Wherein, a i (k) And b i (k) For the adaptive parameter, α i For adaptive gain, beta i E (k) is the rotation speed deviation, xi, alpha i And beta i Three important parameters in the adaptive feedforward control, xi, determine the phase angle margin of the system, alpha i Determining the convergence speed, beta i Affecting the transient response.
4. The control method of a CMG frame permanent magnet synchronous machine according to claim 3, characterized in that: the position and rotating speed feedforward method comprises the following steps:
(4.1) the position feedforward module carries out proportional operation on the position deviation obtained in the step (2.1) to generate a motor q-axis voltage control signal u q3 ；
(4.2) the speed feedforward module carries out proportional operation on the rotating speed deviation e (k) to generate a motor q-axis voltage control signal u q4 Wherein the rotating speed deviation e (k) is a rotating speed setting signal output by the position ring proportional-derivative adjusting moduleWith the rotational speed signal omega m A difference of (d);
the q-axis voltage control signal of the motor is obtained through the operation of the three partsComprises the following steps:
motor d-axis voltage control signalAnd motor q-axis voltage control signalObtained by inverse transformation of two-phase rotating coordinatesAndaccording to obtainingAndthe SVPWM signal for controlling the permanent magnet synchronous motor can be generated by the SVPWM generation method.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201610248632.9A CN105763119B (en) | 2016-04-20 | 2016-04-20 | A kind of control system and control method of CMG frames permanent magnet synchronous motor |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201610248632.9A CN105763119B (en) | 2016-04-20 | 2016-04-20 | A kind of control system and control method of CMG frames permanent magnet synchronous motor |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN105763119A CN105763119A (en) | 2016-07-13 |
| CN105763119B true CN105763119B (en) | 2018-06-01 |
Family
ID=56324439
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201610248632.9A Active CN105763119B (en) | 2016-04-20 | 2016-04-20 | A kind of control system and control method of CMG frames permanent magnet synchronous motor |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN105763119B (en) |
Families Citing this family (14)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN106026834A (en) * | 2016-07-29 | 2016-10-12 | 扬州大学 | Speed sensorless control method of permanent magnet synchronous motor |
| KR102596568B1 (en) * | 2016-08-17 | 2023-11-01 | 현대모비스 주식회사 | Device and method for estimating rotor angle in motor |
| CN106849801B (en) * | 2016-12-20 | 2019-08-27 | 江苏大学 | A kind of induction-type bearingless motor method for estimating rotating speed |
| CN106685296A (en) * | 2016-12-29 | 2017-05-17 | 威灵（芜湖）电机制造有限公司 | Draught fan rotational speed control method and system, and draught fan system |
| JP6707050B2 (en) * | 2017-03-27 | 2020-06-10 | 株式会社日立産機システム | Synchronous motor controller |
| CN107707172B (en) * | 2017-09-14 | 2019-08-09 | 北京控制工程研究所 | A kind of CMG low speed frame quickly wide torque range, high-precision driving control system |
| CN108762096B (en) * | 2018-07-17 | 2020-09-08 | 北京航空航天大学 | Disturbance suppression method for control moment gyro frame system based on discrete nonlinear cascade extended state observer |
| CN110018634B (en) * | 2019-04-28 | 2021-11-16 | 北京控制工程研究所 | Self-adaptive frame control system and method for improving bandwidth of control moment gyroscope |
| CN110456630B (en) * | 2019-08-22 | 2020-07-24 | 北京航空航天大学 | Anti-interference control method for control moment gyro frame servo system |
| CN110943659B (en) * | 2019-10-31 | 2022-01-04 | 北京控制工程研究所 | Laser terminal coarse pointing mechanism working mode identification and position control system |
| CN111030540B (en) * | 2019-12-24 | 2023-05-26 | 新风光电子科技股份有限公司 | Seamless switching method for current source and noninductive vector control of permanent magnet synchronous motor |
| CN111337007B (en) * | 2020-04-09 | 2021-08-10 | 北京控制工程研究所 | Control moment gyro frame zero position stable positioning system and method |
| EP4248558A4 (en) * | 2020-11-17 | 2024-10-02 | Economotor Innovation Ltd | Method and device for synchronous motor control |
| CN112904713B (en) * | 2020-12-31 | 2022-07-19 | 同济大学 | High-precision strong-robustness control method of spatial motion servo mechanism |
Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101709969A (en) * | 2009-11-27 | 2010-05-19 | 北京航空航天大学 | Method for inhibiting moving-gimbal effects of single gimbal magnetically suspended control moment gyroscope |
| CN103825516A (en) * | 2013-12-31 | 2014-05-28 | 清华大学 | Composite controller of synchronization motor |
-
2016
- 2016-04-20 CN CN201610248632.9A patent/CN105763119B/en active Active
Patent Citations (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101709969A (en) * | 2009-11-27 | 2010-05-19 | 北京航空航天大学 | Method for inhibiting moving-gimbal effects of single gimbal magnetically suspended control moment gyroscope |
| CN103825516A (en) * | 2013-12-31 | 2014-05-28 | 清华大学 | Composite controller of synchronization motor |
Non-Patent Citations (1)
| Title |
|---|
| 自抗扰控制力矩陀螺框架角位置伺服系统设计;陈茂胜等;《中国空间科学学会2013年空间光学与机电技术研讨会》;20131231;第305-310页 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN105763119A (en) | 2016-07-13 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN105763119B (en) | A kind of control system and control method of CMG frames permanent magnet synchronous motor | |
| CN110350835B (en) | Permanent magnet synchronous motor position sensorless control method | |
| CN110198150B (en) | Permanent magnet synchronous motor multi-parameter online identification method | |
| CN106208865B (en) | More permanent magnet synchronous motor Virtual-shaft control methods based on Load Torque Observer | |
| CN101964624B (en) | Sensorless control system of permanent magnet synchronous motor | |
| CN109768753B (en) | Novel sliding-mode observer position-sensorless permanent magnet synchronous motor model prediction control method | |
| CN103560735B (en) | Control method for electro-magnetic synchronous motor | |
| CN104901600A (en) | Sensorless control method of permanent magnet synchronous motor in wide rotating speed scope | |
| CN110798116B (en) | Motor vector composite controller based on armature model independent feedforward compensation | |
| Wang et al. | Stability and dynamic performance improvement of speed adaptive full-order observer for sensorless induction motor ultralow speed operation | |
| CN113131817A (en) | Online parameter identification system and method for permanent magnet synchronous motor | |
| CN105048913A (en) | Current compensation-based unbalance vibration control system for bearingless asynchronous motor | |
| CN114598206B (en) | Design method of permanent magnet synchronous motor wide-speed-domain rotor position observer | |
| CN105610369A (en) | Sliding-mode observer based flux linkage observation method of asynchronous motor | |
| Yu et al. | A Stator Current Oriented Closed-Loop $ I\text {--} f $ Control of Sensorless SPMSM With Fully Unknown Parameters for Reverse Rotation Prevention | |
| CN112953335A (en) | Finite time self-adaptive composite control method and system for permanent magnet synchronous motor | |
| CN113890438A (en) | Speed-sensorless control method based on built-in permanent magnet synchronous motor | |
| CN106452256A (en) | Asynchronous motor parameter online correction method based on rotor flux observer | |
| CN115864928A (en) | PMSM model reference self-adaptive rotation speed estimation method based on correction current prediction | |
| CN114465530A (en) | Speed control method and system of permanent magnet synchronous motor | |
| CN106685294B (en) | Asynchronous electromotor rotor resistance and magnetizing inductance decouple antidote | |
| CN113541545A (en) | Fractional order vector control method and system for permanent magnet synchronous motor | |
| CN110492808A (en) | Permanent magnet synchronous motor vector control system and method based on H infinity control | |
| CN113078865B (en) | Built-in permanent magnet synchronous motor sensorless control method | |
| CN111092578B (en) | Vector control method for accurately orienting rotor magnetic field of asynchronous motor |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| C06 | Publication | ||
| PB01 | Publication | ||
| C10 | Entry into substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |